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Low cost 112 G direct detection metro transmission system with reduced bandwidth (10 G) components and MLSE compensation
Optics Communications ∎ (∎∎∎∎) ∎∎∎–∎∎∎
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Optics Communications journal homepage: www.elsevier.com/locate/optcom
Low cost 112 G direct detection metro transmission system with reduced bandwidth (10 G) components and MLSE compensation Albert Gorshtein a,n, Dan Sadot a, Nir Sheffi b, Eduard Sonkin b, Yuval Shachaf b, Don Becker b, Omri Levy b, Gilad Katz b a b
Electrical and Computer Engineering Department, Ben Gurion University of the Negev, Beer Sheva 84105, Israel MultiPhy Ltd., Ness Ziona, Israel
art ic l e i nf o
a b s t r a c t
Article history: Received 18 October 2014 Accepted 5 November 2014
We experimentally demonstrate a 100 G ADC-MLSE-based 4-lanes NRZ direct detection transceiver. The DSP engine enables the use of low cost 10 G optoelectronics and supports 40 km metro transmission suitable for both ASE limited and point-to-point applications. The fully implemented mixed signal ASIC is based on standard 65 nm CMOS process. & 2014 Published by Elsevier B.V.
Keywords: Intensity modulation direct detection Digital signal processing Maximum likelihood sequence estimation (MLSE)
1. Introduction The constant growth in the demand for high bandwidth data transmission leads to higher challenges that should be resolved in the physical layer, and particularly by optical transmission technology. The current high end transmission data rates are in the range of hundreds of Gbits/s. One emerging technology that can support such bitrates for long distances (hundreds of kilometers and above) is coherent transmission and detection [1–8]. On the other hand, direct detection (DD) technology offers the use of lower cost optoelectronic components (O/E), consumes less power and enables overall lower latency solution. These advantages may be critical for metro and short reach applications such as sub-hundred kilometers networks of metro-edge and data centers interconnections. However, direct detection (non-coherent) optical technology is limited to lower bit rates and/or shorter distances. For example, increasing the bitrate from 10 Gbit/s to 25 Gbit/s, results in distance reduction from ∼80 km to ∼15 km, for the same bit error rate (BER) performance. The main reason for this limitation is the inter-symbol interference (ISI) caused by chromatic dispersion (CD) and polarization mode dispersion (PMD). To combat this ISI two techniques are commonly used. The first technique is based on advanced modulation formats, together with partial response signaling [9,10], while the other approach is based on digital signal n
Corresponding author. . E-mail address: [email protected] (A. Gorshtein).
processing (DSP), applying electronic dispersion compensation (EDC). The EDC implementations with maximum likelihood sequence estimation (MLSE) at the receiver (Rx) side, is theoretically the optimal tool to combat ISI, and was very popular for 10 Gbis/s [11,12]. The combination of the two approaches is also possible, and was also theoretically investigated for 4 25 Gbits transmission with the use of reduced bandwidth (BW) components [13]. A more detailed theoretical background on reduced BW components transmission with subsequent MLSE compensation of transmission impairments can be found in [14]. In addition, it should be noted, that in case that nonlinearities occur, the MLSE is capable of coping with self-phase modulation effect, provided that the memory depth of the MLSE is long enough. Here, an inclusive experimental investigation of the 100 Gbit/s transceiver as proposed in [13] is presented and analyzed. All the presented results are based on recorded measurements by means of real time application specific integrated circuit (ASIC). It is shown that the MLSE-based system outperforms its conventional DD counterparts by (a) allowing use of lower cost components, (b) supporting longer reach transmission distances, (c) yielding improved BER results, and (d) doubling the spectral efficiency due to the use of reduced bandwidth components. In particular, transmission over 40 km of a standard single mode fiber (SSMF) is achieved with non-return-to-zero (NRZ) on-off keying (OOK) transmission, supporting an optical signal to noise ratio (OSNR) of 22 dB for a pre-FEC BER of 10 3. The rest of the paper is organized as follows. Section 2 introduces the noise-ISI tradeoff in the MLSE histograms domain. Section 3 describes the high level ASIC architecture and its design
http://dx.doi.org/10.1016/j.optcom.2014.11.015 0030-4018/& 2014 Published by Elsevier B.V.
Please cite this article as: A. Gorshtein, et al., Optics Communications (2014), http://dx.doi.org/10.1016/j.optcom.2014.11.015i
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parameters. Section 4 presents the conducted set of real time experiments via amplified spontaneous emission (ASE) limited transmission system. The setup and real time experimental results of the point-to-point system without optical amplification is given in Section 5. Finally, Section 6 presents conclusive remarks.
2. Noise–ISI tradeoff analysis through the histograms domain Assuming that the optical transmission power is low enough, such that fiber Kerr non-linarites can be neglected, there are two main degradation effects on signal quality in DD systems: ISI and noise. The ISI is generated by CD, PMD, and optical and electrical components BW limitations. The noise, in turn, stems either from optical or electrical amplifiers present in the system, or from the opto-electronic conversion. In ASE limited systems the dominant noise is generated by optical amplifiers [15]. In point-to-point links without optical amplification, however, the dominant noise mechanisms are thermal noise and shot noise [15]. Both phenomena, ISI and noise, can be conveniently represented in the histograms domain as will be illustrated shortly below. The histograms represent the (un-normalized) conditional probability density functions (PDFs), of the received samples rn ,
10GHz Tx, 10GHz Rx,<σ2i > = 1.50, SNRest = 17.6dB
given the transmitted sequence of Nisi consecutive symbols [an , an − 1, ... , an − Nisi+ 1]. The histograms are used by the MLSE decoder in order to obtain the estimated sequence with highest probability that has been transmitted as described in [15]. The Nisi parameter represents the memory depth of the MLSE decoder, and should be chosen such that it exceeds the memory depth of the overall channel (transmitter þfiber þreceiver). Fig. 1(a) and (b) shows examples of histogram sets received in a 28 Gbit/s lane, after a transmission over 25 km amplified optical link with reduced BW (10 GHz) components at the Tx, and 10 GHz vs. 30 GHz components at the Rx side respectively. OSNR of 21 dB and input power of 5 dBm were used in both cases. Each histogram set contains 32 conditional histograms, termed “branches”, corresponding toNisi = 4 . Although the presented histogram shapes do not strictly follow the Gaussian distribution, they are treated as such for the illustration purposes only, to help provide more insight on the noise–ISI tradeoff. As Fig. 1(a) and (b) reveals, there are more than two levels in the received signal, accounting for the amount of ISI present in the system. In case there was no noise in the system, the branches mean values would represent the (non-linear) impulse repose of the channel. However, since the dominant noise source in the presented system configuration is ASE, each branch has a variance, which depends on its mean value.
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Please cite this article as: A. Gorshtein, et al., Optics Communications (2014), http://dx.doi.org/10.1016/j.optcom.2014.11.015i
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It is shown in Fig. 1(c) and (d) that branches with higher mean value also have higher variance (up to a clipping effect introduced by the analog-to-digital conversion). In the ‘noise-free’ scenario, the MLSE is capable of fully compensating for the ISI, provided that the length of the overall channel (Tx þfiberþ Rx) impulse response is less (or equal) than the decoder memory. However, due the presence of signal-dependent noise, the ISI is not fully compensated. This results in performance degradation, and subsequent OSNR (and/or power) penalty. It should be noted, that the MLSE is carrying out the optimal decisions, in the maximum-likelihood sense, at each OSNR (or power) point, selecting the best “compromise” between ISI and noise in terms of optimal bit error rate (BER). To quantify the effect of the noise–ISI tradeoff, the following estimate of the signal-to-noise ratio (SNR) at the MLSE input is proposed:
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where μi and σi2 represent the mean and variance values of each branch, and Nbr = 2Nisi+ 1, is the overall number of branches as proposed in [13]. Comparing the branch histograms in Fig. 1(c) and (d) reveals that the histogram means in both cases are very similar, whereas the variances in case of 1(c) are significantly smaller than in 1(d), while the only difference between the two cases is the Rx bandwidth. It should be noted that in both cases, the signals were sampled by the same analog to digital converter (ADC), with an anti-aliasing filter bandwidth of 14 GHz. The corresponding average noise variances are 1.5 and 2.3 for the 10 GHz and 30 GHz Rx respectively. The corresponding SNRs are 17.6 dB in the reduced BW Rx case vs. 15.8 dB in the full BW Rx case, indicating that the latter will have poorer performance. On the other hand, when OSNR is increased, the system becomes less noise-limited and more ISI limited. In such case, the average noise variance in the full BW (30 GHz) case becomes comparable with the “difference in the residual (uncompensated) ISI” of 10 GHz Rx vs. the 30 GHz Rx cases. Consequently, the BER of the wider bandwidth Rx system will outperform the lower bandwidth Rx case. This result is observed in measurements depicted in Fig. 6. In conclusion, the noise–ISI tradeoff can be applied to optimize the system performance by using reduced BW components that are optimized for a given amount of ISI, MLSE decoder memory depth, and SNR conditions. In the following sections, various systems optimizations are investigated experimentally for a wide range of channel conditions.
3. ASIC architecture and parameters High level architecture of the MP1100Q ASIC and a zoom into its lanes structure is presented in Fig. 2. The MP1100Q DEMUX consists of four 4 28 Gsymbols/s lanes with MLSE based engines. Each MLSE engine follows an analog front end, that contains a digitally-controlled self-calibrated analog to digital converter (ADC) that samples at 28 Gsamples/s, with 5 bit resolution (effective number of bits (ENOB) ∼3.8). In addition, the analog front end contains a digitally-controlled analog clock recovery (CR) and digitally-controlled variable gain amplifier (VGA). The MLSE engine is a 16-state fully implemented Viterbi processor, with an overall latency of under 200 ns. The initialization of the MLSE is fully automatic and follows a unique blind channel acquisition algorithm [15]. In addition, the MLSE is fully adaptive, and is capable of tracking different channel dynamics, in the order of 1 ms. In turn, the 4 28G data stream
coming out of the four lanes is collected to a 4-10 gearbox, that redistributes it into 10 Gbit/s output lanes designated in Fig. 2 as 10 G Tx, according to either IEEE802.3ba CAUI or ITU-T OTU4.10 protocols. Furthermore, MP1100Q has a built-in fully configurable pseudo random bit sequence (PRBS) generator and checker at the output of each 28 G lane that can be used for bit error rate (BER) testing. The ASIC is implemented using 65 nm CMOS technology and has about 15.2 million gates.
4. 4 28Gbit/s transmission in ASE limited network 4.1. Experimental setup The general block diagram of the ASE limited setup, used in various experiments is shown in Fig. 3. The transmit side (Tx) incudes 4 28 Gbit/s signal generator that feeds four off-the-shelf modulator drivers (Mod. drivers), followed by four linear Transmit Optical Sub-Assemblies (TOSAs), which perform the electrical-tooptical conversion. The four optical signals are multiplexed and demultiplexed through a 50 GHz arrayed waveguide gratings (AWG) with 5 dB insertion loss, at transmit and receive sides, respectively. The wavelength division multiplexed (WDM) signal propagates through an SSMF (the length may vary for different experiments). The optical fiber is used to simulate the amount of residual dispersion in metro and regional links. Depending on the network topology and design, the residual dispersion can be either positive or negative. In addition, some networks use symmetric dispersion widows, while others use asymmetric windows. Hence in some experiments, a dispersion compensation fiber (DCF) with negative dispersion coefficient was used. The optical intensity at the fiber output is amplified by erbium doped fiber amplifier (EDFA) and coupled together with an amplified spontaneous emission (ASE) noise source. Variable optical attenuators (VOAs) are used to control the received input power and OSNR value. The OSNR was measured by tapping 5% of the signal into an optical spectrum analyzer (OSA). The receiver (Rx) includes four off-theshelf Receive Optical Sub-Assemblies (ROSAs), which perform the optical-to-electrical conversion, and, in turn, are followed by the MP1100Q MLSE receiver ASIC. As described in the previous section, the MP100Q receives the heavily distorted incoming signal (by optical channel and reduced BW components), recovers the clock, digitizes the signal via the 4 28 Gsamples/s ADCs and recovers the data by the DSP postprocessing engine based on the four MLSE decoders. In the sequel, different scenarios were tested, where the BW of the TOSAs and ROSAs are scenario-dependent. In all the following
Please cite this article as: A. Gorshtein, et al., Optics Communications (2014), http://dx.doi.org/10.1016/j.optcom.2014.11.015i
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Fig. 3. General block diagram of the ASE limited experimental setup.
scenarios, a full symmetry was maintained between the 4-lanse components, both at the Tx and Rx side, i.e., four TOSAs and four ROSAs from the same series were used. The same is true also for the modulator drivers. The difference between the scenarios is the Tx and/or Rx bandwidth, the modulator types, (Mach Zehnder Modulator (MZM) vs. electro-absorption modulator laser (EML)), the modulator chirp parameter (zero vs. negative chirp) and the transmission distance (back-to-back vs. 25 km, vs. 40 km). Hence, the block diagram in Fig. 3 is general and is used to described all the tested scenarios all together. In all the experiments, the optical output power of each TOSA was tuned to be 0 dBm, resulting in 1 dBm of the four subchannels at the AWG output. This optical power was set in order to avoid fiber Kerr nonlinearities. In addition, a pseudo-random bit sequence (PRBS) of length 223 − 1 was used, and the PRBS checkers inside MP1100Q was configured accordingly. The average bit error rate (BER) over the four ASIC lanes was measured. The aggregated transmission rate of 112 Gbit/s (4 28 G) was used in all the experiments, in order to include the forward error correction (FEC) overhead.
4.2. The effect of modulator chirp and residual dispersion, with reduced BW Tx, and full BW Rx In this scenario, reduced BW components were used only in the Tx side. The modulator drivers have 12 GHz BW, while 10 GHz MZMs with zero and negative chirp were examined. At the Rx side full BW (30 GHz) receivers were used. The fiber lengths of 0 km (back to back), 25 km and 40 km were tested. The BER performance vs. OSNR for different fiber lengths and different modulator chirp are presented in Fig. 4. The OSNR is measured per channel according to the formal definition using reference BW of 0.1 nm, and an average OSNR value of the four sub-channels was used for performance evaluation. It can be seen from Fig. 4 that the required OSNR for the preFEC BER value of 10 3 in back-to-back (b2b) scenario is 17.5 dB. This value will be used as a reference baseline for performance degradation comparison for the different fiber lengths. The BER curves for zero and negative chirp modulators in b2b coincide, as expected from the theory. In addition, it can be seen that the required OSNR for the pre-FEC BER value of 10 3 over 25 km of residual dispersion, is 19 dB and 20.5 dB for negative or zero chirp modulators, respectively. Furthermore, Fig. 4 reveals that the
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proposed technique allows digital compensation for transmission over 40 km of residual dispersion using negative or zero chirp modulators, with OSNR of 19.5 dB and 22 dB, respectively, for the same pre-FEC BER value of 10 3. It should be noted that the available OSNR in typical metro and regional networks is 19–30 dB [17], dependent on network design and topology, transmission distance, transmission power etc. Thus, the proposed technique is suitable for a wide range of ASE limited and WDM applications. In this section, the reduced BW components were used in the Tx side only, while in the Rx side consists of a full BW receiver. Theoretical analysis [13] predicts additional performance degradation due to reduced BW Rx as will be shown in the next section.
4.3. Dispersion tolerance with reduced BW Tx and reduced BW Rx The (residual) dispersion tolerance of the proposed technique is shown in Fig. 5. Here, there are two main differences as compared to the setup of the previous section: (i) 10 GHz BW Rx instead of 30 GHz BW Rx and (ii) 10 GHz EML Tx instead of 10 GHz MZM Tx was used. Both modulators exhibit zero chirp. The negative residual dispersion fiber was achieved by using dispersion compensation fiber (DCF). It can be seen that the proposed MLSE-based ASIC allows using low cost reduced BW (10 G) components on both Tx and Rx sides. The residual dispersion equivalent to 7 40 km (solid and dashed lines represent positive and negative dispersion, respectively) can
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be tolerated at OSNR of 23 dB, at the pre-FEC BER value of 10 3. 750 km of SSMF (equivalent to 7800[ps/nm] of CD) requires OSNR of 24 dB at the same BER point. As the inset of Fig. 5 shows, symmetric performance is obtained for negative and positive residual dispersion (up to measurements errors). This can be explained by the fact that the MLSE receiver is affected by amount of inter-symbol interference (ISI) only, regardless of the physical origin. The performance comparison of full BW Rx (30 GHz BW) and reduced BW Rx (10 GHz BW) is summarized in Fig. 6. The observed results can be interpreted as follows. On the one hand, full BW Rx introduces less ISI, but on the other hand, it introduces more noise. The MLSE carries out the optimization between the amount of residual ISI and noise at every OSNR point. Hence, in the case where the system is “noise limited” (OSNRo19 dB for b2b, and o23 dB for 25 km), the reduced BW 10 G Rx performs slightly better. On the other hand, in the case where the system is “ISI limited” (OSNR419 dB for b2b, and 423 dB for 25 km), full BW Rx performs better, since the overall ISI in the system is less, as compared to the reduced BW Rx case. Recall that in both cases, the same 10 G Tx's were used, therefore even in back-to-back case
there is a significant ISI in the system. For 40 km SSMF, the overall ISI in the full BW Rx case is close to the memory depth limit of the implemented MLSE decoder (memory depth of 4 bits). In the reduced BW Rx case, the effect of ISI is further pronounced, resulting in uncompensated ISI (residual ISI), and the system becomes “ISI limited’. Thus, the performance in the 40 km link with the reduced BW Rx is worse than with the full BW Rx. Moreover, since higher residual ISI remains, the slope of the “waterfall” BER curve is less steep as compared to its full BW counterpart. 4.4. PMD tolerance: reduced BW TX and reduced BW Rx To measure the PMD tolerance of the proposed low cost MLSE receiver, polarization scrambler and PMD emulator were added to the setup at the AWG output of the Tx side, as shown in Fig. 7 below. The polarization scrambler speed was 470 Hz, and differential group delay (DGD) of 12 ps between the two polarizations of the transmitted optical field was generated by the PMD emulator. Zero chirp 10 GHz EMLs were used at the Tx side, and 10 GHz PIN photodiodes were used at the Rx side.
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The measured system performance for 0, 25 and 40 km is summarized in Fig. 8. Under the above conditions, negligible performance penalty is observed for all the tested link lengths with DGD of 12 ps at the pre-FEC BER region of 10 3. However, in the 40 km case, as OSNR increases, the performance penalty due to the PMD increases. This can be explained by the fact that the PMD introduces additional ISI while the system is in its “ISI limited” region, thus forming residual ISI. Recall that 40 km of SSMF in addition to reduced BW components at both Tx and Rx sides already introduce ISI which is close to the implemented memory depth of the decoder (4 symbols). As Fig. 8 reveals, when operating at the OSNR region of 26–28 dB, suitable for 40 km, the penalty due to 12 ps of DGD and scrambling speed of 470 Hz is 1 dB, as compared to the corresponding scenario without DGD (40 km, 0 ps DGD). At lower link lengths (0 km and 25 km) the penalty is even lower (less than 0.5 dB). 4.5. APD receiver: Zero chirp reduced BW TX and reduced BW APD Rx Additional advantage of the proposed technique is enabling the option to use 8 GHz avalanche photodiode (APD) receiver (designated as 10 G APD in all the relevant figures), whereas 25 GHz APD
receivers are not commercially available. APD receivers have better sensitivity, thus, from system view point, the use of APD receiver allows the saving of the pre-amplifier EDFA at the Rx side. Fig. 9 summarizes the performance comparison of 10 GHz PIN vs. 8 GHz APD receivers. Zero chirp 10 G EML TOSA was used in both cases. The optical input power at the AWG output per each of the four sub-channels was 6 dBm for PIN and 15 dBm for APD, respectively. It can be seen that in ASE limited networks, 10 GHz PIN Rx performs better than 8 GHz APD. However, it will be shown in the following Section 5.2, that 8 GHz APD outperforms in regard to sensitivity to received optical input power. Yet, when ASE noise dominates, the APD performance degradation can be explained by the fact the APD enhances the signal-spontaneous noise, and particularly the ISI-signal-spontaneous related part in the heavily induced ISI environment of the reduced bandwidth Tx and Rx. Furthermore, the 10 GHz PIN has wider BW than the APD, resulting in less residual ISI, and, in turn, improved performance. Since the observed error floor is one order of magnitude lower than the FEC threshold, there is enough margin for error free postFEC operation.
Please cite this article as: A. Gorshtein, et al., Optics Communications (2014), http://dx.doi.org/10.1016/j.optcom.2014.11.015i
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5. 4 28Gbit/sec point to point (p2p) transmission 5.1. Experimental setup The general block diagram of the experimental setup used to measure the performance of the point-to-point system is depicted on Fig. 10. This scheme is very similar to the ASE limited setup, shown on Fig. 3, and the main difference is the lack of optical amplifier. Consequently, the dominant noise source in the system is thermal noise, and the BER was measured as a function of the optical input power at the AWG output per each of the four subchannels at the receiver. 5.2. APD vs. PIN: Dispersion tolerance of reduced BW Tx and reduced BW Rx system Dispersion tolerance of the proposed technique is summarized in Fig. 11. A zero chirp 10 G EML with 10 GHz driver is used at the Tx side and a 10 GHz PIN or a 8 GHz APD photodiodes are used at the Rx side. Fig. 11 reveals that the proposed technique allows 40 km reach transmission with both 10 GHz PIN and 8 GHz APD receivers. While using 10 GHz PIN receiver, an optical input power of 15 dBm is required to achieve a pre-FEC BER value of 10−3. However, with the use of 8 GHz APD, a significant sensitivity improvement is achieved, and the required optical input power is as low as 22 dBm to achieve the same pre-FEC BER value. On the other hand, BER floor is observed at the ISI limited region (BER o10 5), possibly due to the severe bandwidth limitation introduced by the APD frequency response as well as the inherent higher shot noise at the APD. This limitation, however, still allows error-free post-FEC operation, since the BER floor is 2 orders of magnitude lower than the FEC threshold. 6. Summary and conclusion A low cost 112 G DSP-based MLSE direct-detection receiver is presented. The MLSE receiver enables the use of previous-
generation 10 G optoelectronics including the lasers, modulators, drivers, photo-detectors, and electronic RF amplifiers. Inclusive quantitative analysis of the performance of a fully implemented 100 G (4 28 Gbit/s) NRZ direct detection transceiver is presented. With the help of the DSP engines, the proposed solution allows dramatic cost reduction and improved performance of metro and point-to-point applications. The experimental results show that the proposed technique has a (residual) CD tolerance of up to 800 ps/nm, and less than 1 dB penalty due to 12 ps of PMD. At the Rx side APD can be used instead of PIN, which allows saving optical pre-amplification. In point-to-point applications, 40 km of transmission is achievable. In addition, using APD can provide additional margin as much as 7 dB in power budget. In addition, the reported receiver includes an integrated quad 5-bits ADC with sampling rate of 28G samples/s. The entire ADCþ MLSE receiver is fully implemented in standards CMOS 65 nm process.
Uncited reference [16]
References [1] M.G. Taylor, “Coherent detection method using DSP for demodulation of signal and subsequent equalization of propagation impairments,”, IEEE Photon. Technol. Lett 16 (2) (2004) 674–676. [2] E. Ip, J.M. Kahn, “Digital equalization of chromatic dispersion and polarization mode dispersion,”, J. Lightwave Technol 25 (8) (2007) 2033–2043. [3] S.J. Savory, “Digital filters for coherent optical receivers,”, Opt. Express 16 (2) (2008) 804–817. [4] S.J. Savory, “Compensation of Fibre Impairments in Digital Coherent Systems”, Brussels, , Belgium, 2008 (in ECOC 2008, 21-25 September). [5] C.R.S. Fludger, T. Duthel, D. van den Borne, C. Schulien, E.D. Schmidt, T. Wuth, J. Geyer, E. De Man, G.D. Khoe, H. de Waardt, “Coherent equalization and POLMUX-RZ-DQPSK for robust 100-GE transmission,”, J. Lightwave Technol. 26 (2008) 131–141.
Please cite this article as: A. Gorshtein, et al., Optics Communications (2014), http://dx.doi.org/10.1016/j.optcom.2014.11.015i
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[6] K. Roberts, M. O’Sullivan, K.T. Wu, H. Sun, A. Awadalla, D.J. Krause, C. Laperle, “Performance of dual-polarization QPSK for optical transport systems,”, J. Lightwave Technol. 27 (2009) 3546–3559. [7] M. Kuschnerov, F.N. Hauske, K. Piyawanno, B. Spinnler, M.S. Alfiad, A. Napoli, B. Lankl, “DSP for coherent single-carrier receivers,”, J. Lightwave Technol. 27 (16) (Aug. 2009) 3614–3622. [8] A. Gorshtein, O. Levy, G. Katz, D. Sadot, “Coherent Compensation for 100 G DPQPSK with one sample per symbol based on anti-aliasing filtering and blind equalization MLSE”, IEEE Photon. Technol. Lett 22 (16) (2010) 1208–1210. [9] P.J. Winzer, R.-J. Essiambre, “Advanced optical modulation formats,”, Proc. IEEE 94 (2006) 952–985. [10] I. Lyubomirsky, “Advanced modulation formats for ultra-dense wavelength division multiplexing,” 〈http://www.journalogy.net/Publication/5674594/ad vanced-modulation-formats-for-ultra-dense-wavelength-divisionmultiplexing〉. [11] J.-P. Elbers, H. Wernz, H. Griesser, C. Glingene, A. Faerbert, S. Langenbach, N. Stojanovic, C. Dorschky, T. Kupfer, C. Schuliea, “Measurement of the dispersion tolerance of optical duobinary with an MLSE-receiver at 10.7 Gb/s,” Proc. OFC 2005, OThJ4.
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67 [12] J.D. Downie and J. Hurle, “Chromatic dispersion compensation effectiveness of an MLSE-EDC receiver for three variants of duobinary,” IEEE/LEOS Summer 68 Topical Meetings, TuA1.2, 29-30 (2007). 69 [13] A. Gorshtein, O. Levy, G. Katz, D. Sadot, “ Low cost 112G direct detection metro 70 transmission system with reduced bandwidth (10G) components and MLSE compensation,” SPPCoM 2011, SPWC4. 71 [14] A. Gorshtein, D. Sadot, Advanced Modulation Formats and MLSE Based Digital 72 Signal Processing for 100Gbit/sec Communication Through Optical Fibers, 73 Optical Fiber Communications and Devices, in: Dr Moh Yasin (Ed.), InTech, ISBN 978-953-307-954-7http://dx.doi.org/10.5772/27062 (Available from). 74 [15] G. Agrawal, Fiber Optic Communications Systems, John Wiley & Sons, Inc., Q3 75 2002. 76 [16] A. Gorshtein, O. Levy, G. Katz, D. Sadot, “Blind channel estimation for MLSE receiver in high speed optical communications: theory and ASIC im77 plementation,”, Opt. Express 21 (2013) 21766–21789. 78 [17] M.H. Eiselt and B.T. Teipen, “ Requirements for 100-Gb/s Metro Networks,” 79 Proc. OFC 2009, OTuN6.
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Contents lists available at ScienceDirect
Optics Communications journal homepage: www.elsevier.com/locate/optcom
Low cost 112 G direct detection metro transmission system with reduced bandwidth (10 G) components and MLSE compensation Albert Gorshtein a,n, Dan Sadot a, Nir Sheffi b, Eduard Sonkin b, Yuval Shachaf b, Don Becker b, Omri Levy b, Gilad Katz b a b
Electrical and Computer Engineering Department, Ben Gurion University of the Negev, Beer Sheva 84105, Israel MultiPhy Ltd., Ness Ziona, Israel
art ic l e i nf o
a b s t r a c t
Article history: Received 18 October 2014 Accepted 5 November 2014
We experimentally demonstrate a 100 G ADC-MLSE-based 4-lanes NRZ direct detection transceiver. The DSP engine enables the use of low cost 10 G optoelectronics and supports 40 km metro transmission suitable for both ASE limited and point-to-point applications. The fully implemented mixed signal ASIC is based on standard 65 nm CMOS process. & 2014 Published by Elsevier B.V.
Keywords: Intensity modulation direct detection Digital signal processing Maximum likelihood sequence estimation (MLSE)
1. Introduction The constant growth in the demand for high bandwidth data transmission leads to higher challenges that should be resolved in the physical layer, and particularly by optical transmission technology. The current high end transmission data rates are in the range of hundreds of Gbits/s. One emerging technology that can support such bitrates for long distances (hundreds of kilometers and above) is coherent transmission and detection [1–8]. On the other hand, direct detection (DD) technology offers the use of lower cost optoelectronic components (O/E), consumes less power and enables overall lower latency solution. These advantages may be critical for metro and short reach applications such as sub-hundred kilometers networks of metro-edge and data centers interconnections. However, direct detection (non-coherent) optical technology is limited to lower bit rates and/or shorter distances. For example, increasing the bitrate from 10 Gbit/s to 25 Gbit/s, results in distance reduction from ∼80 km to ∼15 km, for the same bit error rate (BER) performance. The main reason for this limitation is the inter-symbol interference (ISI) caused by chromatic dispersion (CD) and polarization mode dispersion (PMD). To combat this ISI two techniques are commonly used. The first technique is based on advanced modulation formats, together with partial response signaling [9,10], while the other approach is based on digital signal n
Corresponding author. . E-mail address: [email protected] (A. Gorshtein).
processing (DSP), applying electronic dispersion compensation (EDC). The EDC implementations with maximum likelihood sequence estimation (MLSE) at the receiver (Rx) side, is theoretically the optimal tool to combat ISI, and was very popular for 10 Gbis/s [11,12]. The combination of the two approaches is also possible, and was also theoretically investigated for 4 25 Gbits transmission with the use of reduced bandwidth (BW) components [13]. A more detailed theoretical background on reduced BW components transmission with subsequent MLSE compensation of transmission impairments can be found in [14]. In addition, it should be noted, that in case that nonlinearities occur, the MLSE is capable of coping with self-phase modulation effect, provided that the memory depth of the MLSE is long enough. Here, an inclusive experimental investigation of the 100 Gbit/s transceiver as proposed in [13] is presented and analyzed. All the presented results are based on recorded measurements by means of real time application specific integrated circuit (ASIC). It is shown that the MLSE-based system outperforms its conventional DD counterparts by (a) allowing use of lower cost components, (b) supporting longer reach transmission distances, (c) yielding improved BER results, and (d) doubling the spectral efficiency due to the use of reduced bandwidth components. In particular, transmission over 40 km of a standard single mode fiber (SSMF) is achieved with non-return-to-zero (NRZ) on-off keying (OOK) transmission, supporting an optical signal to noise ratio (OSNR) of 22 dB for a pre-FEC BER of 10 3. The rest of the paper is organized as follows. Section 2 introduces the noise-ISI tradeoff in the MLSE histograms domain. Section 3 describes the high level ASIC architecture and its design
http://dx.doi.org/10.1016/j.optcom.2014.11.015 0030-4018/& 2014 Published by Elsevier B.V.
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parameters. Section 4 presents the conducted set of real time experiments via amplified spontaneous emission (ASE) limited transmission system. The setup and real time experimental results of the point-to-point system without optical amplification is given in Section 5. Finally, Section 6 presents conclusive remarks.
2. Noise–ISI tradeoff analysis through the histograms domain Assuming that the optical transmission power is low enough, such that fiber Kerr non-linarites can be neglected, there are two main degradation effects on signal quality in DD systems: ISI and noise. The ISI is generated by CD, PMD, and optical and electrical components BW limitations. The noise, in turn, stems either from optical or electrical amplifiers present in the system, or from the opto-electronic conversion. In ASE limited systems the dominant noise is generated by optical amplifiers [15]. In point-to-point links without optical amplification, however, the dominant noise mechanisms are thermal noise and shot noise [15]. Both phenomena, ISI and noise, can be conveniently represented in the histograms domain as will be illustrated shortly below. The histograms represent the (un-normalized) conditional probability density functions (PDFs), of the received samples rn ,
10GHz Tx, 10GHz Rx,<σ2i > = 1.50, SNRest = 17.6dB
given the transmitted sequence of Nisi consecutive symbols [an , an − 1, ... , an − Nisi+ 1]. The histograms are used by the MLSE decoder in order to obtain the estimated sequence with highest probability that has been transmitted as described in [15]. The Nisi parameter represents the memory depth of the MLSE decoder, and should be chosen such that it exceeds the memory depth of the overall channel (transmitter þfiber þreceiver). Fig. 1(a) and (b) shows examples of histogram sets received in a 28 Gbit/s lane, after a transmission over 25 km amplified optical link with reduced BW (10 GHz) components at the Tx, and 10 GHz vs. 30 GHz components at the Rx side respectively. OSNR of 21 dB and input power of 5 dBm were used in both cases. Each histogram set contains 32 conditional histograms, termed “branches”, corresponding toNisi = 4 . Although the presented histogram shapes do not strictly follow the Gaussian distribution, they are treated as such for the illustration purposes only, to help provide more insight on the noise–ISI tradeoff. As Fig. 1(a) and (b) reveals, there are more than two levels in the received signal, accounting for the amount of ISI present in the system. In case there was no noise in the system, the branches mean values would represent the (non-linear) impulse repose of the channel. However, since the dominant noise source in the presented system configuration is ASE, each branch has a variance, which depends on its mean value.
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It is shown in Fig. 1(c) and (d) that branches with higher mean value also have higher variance (up to a clipping effect introduced by the analog-to-digital conversion). In the ‘noise-free’ scenario, the MLSE is capable of fully compensating for the ISI, provided that the length of the overall channel (Tx þfiberþ Rx) impulse response is less (or equal) than the decoder memory. However, due the presence of signal-dependent noise, the ISI is not fully compensated. This results in performance degradation, and subsequent OSNR (and/or power) penalty. It should be noted, that the MLSE is carrying out the optimal decisions, in the maximum-likelihood sense, at each OSNR (or power) point, selecting the best “compromise” between ISI and noise in terms of optimal bit error rate (BER). To quantify the effect of the noise–ISI tradeoff, the following estimate of the signal-to-noise ratio (SNR) at the MLSE input is proposed:
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where μi and σi2 represent the mean and variance values of each branch, and Nbr = 2Nisi+ 1, is the overall number of branches as proposed in [13]. Comparing the branch histograms in Fig. 1(c) and (d) reveals that the histogram means in both cases are very similar, whereas the variances in case of 1(c) are significantly smaller than in 1(d), while the only difference between the two cases is the Rx bandwidth. It should be noted that in both cases, the signals were sampled by the same analog to digital converter (ADC), with an anti-aliasing filter bandwidth of 14 GHz. The corresponding average noise variances are 1.5 and 2.3 for the 10 GHz and 30 GHz Rx respectively. The corresponding SNRs are 17.6 dB in the reduced BW Rx case vs. 15.8 dB in the full BW Rx case, indicating that the latter will have poorer performance. On the other hand, when OSNR is increased, the system becomes less noise-limited and more ISI limited. In such case, the average noise variance in the full BW (30 GHz) case becomes comparable with the “difference in the residual (uncompensated) ISI” of 10 GHz Rx vs. the 30 GHz Rx cases. Consequently, the BER of the wider bandwidth Rx system will outperform the lower bandwidth Rx case. This result is observed in measurements depicted in Fig. 6. In conclusion, the noise–ISI tradeoff can be applied to optimize the system performance by using reduced BW components that are optimized for a given amount of ISI, MLSE decoder memory depth, and SNR conditions. In the following sections, various systems optimizations are investigated experimentally for a wide range of channel conditions.
3. ASIC architecture and parameters High level architecture of the MP1100Q ASIC and a zoom into its lanes structure is presented in Fig. 2. The MP1100Q DEMUX consists of four 4 28 Gsymbols/s lanes with MLSE based engines. Each MLSE engine follows an analog front end, that contains a digitally-controlled self-calibrated analog to digital converter (ADC) that samples at 28 Gsamples/s, with 5 bit resolution (effective number of bits (ENOB) ∼3.8). In addition, the analog front end contains a digitally-controlled analog clock recovery (CR) and digitally-controlled variable gain amplifier (VGA). The MLSE engine is a 16-state fully implemented Viterbi processor, with an overall latency of under 200 ns. The initialization of the MLSE is fully automatic and follows a unique blind channel acquisition algorithm [15]. In addition, the MLSE is fully adaptive, and is capable of tracking different channel dynamics, in the order of 1 ms. In turn, the 4 28G data stream
coming out of the four lanes is collected to a 4-10 gearbox, that redistributes it into 10 Gbit/s output lanes designated in Fig. 2 as 10 G Tx, according to either IEEE802.3ba CAUI or ITU-T OTU4.10 protocols. Furthermore, MP1100Q has a built-in fully configurable pseudo random bit sequence (PRBS) generator and checker at the output of each 28 G lane that can be used for bit error rate (BER) testing. The ASIC is implemented using 65 nm CMOS technology and has about 15.2 million gates.
4. 4 28Gbit/s transmission in ASE limited network 4.1. Experimental setup The general block diagram of the ASE limited setup, used in various experiments is shown in Fig. 3. The transmit side (Tx) incudes 4 28 Gbit/s signal generator that feeds four off-the-shelf modulator drivers (Mod. drivers), followed by four linear Transmit Optical Sub-Assemblies (TOSAs), which perform the electrical-tooptical conversion. The four optical signals are multiplexed and demultiplexed through a 50 GHz arrayed waveguide gratings (AWG) with 5 dB insertion loss, at transmit and receive sides, respectively. The wavelength division multiplexed (WDM) signal propagates through an SSMF (the length may vary for different experiments). The optical fiber is used to simulate the amount of residual dispersion in metro and regional links. Depending on the network topology and design, the residual dispersion can be either positive or negative. In addition, some networks use symmetric dispersion widows, while others use asymmetric windows. Hence in some experiments, a dispersion compensation fiber (DCF) with negative dispersion coefficient was used. The optical intensity at the fiber output is amplified by erbium doped fiber amplifier (EDFA) and coupled together with an amplified spontaneous emission (ASE) noise source. Variable optical attenuators (VOAs) are used to control the received input power and OSNR value. The OSNR was measured by tapping 5% of the signal into an optical spectrum analyzer (OSA). The receiver (Rx) includes four off-theshelf Receive Optical Sub-Assemblies (ROSAs), which perform the optical-to-electrical conversion, and, in turn, are followed by the MP1100Q MLSE receiver ASIC. As described in the previous section, the MP100Q receives the heavily distorted incoming signal (by optical channel and reduced BW components), recovers the clock, digitizes the signal via the 4 28 Gsamples/s ADCs and recovers the data by the DSP postprocessing engine based on the four MLSE decoders. In the sequel, different scenarios were tested, where the BW of the TOSAs and ROSAs are scenario-dependent. In all the following
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scenarios, a full symmetry was maintained between the 4-lanse components, both at the Tx and Rx side, i.e., four TOSAs and four ROSAs from the same series were used. The same is true also for the modulator drivers. The difference between the scenarios is the Tx and/or Rx bandwidth, the modulator types, (Mach Zehnder Modulator (MZM) vs. electro-absorption modulator laser (EML)), the modulator chirp parameter (zero vs. negative chirp) and the transmission distance (back-to-back vs. 25 km, vs. 40 km). Hence, the block diagram in Fig. 3 is general and is used to described all the tested scenarios all together. In all the experiments, the optical output power of each TOSA was tuned to be 0 dBm, resulting in 1 dBm of the four subchannels at the AWG output. This optical power was set in order to avoid fiber Kerr nonlinearities. In addition, a pseudo-random bit sequence (PRBS) of length 223 − 1 was used, and the PRBS checkers inside MP1100Q was configured accordingly. The average bit error rate (BER) over the four ASIC lanes was measured. The aggregated transmission rate of 112 Gbit/s (4 28 G) was used in all the experiments, in order to include the forward error correction (FEC) overhead.
4.2. The effect of modulator chirp and residual dispersion, with reduced BW Tx, and full BW Rx In this scenario, reduced BW components were used only in the Tx side. The modulator drivers have 12 GHz BW, while 10 GHz MZMs with zero and negative chirp were examined. At the Rx side full BW (30 GHz) receivers were used. The fiber lengths of 0 km (back to back), 25 km and 40 km were tested. The BER performance vs. OSNR for different fiber lengths and different modulator chirp are presented in Fig. 4. The OSNR is measured per channel according to the formal definition using reference BW of 0.1 nm, and an average OSNR value of the four sub-channels was used for performance evaluation. It can be seen from Fig. 4 that the required OSNR for the preFEC BER value of 10 3 in back-to-back (b2b) scenario is 17.5 dB. This value will be used as a reference baseline for performance degradation comparison for the different fiber lengths. The BER curves for zero and negative chirp modulators in b2b coincide, as expected from the theory. In addition, it can be seen that the required OSNR for the pre-FEC BER value of 10 3 over 25 km of residual dispersion, is 19 dB and 20.5 dB for negative or zero chirp modulators, respectively. Furthermore, Fig. 4 reveals that the
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Please cite this article as: A. Gorshtein, et al., Optics Communications (2014), http://dx.doi.org/10.1016/j.optcom.2014.11.015i
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proposed technique allows digital compensation for transmission over 40 km of residual dispersion using negative or zero chirp modulators, with OSNR of 19.5 dB and 22 dB, respectively, for the same pre-FEC BER value of 10 3. It should be noted that the available OSNR in typical metro and regional networks is 19–30 dB [17], dependent on network design and topology, transmission distance, transmission power etc. Thus, the proposed technique is suitable for a wide range of ASE limited and WDM applications. In this section, the reduced BW components were used in the Tx side only, while in the Rx side consists of a full BW receiver. Theoretical analysis [13] predicts additional performance degradation due to reduced BW Rx as will be shown in the next section.
4.3. Dispersion tolerance with reduced BW Tx and reduced BW Rx The (residual) dispersion tolerance of the proposed technique is shown in Fig. 5. Here, there are two main differences as compared to the setup of the previous section: (i) 10 GHz BW Rx instead of 30 GHz BW Rx and (ii) 10 GHz EML Tx instead of 10 GHz MZM Tx was used. Both modulators exhibit zero chirp. The negative residual dispersion fiber was achieved by using dispersion compensation fiber (DCF). It can be seen that the proposed MLSE-based ASIC allows using low cost reduced BW (10 G) components on both Tx and Rx sides. The residual dispersion equivalent to 7 40 km (solid and dashed lines represent positive and negative dispersion, respectively) can
Please cite this article as: A. Gorshtein, et al., Optics Communications (2014), http://dx.doi.org/10.1016/j.optcom.2014.11.015i
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be tolerated at OSNR of 23 dB, at the pre-FEC BER value of 10 3. 750 km of SSMF (equivalent to 7800[ps/nm] of CD) requires OSNR of 24 dB at the same BER point. As the inset of Fig. 5 shows, symmetric performance is obtained for negative and positive residual dispersion (up to measurements errors). This can be explained by the fact that the MLSE receiver is affected by amount of inter-symbol interference (ISI) only, regardless of the physical origin. The performance comparison of full BW Rx (30 GHz BW) and reduced BW Rx (10 GHz BW) is summarized in Fig. 6. The observed results can be interpreted as follows. On the one hand, full BW Rx introduces less ISI, but on the other hand, it introduces more noise. The MLSE carries out the optimization between the amount of residual ISI and noise at every OSNR point. Hence, in the case where the system is “noise limited” (OSNRo19 dB for b2b, and o23 dB for 25 km), the reduced BW 10 G Rx performs slightly better. On the other hand, in the case where the system is “ISI limited” (OSNR419 dB for b2b, and 423 dB for 25 km), full BW Rx performs better, since the overall ISI in the system is less, as compared to the reduced BW Rx case. Recall that in both cases, the same 10 G Tx's were used, therefore even in back-to-back case
there is a significant ISI in the system. For 40 km SSMF, the overall ISI in the full BW Rx case is close to the memory depth limit of the implemented MLSE decoder (memory depth of 4 bits). In the reduced BW Rx case, the effect of ISI is further pronounced, resulting in uncompensated ISI (residual ISI), and the system becomes “ISI limited’. Thus, the performance in the 40 km link with the reduced BW Rx is worse than with the full BW Rx. Moreover, since higher residual ISI remains, the slope of the “waterfall” BER curve is less steep as compared to its full BW counterpart. 4.4. PMD tolerance: reduced BW TX and reduced BW Rx To measure the PMD tolerance of the proposed low cost MLSE receiver, polarization scrambler and PMD emulator were added to the setup at the AWG output of the Tx side, as shown in Fig. 7 below. The polarization scrambler speed was 470 Hz, and differential group delay (DGD) of 12 ps between the two polarizations of the transmitted optical field was generated by the PMD emulator. Zero chirp 10 GHz EMLs were used at the Tx side, and 10 GHz PIN photodiodes were used at the Rx side.
Please cite this article as: A. Gorshtein, et al., Optics Communications (2014), http://dx.doi.org/10.1016/j.optcom.2014.11.015i
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The measured system performance for 0, 25 and 40 km is summarized in Fig. 8. Under the above conditions, negligible performance penalty is observed for all the tested link lengths with DGD of 12 ps at the pre-FEC BER region of 10 3. However, in the 40 km case, as OSNR increases, the performance penalty due to the PMD increases. This can be explained by the fact that the PMD introduces additional ISI while the system is in its “ISI limited” region, thus forming residual ISI. Recall that 40 km of SSMF in addition to reduced BW components at both Tx and Rx sides already introduce ISI which is close to the implemented memory depth of the decoder (4 symbols). As Fig. 8 reveals, when operating at the OSNR region of 26–28 dB, suitable for 40 km, the penalty due to 12 ps of DGD and scrambling speed of 470 Hz is 1 dB, as compared to the corresponding scenario without DGD (40 km, 0 ps DGD). At lower link lengths (0 km and 25 km) the penalty is even lower (less than 0.5 dB). 4.5. APD receiver: Zero chirp reduced BW TX and reduced BW APD Rx Additional advantage of the proposed technique is enabling the option to use 8 GHz avalanche photodiode (APD) receiver (designated as 10 G APD in all the relevant figures), whereas 25 GHz APD
receivers are not commercially available. APD receivers have better sensitivity, thus, from system view point, the use of APD receiver allows the saving of the pre-amplifier EDFA at the Rx side. Fig. 9 summarizes the performance comparison of 10 GHz PIN vs. 8 GHz APD receivers. Zero chirp 10 G EML TOSA was used in both cases. The optical input power at the AWG output per each of the four sub-channels was 6 dBm for PIN and 15 dBm for APD, respectively. It can be seen that in ASE limited networks, 10 GHz PIN Rx performs better than 8 GHz APD. However, it will be shown in the following Section 5.2, that 8 GHz APD outperforms in regard to sensitivity to received optical input power. Yet, when ASE noise dominates, the APD performance degradation can be explained by the fact the APD enhances the signal-spontaneous noise, and particularly the ISI-signal-spontaneous related part in the heavily induced ISI environment of the reduced bandwidth Tx and Rx. Furthermore, the 10 GHz PIN has wider BW than the APD, resulting in less residual ISI, and, in turn, improved performance. Since the observed error floor is one order of magnitude lower than the FEC threshold, there is enough margin for error free postFEC operation.
Please cite this article as: A. Gorshtein, et al., Optics Communications (2014), http://dx.doi.org/10.1016/j.optcom.2014.11.015i
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5. 4 28Gbit/sec point to point (p2p) transmission 5.1. Experimental setup The general block diagram of the experimental setup used to measure the performance of the point-to-point system is depicted on Fig. 10. This scheme is very similar to the ASE limited setup, shown on Fig. 3, and the main difference is the lack of optical amplifier. Consequently, the dominant noise source in the system is thermal noise, and the BER was measured as a function of the optical input power at the AWG output per each of the four subchannels at the receiver. 5.2. APD vs. PIN: Dispersion tolerance of reduced BW Tx and reduced BW Rx system Dispersion tolerance of the proposed technique is summarized in Fig. 11. A zero chirp 10 G EML with 10 GHz driver is used at the Tx side and a 10 GHz PIN or a 8 GHz APD photodiodes are used at the Rx side. Fig. 11 reveals that the proposed technique allows 40 km reach transmission with both 10 GHz PIN and 8 GHz APD receivers. While using 10 GHz PIN receiver, an optical input power of 15 dBm is required to achieve a pre-FEC BER value of 10−3. However, with the use of 8 GHz APD, a significant sensitivity improvement is achieved, and the required optical input power is as low as 22 dBm to achieve the same pre-FEC BER value. On the other hand, BER floor is observed at the ISI limited region (BER o10 5), possibly due to the severe bandwidth limitation introduced by the APD frequency response as well as the inherent higher shot noise at the APD. This limitation, however, still allows error-free post-FEC operation, since the BER floor is 2 orders of magnitude lower than the FEC threshold. 6. Summary and conclusion A low cost 112 G DSP-based MLSE direct-detection receiver is presented. The MLSE receiver enables the use of previous-
generation 10 G optoelectronics including the lasers, modulators, drivers, photo-detectors, and electronic RF amplifiers. Inclusive quantitative analysis of the performance of a fully implemented 100 G (4 28 Gbit/s) NRZ direct detection transceiver is presented. With the help of the DSP engines, the proposed solution allows dramatic cost reduction and improved performance of metro and point-to-point applications. The experimental results show that the proposed technique has a (residual) CD tolerance of up to 800 ps/nm, and less than 1 dB penalty due to 12 ps of PMD. At the Rx side APD can be used instead of PIN, which allows saving optical pre-amplification. In point-to-point applications, 40 km of transmission is achievable. In addition, using APD can provide additional margin as much as 7 dB in power budget. In addition, the reported receiver includes an integrated quad 5-bits ADC with sampling rate of 28G samples/s. The entire ADCþ MLSE receiver is fully implemented in standards CMOS 65 nm process.
Uncited reference [16]
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