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Wideband slot-coupled dielectric resonator-based filter
Journal of Alloys and Compounds 785 (2019) 1264e1269
Contents lists available at ScienceDirect
Journal of Alloys and Compounds journal homepage: http://www.elsevier.com/locate/jalcom
Wideband slot-coupled dielectric resonator-based filter Ke Bi a, Xuying Wang a, Yanan Hao a, Ming Lei a, *, Guoyan Dong b, **, Ji Zhou c, *** a State Key Laboratory of Information Photonics and Optical Communications, School of Science, Beijing University of Posts and Telecommunications, Beijing 100876, China b College of Materials Science and Opto-Electronic Technology, University of Chinese Academy of Sciences, Beijing 100049, China c State Key Laboratory of New Ceramics and Fine Processing, School of Materials Science and Engineering, Tsinghua University, Beijing 100084, China
a r t i c l e i n f o
a b s t r a c t
Article history: Received 28 November 2018 Accepted 24 January 2019 Available online 25 January 2019
A simple and novel slot-coupled dielectric resonator-based filter (DRF) with wideband performance is proposed, and the designed geometric parameters of the filter are discussed in detail. The proposed filter comprises a substrate, a microstrip line, and three dielectric resonators (DRs), which are positioned on the center of the corresponding slots to obtain a wide stopband response. Numerical simulations show that the wideband DRF has a center frequency of 6.5 GHz, which is consistent with the experimental measurement results. The simulation, as well as the experimental results demonstrate that the distance between the adjacent DRs is a key parameter that can be used to tune the bandwidth of the wideband DMF. The physical mechanism of the DRs is also revealed through analysis of the electromagnetic field distribution results. Additionally, the influence of the DRs on the stopband frequency is also discussed. The results indicate that the DRs play a critical role in the proposed filter. Furthermore, this paper presents the first report on a wideband DRF. © 2019 Elsevier B.V. All rights reserved.
Keywords: Wideband Dielectric resonator filter Band-stop filter Microwave filter
1. Introduction With the rapid development of mobile wireless communication systems, microwave filters have been progressively attracting an increasing amount of attention [1,2]. Both band-pass and band-stop filters play the role of important selective signal devices in various wireless communication systems [3,4]. Band-pass filters enable the transmission of desired signals. On the contrary, band-stop filters suppress the undesired signals in various applications [5,6]. Recent advances in microwave transmission systems have increased the demand for band-stop filters with good frequency response, such as filters with a wideband rejection feature, or those with excellent in-band performance [7,8]. Traditional band-stop filters include an electronic bandgap and defected ground structures such as double spiral, which place restrictions on miniaturization [9,10]. Substrateintegrated waveguide structures are also implemented in bandstop filters, which is typically integrated into microstrip or coplanar waveguide, and also other planar circuits [11]. However, it
* Corresponding author. ** Corresponding author. *** Corresponding author. E-mail addresses: [email protected] (M. Lei), [email protected] (G. Dong), [email protected] (J. Zhou). https://doi.org/10.1016/j.jallcom.2019.01.286 0925-8388/© 2019 Elsevier B.V. All rights reserved.
is difficult to manufacture band-stop filters on a large scale because of their complex structures [12]. Dielectric resonator (DR) is widely employed in wireless communications system due to its unique characteristic, such as highquality factor [13]. It is known that the propagation of electromagnetic waves through the substrate can be regulated by simply controlling the interaction between electromagnetic waves and dielectric materials [14e16]. Recently, dielectric resonators (DRs) have garnered a considerable amount of interest owing to their ability to be easily integrated into communication systems [17,18]. The compact size and low production cost of dielectric resonatorbased filters (DRFs) make them a promising substitute for traditional filters in wireless communication systems [19]. Despite these desirable characteristics, most DRFs are reported to possess narrow bandwidth properties, which are not advantageous in practical applications [20,21]. Recently, we have designed a dielectric resonator filter with thermally tunable characteristic [22]. But, this filter still exhibits a narrow band property. In this work, a novel wideband slot-coupled dielectric resonator-based band-stop filter is developed. In the proposed DRF design, three DRs are respectively mounted onto three slots. This design is based on the idea that the stopband bandwidth can be varied by adjusting the distance between adjacent DRs, while also minimizing deviation from the 6.5-GHz center frequency of the
K. Bi et al. / Journal of Alloys and Compounds 785 (2019) 1264e1269
stopband. The obtained numerical simulation results are found to agree well with the corresponding experimental measurement results. 2. Prototype filter design Fig. 1 draws the 3-dimensional geometry of the single slotcoupled DMF. On the ground plane, a DR with side length dimensions of a ¼ 2 mm and c ¼ 3.8 mm, and dielectric constant ε ¼ 35, is fixed above the substrate, at the center of the slot, as is shown in Fig. 1(a). Corresponding to the top view in Fig. 1(b), the
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F4B substrate has a dimension of L ¼ 40 mm, W ¼ 20 mm, and thickness t ¼ 1.0 mm. Its relative permittivity and loss tangent are εr ¼ 2.65 and tand ¼ 0.002, respectively. A slot with length b and width d is etched into the center of the substrate. A microstrip line with dimensions of L w ¼ 40 2.7 mm, a thickness of 0.035 mm, and a resistance of 50 U is printed onto the F4B substrate, as is shown in Fig. 1(c). CST Microwave Studio 2011 based on the finite integration technique (FIT) was used to numerically simulate the varying geometric conditions of the proposed filter. Fig. 2(a) illustrates the effects of varying the slot length b on the simulation results of the
Fig. 1. Schematic diagrams for the single slot-coupled dielectric resonator-based filter. (a) 3D view, (b) top view, and (c) bottom view.
Fig. 2. The simulation results of transmission spectra for the single slot-coupled dielectric resonator-based filter with different structure parameters of (a) b, (b) d, (c) L, and (d) w.
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transmission spectra for the single slot-coupled DRF. In this simulation, the parameters of slot width, microstrip line length, and width were set as d ¼ 1.6, L ¼ 40, and w ¼ 2.7 mm, respectively. From Fig. 2(a), it is obvious that the increase of slot length b reduces the filter resonance frequency, whereas the rejection bandwidth is not significantly changed. The effects of varying the slot width d on the simulated transmission spectra are illustrated in Fig. 2(b); the relationship between the slot width d and resonance frequency can be clearly observed. In this simulation, the parameters of slot length, microstrip line length, and width were set as b ¼ 8.0, L ¼ 40, and w ¼ 2.7 mm, respectively. When increasing the slot width d from 2.0 to 3.2 mm, the resonance frequency and rejection bandwidth increase. Fig. 2(c) illustrates the effects of varying the microstrip line length L on the simulation results of the transmission spectra for the single slot-coupled DRF. In this simulation, the microstrip line width, slot length, and width were set as w ¼ 2.7, b ¼ 8.0, and d ¼ 1.6 mm, respectively. The increase of the microstrip line length L can be observed to have negligible influence on the resonance frequency, whereas the insertion loss exhibited an initial increase and subsequent decrease. The effects of the microstrip line width w variation on the simulated transmission spectra of the single slot-coupled DRF are illustrated in Fig. 2(d). In this simulation, the parameters of microstrip line length, slot length, and width were set as L ¼ 40, b ¼ 8.0, and d ¼ 1.6 mm, respectively. We can see that both the resonance frequency and insertion loss slightly increase in response to the microstrip line width w being increased. It is well known that the energy attenuation can be increased by the decrease of the insertion loss. On account of the results of the aforementioned numerical simulations, the optimized geometric parameters b, d, L, and w of the single slot-coupled DMF were chosen as 8.0, 1.6, 40, and 2.7 mm, respectively. Fig. 3 gives the schematic diagrams of the configuration and geometry for the designed wideband slot-coupled DRF. The slot and DR structures shown in Fig. 1(a) were duplicated and integrated into the original structure, as is illustrated in Fig. 3(a). Furthermore, Fig. 3(b) diagrammatizes the added parameter g, which is the distance between two adjacent DRs. The dimensions of the substrate, slot, and microstrip are the same as those illustrated in Fig. 1(b) and (c). To excite the electromagnetic resonance of each DR, three DRs are respectively mounted onto three slots.
3. Simulation and measurement results In order to verify the excellent performance of the designed DRF, a filter with a center frequency of 7.4 GHz was fabricated (Fig. 4). The proposed filter was fabricated on a printed circuit board (PCB) [23], which was then placed in a metal cavity. We have fabricated
Fig. 4. Photograph of the fabricated wideband slot-coupled dielectric resonator-based filter.
Fig. 3. Schematic diagrams of (a) 3D, (b) top, and (c) bottom views for the wideband slot-coupled dielectric resonator-based filter.
K. Bi et al. / Journal of Alloys and Compounds 785 (2019) 1264e1269
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four filters with different g values, such as g ¼ 4 mm, g ¼ 6 mm, g ¼ 8 mm, and g ¼ 10 mm. A diamond wire cutting machine was used to cut the ceramic to dimension of 2 mm 2 mm 3.8 mm for the proposed DR [24]. As can be seen, three DRs are positioned on the slot in turn. Note that all DRs are identical, made of BaZn0.33Nb0.67O3-Ba3CoNb2O9 (0.5BZN-0.5BCN, BCZN) ceramic with high quality factor Q f and low sintering temperature, and have a relative permittivity of 35 and a tangent loss of 0.002. The measured X-ray diffraction (XRD) pattern and dielectric constant of the BCZN ceramic are shown in Fig. 5. From Fig. 5(a), it can be seen that all peaks can be attributed to BCZN, and no additional peaks for impurity or intermediate phases are found. Fig. 5(b) displays that the dielectric constant of the BCZN ceramic is nearly constant within the frequency of 1 kHze10 MHz. Fig. 6 depicts the simulated and measured transmission spectra of the proposed DRF, which are in good agreement. The simulated transmission spectra are shown in Fig. 6(a) under the condition of an approximately 6.5 GHz center frequency. It can be observed that the bandwidth decreases as the distance g increases, and that the filter has an insertion loss of more than 19 dB in the wideband rejection region. Alternatively, experimental measurements were
Fig. 6. Effects of varying the distance g between the neighboring DRs on the (a) simulated and (b) measured transmission spectra for the filter.
Fig. 5. Measured XRD pattern and (b) dielectric constant of the BCZN ceramic.
carried out by an E5063A vector network analyzer to directly measure the two-port S-parameters. We have measured four filters with different g values, such as g ¼ 4 mm, g ¼ 6 mm, g ¼ 8 mm, and g ¼ 10 mm. In Fig. 6(b), the measured transmission spectra are centered at 7.4 GHz; moreover, under these conditions, the insertion loss in the wideband stopband is higher than 18 dB. Furthermore, the bandwidth can be observed to decrease as the distance g is increased from 4 to 10 mm. Considering the above analysis, all of the DRs are believed to be operating as based on the same principle: the coupling of the three slot-DR pairs results in wideband characteristics. Moreover, it can be seen that the mutual coupling effect decreases as the distance g increases. Table 1 presents a detailed comparison of the simulated and measured results for the proposed wideband DRF, and confirms the broadband feature of the filter. Both the simulation and the measured values indicate that the electromagnetic energy in the designed wideband DRF sharply attenuates within a specific frequency range. And due to the manufacturing tolerance, the simulated and measured results have minor differences. In the designed filter, the physical mechanism of the DRs is
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Table 1 Comparison of simulated and measured results for the proposed DRF. g (mm)
Sim.10-dB FBW
Mea. 10-dB FBW
Sim. IL (dB)
Mea. IL (dB)
4 6 8 10
34.8% 31.1% 28.0% 26.2%
46.5% 44.6% 39.9% 39.5%
19 25 28 33
18 19 21 20
FBW: Fractional Band-Width, IL: Insertion Loss.
revealed by the subsequent investigation into the electric and magnetic field distributions at 6.5 GHz. Fig. 7(a) shows the simulation result of the electric field distribution of the proposed wideband DRF under the conditions of g ¼ 6 mm at 6.5 GHz. In the DR, the displacement current is horizontal, i.e., y-directional. Fig. 7(b) shows the simulated magnetic field distribution in the DR, which is clearly observed to be strong, and exhibits a circular pattern. Particularly, in the DR, the generated strong displacement current as a result of the electric field of the electromagnetic waves caused a surrounding circular magnetic field; this phenomenon is considered to be electric resonance [25]. It should be noted that the strong magnetic field intensity is still observed above each slot, which is caused by the electromagnetic waves oscillation between the microstrip line and the ground plane along the z-direction. The influence of the DR on the proposed wideband DRF were also studied. Fig. 8 displays the simulated electric energy density distributions in the proposed filter under the conditions of g ¼ 10 mm and varying frequencies. Fig. 8(a) shows the three DRs coupled to their corresponding slots and generating electromagnetic waves that propagate through the substrate from one port to the other. Fig. 8(b) shows only the first DR to be coupled to the slot, demonstrating that the electromagnetic energy is mostly confined within the first DR; this means that electromagnetic waves cannot transmit through the substrate, and that the first DR significantly influences the stopband frequency of 6.5 GHz. Fig. 8(c) shows the first two DRs coupled to their corresponding slots, demonstrating that the electromagnetic energy is basically confined within the first two DRs; this indicates that the first two DRs are responsible for the stopband frequency of 7.0 GHz. Fig. 8(d) shows all three of the DRs coupled to their corresponding slots; the propagation of
Fig. 8. Simulated electric energy density distributions of the wideband DRF with g ¼ 10 mm at (a) 4.0 GHz, (b) 6.5 GHz, (c) 7.0 GHz, and (d) 9.0 GHz.
electromagnetic waves through the substrate demonstrates good agreement with the results illustrated in Fig. 6. Considering these results, it can be concluded that the stopband frequency is significantly dependent on the mutual coupling effect between the slotDM pairs. 4. Conclusions A novel wideband slot-coupled DRF has been fabricated and evaluated in this study. The consistency between the simulation and the measurement confirm that the proposed filter has the ability to suppress undesirable wideband signals. Moreover, the electric and magnetic field distributions provide insight into the electric resonance mechanism of the DMs. The electric energy density distributions further demonstrate the important role of DRs in the filter. The proposed structure has strong implications for future wideband DRF designs, and thus has the potential to improve the design of wireless communication systems. Acknowledgements This work was supported by the National Natural Science Foundation of China under (Grant Nos. 61774020, 61671085, 61377097, 11574311 and 61690195), Fund of IPOC BUPT (Grant No. IPOC2017ZT06) and Fundamental Research Funds for the Central Universities (Grant No. 2018XKJC05), China. References
Fig. 7. The simulation results of (a) electric and (b) magnetic field distributions of the proposed wideband DRF, here g ¼ 6 mm at 6.5 GHz.
mez-García, A.C. Guyette, IEEE Trans. Microw. Theor. Tech. 63 (2015) [1] R. Go 1294. [2] R. Ramírez-Villegas, I. Huynen, L. Piraux, A. Encinas, J.D.L.T. Medina, IEEE Trans. Microw. Theor. Tech. 65 (2017) 72. [3] Z. Li, K.L. Wu, IEEE Trans. Microw. Theor. Tech. 65 (2017) 3789. [4] J. Lee, E.J. Naglich, H.H. Sigmarsson, D. Peroulis, W.J. Chappell, IEEE Trans. Microw. Theor. Tech. 61 (2013) 1114. [5] Q. Xue, J.Y. Jin, IEEE Trans. Microw. Theor. Tech. 65 (2017) 4103. [6] H.J. Xu, K. Bi, Y.N. Hao, J.M. Zhang, J.C. Xu, J. Dai, K. Xu, J. Zhou, IEEE Antennas Wirel. Propag. Lett. 17 (2018) 2494. [7] S.H. Chien, Y.S. Lin, IEEE Access 5 (2017) 18847. [8] K. Bi, L. Zeng, K. Chai, Z. Fan, L. Liu, Q. Wang, M. Lei, Appl. Phys. Lett. 107 (2015), 064103. [9] R. Azadegan, K. Sarabandi, IEEE Trans. Microw. Theor. Tech. 52 (2008) 1548. [10] S.Y. Eom, I.P. Hong, Microw. Opt. Technol. Lett. 52 (2010) 1880.
K. Bi et al. / Journal of Alloys and Compounds 785 (2019) 1264e1269 [11] M. Esmaeili, J. Bornemann, P. Krauss, IET Microw. Antennas Propag. 9 (2015) 1307. [12] M. Esmaeili, J. Bornemann, IEEE Microw. Wireless Compon. Lett. 27 (2017) 40. [13] R.R. Mansour, IEEE Microw. Mag. 10 (2009) 84. [14] X.X. Yuan, L.H. Zhou, J.X. Chen, IEEE Microw. Wireless Compon. Lett. 28 (2018) 789. [15] J.X. Xu, X.Y. Zhang, IEEE Trans. Microw. Theor. Tech. 66 (2018) 3297. [16] Y.Y. Zhu, Y.L. Li, J.X. Chen, IEEE Microw. Wireless Compon. Lett. 28 (2018) 591. [17] J.X. Chen, J. Li, J. Shi, IEEE Microw. Wireless Compon. Lett. 28 (2018) 657. [18] M. Song, I. Iorsh, P. Kapitanova, E. Nenasheva, P. Belov, Appl. Phys. Lett. 108
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(2016), 023902. I.M. Reaney, D. Iddles, J. Am. Ceram. Soc. 89 (2006) 2063. H. Hughes, D.M. Iddles, Appl. Phys. Lett. 79 (2001) 2952. X.Y. Zhang, J.X. Xu, J.X. Chen, IEEE Trans. Microw. Theor. Tech. 65 (2017) 2101. X. Wang, K. Bi, Y. Hao, M. Lei, J. Alloys Compd. 749 (2018) 363. X.P. Chen, K. Wu, IEEE Trans. Microw. Theor. Tech. 56 (2008) 142. Y. Hao, Q. Wang, X. Gao, S. Huang, K. Bi, J. Alloys Compd. 702 (2017) 664. F. Zhang, V. Sadaune, L. Kang, Q. Zhao, J. Zhou, D. Lippens, Prog. Electromagn. Res. 132 (2012) 587.
Contents lists available at ScienceDirect
Journal of Alloys and Compounds journal homepage: http://www.elsevier.com/locate/jalcom
Wideband slot-coupled dielectric resonator-based filter Ke Bi a, Xuying Wang a, Yanan Hao a, Ming Lei a, *, Guoyan Dong b, **, Ji Zhou c, *** a State Key Laboratory of Information Photonics and Optical Communications, School of Science, Beijing University of Posts and Telecommunications, Beijing 100876, China b College of Materials Science and Opto-Electronic Technology, University of Chinese Academy of Sciences, Beijing 100049, China c State Key Laboratory of New Ceramics and Fine Processing, School of Materials Science and Engineering, Tsinghua University, Beijing 100084, China
a r t i c l e i n f o
a b s t r a c t
Article history: Received 28 November 2018 Accepted 24 January 2019 Available online 25 January 2019
A simple and novel slot-coupled dielectric resonator-based filter (DRF) with wideband performance is proposed, and the designed geometric parameters of the filter are discussed in detail. The proposed filter comprises a substrate, a microstrip line, and three dielectric resonators (DRs), which are positioned on the center of the corresponding slots to obtain a wide stopband response. Numerical simulations show that the wideband DRF has a center frequency of 6.5 GHz, which is consistent with the experimental measurement results. The simulation, as well as the experimental results demonstrate that the distance between the adjacent DRs is a key parameter that can be used to tune the bandwidth of the wideband DMF. The physical mechanism of the DRs is also revealed through analysis of the electromagnetic field distribution results. Additionally, the influence of the DRs on the stopband frequency is also discussed. The results indicate that the DRs play a critical role in the proposed filter. Furthermore, this paper presents the first report on a wideband DRF. © 2019 Elsevier B.V. All rights reserved.
Keywords: Wideband Dielectric resonator filter Band-stop filter Microwave filter
1. Introduction With the rapid development of mobile wireless communication systems, microwave filters have been progressively attracting an increasing amount of attention [1,2]. Both band-pass and band-stop filters play the role of important selective signal devices in various wireless communication systems [3,4]. Band-pass filters enable the transmission of desired signals. On the contrary, band-stop filters suppress the undesired signals in various applications [5,6]. Recent advances in microwave transmission systems have increased the demand for band-stop filters with good frequency response, such as filters with a wideband rejection feature, or those with excellent in-band performance [7,8]. Traditional band-stop filters include an electronic bandgap and defected ground structures such as double spiral, which place restrictions on miniaturization [9,10]. Substrateintegrated waveguide structures are also implemented in bandstop filters, which is typically integrated into microstrip or coplanar waveguide, and also other planar circuits [11]. However, it
* Corresponding author. ** Corresponding author. *** Corresponding author. E-mail addresses: [email protected] (M. Lei), [email protected] (G. Dong), [email protected] (J. Zhou). https://doi.org/10.1016/j.jallcom.2019.01.286 0925-8388/© 2019 Elsevier B.V. All rights reserved.
is difficult to manufacture band-stop filters on a large scale because of their complex structures [12]. Dielectric resonator (DR) is widely employed in wireless communications system due to its unique characteristic, such as highquality factor [13]. It is known that the propagation of electromagnetic waves through the substrate can be regulated by simply controlling the interaction between electromagnetic waves and dielectric materials [14e16]. Recently, dielectric resonators (DRs) have garnered a considerable amount of interest owing to their ability to be easily integrated into communication systems [17,18]. The compact size and low production cost of dielectric resonatorbased filters (DRFs) make them a promising substitute for traditional filters in wireless communication systems [19]. Despite these desirable characteristics, most DRFs are reported to possess narrow bandwidth properties, which are not advantageous in practical applications [20,21]. Recently, we have designed a dielectric resonator filter with thermally tunable characteristic [22]. But, this filter still exhibits a narrow band property. In this work, a novel wideband slot-coupled dielectric resonator-based band-stop filter is developed. In the proposed DRF design, three DRs are respectively mounted onto three slots. This design is based on the idea that the stopband bandwidth can be varied by adjusting the distance between adjacent DRs, while also minimizing deviation from the 6.5-GHz center frequency of the
K. Bi et al. / Journal of Alloys and Compounds 785 (2019) 1264e1269
stopband. The obtained numerical simulation results are found to agree well with the corresponding experimental measurement results. 2. Prototype filter design Fig. 1 draws the 3-dimensional geometry of the single slotcoupled DMF. On the ground plane, a DR with side length dimensions of a ¼ 2 mm and c ¼ 3.8 mm, and dielectric constant ε ¼ 35, is fixed above the substrate, at the center of the slot, as is shown in Fig. 1(a). Corresponding to the top view in Fig. 1(b), the
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F4B substrate has a dimension of L ¼ 40 mm, W ¼ 20 mm, and thickness t ¼ 1.0 mm. Its relative permittivity and loss tangent are εr ¼ 2.65 and tand ¼ 0.002, respectively. A slot with length b and width d is etched into the center of the substrate. A microstrip line with dimensions of L w ¼ 40 2.7 mm, a thickness of 0.035 mm, and a resistance of 50 U is printed onto the F4B substrate, as is shown in Fig. 1(c). CST Microwave Studio 2011 based on the finite integration technique (FIT) was used to numerically simulate the varying geometric conditions of the proposed filter. Fig. 2(a) illustrates the effects of varying the slot length b on the simulation results of the
Fig. 1. Schematic diagrams for the single slot-coupled dielectric resonator-based filter. (a) 3D view, (b) top view, and (c) bottom view.
Fig. 2. The simulation results of transmission spectra for the single slot-coupled dielectric resonator-based filter with different structure parameters of (a) b, (b) d, (c) L, and (d) w.
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transmission spectra for the single slot-coupled DRF. In this simulation, the parameters of slot width, microstrip line length, and width were set as d ¼ 1.6, L ¼ 40, and w ¼ 2.7 mm, respectively. From Fig. 2(a), it is obvious that the increase of slot length b reduces the filter resonance frequency, whereas the rejection bandwidth is not significantly changed. The effects of varying the slot width d on the simulated transmission spectra are illustrated in Fig. 2(b); the relationship between the slot width d and resonance frequency can be clearly observed. In this simulation, the parameters of slot length, microstrip line length, and width were set as b ¼ 8.0, L ¼ 40, and w ¼ 2.7 mm, respectively. When increasing the slot width d from 2.0 to 3.2 mm, the resonance frequency and rejection bandwidth increase. Fig. 2(c) illustrates the effects of varying the microstrip line length L on the simulation results of the transmission spectra for the single slot-coupled DRF. In this simulation, the microstrip line width, slot length, and width were set as w ¼ 2.7, b ¼ 8.0, and d ¼ 1.6 mm, respectively. The increase of the microstrip line length L can be observed to have negligible influence on the resonance frequency, whereas the insertion loss exhibited an initial increase and subsequent decrease. The effects of the microstrip line width w variation on the simulated transmission spectra of the single slot-coupled DRF are illustrated in Fig. 2(d). In this simulation, the parameters of microstrip line length, slot length, and width were set as L ¼ 40, b ¼ 8.0, and d ¼ 1.6 mm, respectively. We can see that both the resonance frequency and insertion loss slightly increase in response to the microstrip line width w being increased. It is well known that the energy attenuation can be increased by the decrease of the insertion loss. On account of the results of the aforementioned numerical simulations, the optimized geometric parameters b, d, L, and w of the single slot-coupled DMF were chosen as 8.0, 1.6, 40, and 2.7 mm, respectively. Fig. 3 gives the schematic diagrams of the configuration and geometry for the designed wideband slot-coupled DRF. The slot and DR structures shown in Fig. 1(a) were duplicated and integrated into the original structure, as is illustrated in Fig. 3(a). Furthermore, Fig. 3(b) diagrammatizes the added parameter g, which is the distance between two adjacent DRs. The dimensions of the substrate, slot, and microstrip are the same as those illustrated in Fig. 1(b) and (c). To excite the electromagnetic resonance of each DR, three DRs are respectively mounted onto three slots.
3. Simulation and measurement results In order to verify the excellent performance of the designed DRF, a filter with a center frequency of 7.4 GHz was fabricated (Fig. 4). The proposed filter was fabricated on a printed circuit board (PCB) [23], which was then placed in a metal cavity. We have fabricated
Fig. 4. Photograph of the fabricated wideband slot-coupled dielectric resonator-based filter.
Fig. 3. Schematic diagrams of (a) 3D, (b) top, and (c) bottom views for the wideband slot-coupled dielectric resonator-based filter.
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four filters with different g values, such as g ¼ 4 mm, g ¼ 6 mm, g ¼ 8 mm, and g ¼ 10 mm. A diamond wire cutting machine was used to cut the ceramic to dimension of 2 mm 2 mm 3.8 mm for the proposed DR [24]. As can be seen, three DRs are positioned on the slot in turn. Note that all DRs are identical, made of BaZn0.33Nb0.67O3-Ba3CoNb2O9 (0.5BZN-0.5BCN, BCZN) ceramic with high quality factor Q f and low sintering temperature, and have a relative permittivity of 35 and a tangent loss of 0.002. The measured X-ray diffraction (XRD) pattern and dielectric constant of the BCZN ceramic are shown in Fig. 5. From Fig. 5(a), it can be seen that all peaks can be attributed to BCZN, and no additional peaks for impurity or intermediate phases are found. Fig. 5(b) displays that the dielectric constant of the BCZN ceramic is nearly constant within the frequency of 1 kHze10 MHz. Fig. 6 depicts the simulated and measured transmission spectra of the proposed DRF, which are in good agreement. The simulated transmission spectra are shown in Fig. 6(a) under the condition of an approximately 6.5 GHz center frequency. It can be observed that the bandwidth decreases as the distance g increases, and that the filter has an insertion loss of more than 19 dB in the wideband rejection region. Alternatively, experimental measurements were
Fig. 6. Effects of varying the distance g between the neighboring DRs on the (a) simulated and (b) measured transmission spectra for the filter.
Fig. 5. Measured XRD pattern and (b) dielectric constant of the BCZN ceramic.
carried out by an E5063A vector network analyzer to directly measure the two-port S-parameters. We have measured four filters with different g values, such as g ¼ 4 mm, g ¼ 6 mm, g ¼ 8 mm, and g ¼ 10 mm. In Fig. 6(b), the measured transmission spectra are centered at 7.4 GHz; moreover, under these conditions, the insertion loss in the wideband stopband is higher than 18 dB. Furthermore, the bandwidth can be observed to decrease as the distance g is increased from 4 to 10 mm. Considering the above analysis, all of the DRs are believed to be operating as based on the same principle: the coupling of the three slot-DR pairs results in wideband characteristics. Moreover, it can be seen that the mutual coupling effect decreases as the distance g increases. Table 1 presents a detailed comparison of the simulated and measured results for the proposed wideband DRF, and confirms the broadband feature of the filter. Both the simulation and the measured values indicate that the electromagnetic energy in the designed wideband DRF sharply attenuates within a specific frequency range. And due to the manufacturing tolerance, the simulated and measured results have minor differences. In the designed filter, the physical mechanism of the DRs is
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Table 1 Comparison of simulated and measured results for the proposed DRF. g (mm)
Sim.10-dB FBW
Mea. 10-dB FBW
Sim. IL (dB)
Mea. IL (dB)
4 6 8 10
34.8% 31.1% 28.0% 26.2%
46.5% 44.6% 39.9% 39.5%
19 25 28 33
18 19 21 20
FBW: Fractional Band-Width, IL: Insertion Loss.
revealed by the subsequent investigation into the electric and magnetic field distributions at 6.5 GHz. Fig. 7(a) shows the simulation result of the electric field distribution of the proposed wideband DRF under the conditions of g ¼ 6 mm at 6.5 GHz. In the DR, the displacement current is horizontal, i.e., y-directional. Fig. 7(b) shows the simulated magnetic field distribution in the DR, which is clearly observed to be strong, and exhibits a circular pattern. Particularly, in the DR, the generated strong displacement current as a result of the electric field of the electromagnetic waves caused a surrounding circular magnetic field; this phenomenon is considered to be electric resonance [25]. It should be noted that the strong magnetic field intensity is still observed above each slot, which is caused by the electromagnetic waves oscillation between the microstrip line and the ground plane along the z-direction. The influence of the DR on the proposed wideband DRF were also studied. Fig. 8 displays the simulated electric energy density distributions in the proposed filter under the conditions of g ¼ 10 mm and varying frequencies. Fig. 8(a) shows the three DRs coupled to their corresponding slots and generating electromagnetic waves that propagate through the substrate from one port to the other. Fig. 8(b) shows only the first DR to be coupled to the slot, demonstrating that the electromagnetic energy is mostly confined within the first DR; this means that electromagnetic waves cannot transmit through the substrate, and that the first DR significantly influences the stopband frequency of 6.5 GHz. Fig. 8(c) shows the first two DRs coupled to their corresponding slots, demonstrating that the electromagnetic energy is basically confined within the first two DRs; this indicates that the first two DRs are responsible for the stopband frequency of 7.0 GHz. Fig. 8(d) shows all three of the DRs coupled to their corresponding slots; the propagation of
Fig. 8. Simulated electric energy density distributions of the wideband DRF with g ¼ 10 mm at (a) 4.0 GHz, (b) 6.5 GHz, (c) 7.0 GHz, and (d) 9.0 GHz.
electromagnetic waves through the substrate demonstrates good agreement with the results illustrated in Fig. 6. Considering these results, it can be concluded that the stopband frequency is significantly dependent on the mutual coupling effect between the slotDM pairs. 4. Conclusions A novel wideband slot-coupled DRF has been fabricated and evaluated in this study. The consistency between the simulation and the measurement confirm that the proposed filter has the ability to suppress undesirable wideband signals. Moreover, the electric and magnetic field distributions provide insight into the electric resonance mechanism of the DMs. The electric energy density distributions further demonstrate the important role of DRs in the filter. The proposed structure has strong implications for future wideband DRF designs, and thus has the potential to improve the design of wireless communication systems. Acknowledgements This work was supported by the National Natural Science Foundation of China under (Grant Nos. 61774020, 61671085, 61377097, 11574311 and 61690195), Fund of IPOC BUPT (Grant No. IPOC2017ZT06) and Fundamental Research Funds for the Central Universities (Grant No. 2018XKJC05), China. References
Fig. 7. The simulation results of (a) electric and (b) magnetic field distributions of the proposed wideband DRF, here g ¼ 6 mm at 6.5 GHz.
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