A chip-scale frequency down-conversion realized by MEMS-based filter and local oscillator

Journal Pre-proof A Chip-Scale Frequency Down-Conversion Realized by MEMS-Based Filter and Local Oscillator Jyoti Satija, Sukomal Dey, Shashwat Bhattacharya, Gayathri Pillai, Sheng-Shian Li

PII:

S0924-4247(19)30544-8

DOI:

https://doi.org/10.1016/j.sna.2019.111787

Reference:

SNA 111787

To appear in:

Sensors and Actuators: A. Physical

Received Date:

29 March 2019

Revised Date:

13 November 2019

Accepted Date:

6 December 2019

Please cite this article as: Satija J, Dey S, Bhattacharya S, Pillai G, Li S-Shian, A Chip-Scale Frequency Down-Conversion Realized by MEMS-Based Filter and Local Oscillator, Sensors and Actuators: A. Physical (2019), doi: https://doi.org/10.1016/j.sna.2019.111787

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A Chip-Scale Frequency Down-Conversion Realized by MEMSBased Filter and Local Oscillator Authors: Jyoti Satija1, Sukomal Dey2, Shashwat Bhattacharya1, Gayathri Pillai1, and ShengShian Li1,3

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Affiliations: 1 Institute of NanoEngineering and MicroSystems, National Tsing Hua University, Hsinchu, Taiwan 2 Department of Electrical Engineering, Indian Institute of Technology, Palakkad, India 3 Department of Power Mechanical Engineering, National Tsing Hua University, Hsinchu, Taiwan

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Fax: +886-3-574-5454 E-mail: [email protected]

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Corresponding author: Sheng-Shian Li Institute of NanoEngineering & MicroSystems, National Tsing Hua University, Hsinchu, Taiwan Tel: +886-3-576-2401

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Graphical abstract

Highlights 1.

Wide clamped-clamped (CC) beam resonator and filter has been designed and characterized to achieve mixler operation on a chip.

2.

Performance comparison has been provided for high-stiffness and low-stiffness driving configuration in CC-beam resonator. The resonator has been operated in the non-linear region to improve the phase noise of the oscillator using phase-feedback mechanism.

The local oscillator signal (LO) required for the mixler operation is provided using

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3.

the designed resonator thus avoiding the requirement of an external instrument or

The proposed mixler achieves combined loss of 14.4dB and mixer’s conversion

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a function generator.

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loss of 8.7dB, with RF-LO-IF isolation better than 40dB, where RF and IF are radio frequency and intermediate frequency signals.

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Abstract

A wide-width clamped-clamped beam (CC-beam) resonator and a filter realized

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using mechanically-coupled resonator tanks have been extensively characterized to attain mixler (i.e., mixer-filter) functionality for a chip-scale frequency translation. Such a resonator has been driven into the nonlinear regime to explore its effect on the mixler operation as well as to obtain the best phase noise performance using the phase-feedback method. The proposed electrostatically transduced mixler operates by utilizing the local oscillator (LO) signal generated from the resonator itself, thus eliminating the need of a function generator or an external quartz crystal oscillator. This technique implemented with all the components on the same silicon substrate shows great potential for miniaturization 2

in wireless communication while achieving 14.4dB overall mixler loss.

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Keywords: Resonator, Filter, Phase-feedback, Nonlinearity, Local Oscillator, Mixler.

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1. Introduction Micro-Electro-Mechanical

Systems

(MEMS)

components

for

wireless

communication have already been explored for decades and have proven themselves to be very useful in many applications [1] [2] due to the ability of integration on a single chip and their very small form factor. This is why even after the exceptional performance of the quartz crystal oscillators (aging, thermal stability and excellent Quality factor (Q)), micromechanical resonators have been developed to provide decent Q value and power handling capability, for frequency selection [3] and timing reference [4], respectively, over the past few decades.

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Nevertheless, there are still two major challenges for capacitive resonators and oscillators to optimize their use in different applications mentioned above, including (i) power handling and (ii) phase noise [5]. To address these issues, choosing the proper resonator design and the electrode configuration is of paramount importance. In this work, a flexural beam resonator design is adopted and the power handling of the device is improved by using two schemes. One is the wide beam design and the other is choosing high-stiffness driving over low-stiffness driving configuration [6]. CC-beam has been considered as the simplest design so

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far along with its simple surface micromachining process making it attractive for low cost, on-chip oscillator applications in both academia [4] [7] and industry [8]. The same resonator design and configuration [6] have been used in this paper for further analysis. To make it function as a high frequency (HF) filter, this resonator is mechanically coupled with another identical resonator designed at the same frequency [3].

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Furthermore, there are various techniques that are being embraced in the nonlinear regime [9] to improve the phase noise and power handling performance such as by utilizing the bifurcation points [10], bias-dependent nonlinearity [11], cancellation between electrostatic and mechanical nonlinearity by finding intermediate voltage level [12], etc. In this work, the phase-feedback approach [13] has been used which utilizes the oscillation loop to attune the resonant frequency by tuning the phase of PLL (phase-locked loop) which has been used as a sustaining circuitry here. This technique is experimentally applied for an electrostatically actuated CC-beam resonator operating in the nonlinear region. And using this, phase noise is minimized by finding the operating point that can control both the frequency as well as the output signal power. Applications of a resonator are well-known from its ability to perform as an 4

oscillator and a filter (coupled resonators), which are then commingled to act as a mixler. A mixler is able to serve the function of both mixer and filter, which are the core components of a superheterodyne receiver. Directing a considerable effort in operating them together instead of using two off-chip components offers benefits like [14] (i) low power consumption, (ii) size reduction, and (iii) no need for additional impedance mismatching circuits between the mixer and the IF filter.

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However, it would be promising if the mixler operation can be achieved on a single chip without the need for an external function generator or crystal oscillator which has always been used to generate the LO signal. And this defines the main goal of this paper. The resonator and filter designed together on the same chip successfully achieves the main requirement which is filtering for the frequency selection and mixing to up-convert or down-convert the radio frequency (RF) signal to intermediate frequency (IF) signal. The mixler functionality is realized through

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capacitive electromechanical mixing, utilizing the nonlinear force quadratic equation which will be described later. The schematic representing the intended on-chip mixler operation is shown in Fig. 1.

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Fig. 1. Schematic of the proposed on-chip frequency translation through the MEMSbased mixler and resonator (serves as local oscillator). This paper has been organized as follows: Section II starts with the design part of the MEMS resonator, oscillator, and filter for the mixler application. Section III describes the test setup followed by section IV that briefly explains the traditional polysilicon surface micromachining fabrication process overview. Section V is subdivided into sections explaining the detailed experimental results corresponding to resonator nonlinearity, filter, and mixler response. Finally, section VI provides the concluding remarks of the work. 5

2. Design and Characterization 2.1. Resonator Design Fig. 2(a) presents the design of a wide beam high-stiffness driven CC-beam resonator. As shown, this device consists of a single beam anchored to the substrate at its ends and suspended above the electrodes. By employing the wide beam design and asymmetric drive and sense configuration (in which sensing is done at low-stiffness or high-velocity area whereas driving is done at high-stiffness or low-velocity location), low motional resistance and better power handling can be achieved, thus reducing phase noise when implemented as an oscillator.

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Resonator’s cross-sectional view is shown in Fig. 2(b), where Lr, h, and do are the original length, thickness and gap of the resonator, ws and wd are sensing electrode

and driving electrode widths, respectively. Under normal operation using the excitation scheme shown in Fig. 2(a), a dc-bias voltage, VP is applied to the

resonator structure while an ac signal, vac, is delivered to the underlying input given by Cd vac x

(1)

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Fd  2VP

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electrodes. These two voltages together generate an electrostatic driving force, Fd,

f ow

1/2

k E h   ke    1.03 1     2 m (1  )  L2   km  

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1  2

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where ∂Cd /∂x is the change in capacitance with the displacement of the resonator above the drive electrode and the multiplication factor 2 in the equation accounts for the two drive electrodes. This force then drives the beam into resonance when vac matches the frequency, fow, of the wide-width resonator, given by [15] (2)

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where E and ρ represent Young’s modulus and density of the structural material, ν is the Poisson’s ratio taken into account for a wide beam, h and Lr are the geometrical parameters, and κ is the scaling factor that depicts the resonant frequency dependence on the surface topography. Value of “κ” depends on the setup of anchor and finite elasticity effects. It can be predicted using finite element analysis [3]. ke and km are the electrical and mechanical stiffness [4]. This vibrating resonator beam then sources a dc biased capacitance change attributing to an output current, io, given by

io  Vp

Co Co x x  Vp  t x t t

(3)

The electrical current is then sensed at the output electrode. η in (3) defines the electrostatic transduction efficiency and ∂Co/∂x is the change in capacitance with 6

resonator displacement at the sensing port given by Co  oA o  2 x do

(4)

where, Ao=wrwe, is defined by the resonator and electrode overlap area and do is the

(a)

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original gap spacing.

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Fig. 2. (a) Schematic-view of a high-stiffness driven CC-beam resonator. (b) Resonator cross-sectional view with critical dimensions labeled.

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2.2. Oscillator Design and Operation The resonator operates as an oscillator if a feedback amplifier is augmented to it to form a closed loop as shown in Fig. 3. The sustaining amplifier shown can be any of the following: TIA (Transimpedance Amplifier), PLL (instrumentation, used here), off-chip commercially available amplifier or a tape-out IC.

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To maintain the oscillation across the closed loop, Barkhausen criterion needs to be followed which has two main conditions: (i) gain, A=(Ramp/Rtot) >1, where Ramp is the amplifier gain and Rtot=Rm+Ri+Ro is the total resistance, and (ii) phase around the positive feedback loop should be 0°. Here Ramp, Ri and Ro are the gain, input and output resistance of the amplifier, and Rm is the motional resistance of the CCbeam resonator. Once the criterion is met, a time-varying continuous sinusoidal output is observed in an oscilloscope and the oscillator should oscillate at just one frequency, providing delta function in the frequency domain. But practically, there are many losses or fluctuations resulting in sideband power other than the power at the fundamental frequency, treated as the phase noise [5] [16]. There are two components of noise in an oscillator: amplitude noise and phase noise. Generally, the amplitude output remains constant in an oscillator using automatic level control thus reducing the amplitude noise and making the phase noise dominant. Therefore, phase noise is a very important parameter to characterize an 7

oscillator. To reduce the oscillator phase noise, there are two important factors [5]: (i) Increasing the Q-factor, which can remove most of the noise at close-to-carrier frequency offsets, and (ii) Increasing the oscillation signal power, Po, which can improve the phase noise performance at far-from-carrier frequency offsets by

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increasing the signal-to-noise ratio. In this work, close-to-carrier phase noise has been improved using a wide beam high-stiffness driving scheme and phase-feedback approach.

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Fig. 3. Perspective schematic view of an oscillator driven at low-stiffness location for nonlinearity exploration.

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2.3. Filter Design Filters that make an important part of superheterodyne transceivers can now be miniaturized and integrated on-chip using the existing micromachining technologies [3]. It is designed in the HF range using two CC-beam resonators mechanically coupled together, and fabricated on the same chip as the resonator for LO, using polysilicon surface micromachining technology. The perspective view of the schematic is shown in Fig. 4 along with the biasing conditions. The mechanically coupled beam is a network of springs. Since there are two resonators coupled, its transmission response exhibits two peaks with closely spaced frequencies but separated by some offset that defines the filter bandwidth. This bandwidth can be tuned by changing the stiffness of the coupling springs. And the constituent resonators themselves define the center frequency of the filter. To flatten the passband or to move both the peaks at the same level, termination resistors are used [3]. The two resistors (RQ1 and RQ2) are applied at the input and output of the filter, respectively. Coupling beam dimensions are chosen as per the quarter wavelength of the center frequency of the filter, fIF, and its location is decided according to the required filter bandwidth [17]. Just like the resonator, it should have high Q as low resonator Q of a given filter corresponds to high insertion loss (I.L.). Ideally, the insertion loss for 8

a filter should be 0dB. Similar to the resonator, for the filter operation, VP is applied directly to the resonator together with an ac drive voltage, vin, that passes the signal to the first resonator through RQ1. This generates a force and drives the resonator when the frequency of vin falls within the filter passband. Then through the coupling

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spring, this energy of vibration is transferred to the other resonator making it resonate as well thus creating an output current, io, as given by (3). The output current gets converted to the voltage at the output terminal through RQ2.

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Fig. 4. Schematic view of a mechanically coupled filter using two wide-width CCbeam resonators for power handling enhancement. 2.4. Nonlinearity of Resonator and Oscillator Owing to the low phase noise requirement of the frequency stable oscillators, they

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ought to have a high signal to noise ratio. The nonlinear resonator can cause various instabilities and frequency fluctuation with amplitude variation (A-f effect), which tends to limit the device power handling. Thus, to suppress the

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nonlinearity, various ways have been reported [9] [12]. However, it is observed that operating the resonator at a particular condition in the nonlinear regime

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tremendously improves the device’s performance by suppressing A-f effect [13]. It is already proven that other than the high-stiffness driving design [6], and the resonator array design [18], close-to-carrier phase noise can be improved by operating the resonator in the nonlinear/anharmonic region [11]. Thus, nonlinearity should be exploited more to achieve potential benefits like improving the phase noise performance, SNR, extending the dynamic range of the resonator, etc. This work has achieved the same using the phase-feedback control method which will be explained in the results section.

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2.5. Mixler Mixer-filter is an amalgamation of both active and passive components like oscillator and filter, respectively [19]. Thus, a resonator and a filter are used to describe the mixler system using their equivalent circuits as shown in Fig. 5. As mentioned, a CC-beam

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resonator based on the high-stiffness driving principle is designed to have a resonant frequency of 9.7MHz. A filter is designed based on the same principle.

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Fig. 5. Schematic-view of the proposed MEMS-based mixler and LO using equivalent circuit diagrams of their constituent filter and resonator.

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From the nonlinear force-capacitance quadratic equation,

 1  CV 2   x  2 

(5)

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F

where V comprises DC, RF and LO voltage signals as the input. This leads to the

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combination of different input product terms providing sum and difference, which is the key to mixing. The force after applying the combination of input voltages turns out to be [ 1 4 ] F

1 C 2 vRF  vLO  VP  2 x

(6)

where vRF  VRF .cos RF t  and vLO  VLO cos LOt  . Expanding the quadratic equation,

F

1 C C C C vRF 2  vLO 2  VP 2   vRFVP  vLOVP  vRF vLO  2 x x x x 10

(7)

The term with interest to mixing is the one underlined above as it consists of both RF and LO signals which when multiplied together gives the resulting output force, Fo, constituted of the difference of frequencies corresponding to IF signal as shown, Fo 

C C vRF vLO  VRF cos RF t VLO cos LOt  x x

C VRFVLO cos  (RF  LO )t  x C  VRFVLO cos IF t  . x 

(8)

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3. Measurement Setup

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3.1. Resonator and Filter Standalone two-port CC-beam resonators with different widths (20, 35, 50 and 60µm) have been measured using the setup shown in Fig. 6. The primary parameter for resonator characterization is the natural frequency (ωo) at its resonance. In addition to the resonance frequency, Q-factor, motional resistance, and phase can also be extracted from the measurement results. The performance of the resonator is affected by the environment. Vacuum condition reduces the loss due to air damping, thus enhancing

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the quality factor, Q. Vacuum condition for the resonator is maintained by an additional pump to the Lakeshore probe station. The pressure in the chamber is expected to be around 10-4 Torr. Before starting the measurements, calibration is performed to remove the parasitic effects of cables by open, short, load and through test (i.e., SLOT).

Fig. 6. Measurement setup for a wide-width CC-beam high-stiffness driven resonator operating at vacuum. Agilent’s E5071C network analyzer (NA) with resolution of 1601 points has been used 11

to provide an ac input voltage to the drive electrode of the device. Keithley source measuring units (SMU) on the other hand has been used to provide dc bias directly to the structure. The cable through the sense electrode has been connected to the other port of the NA to record the output. For measuring the filter, the setup would be the same as the resonator.

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3.2. Oscillator Fig. 7 shows the perspective test setup for the oscillator measurement using the Zurich HF2LI lock-in amplifier and PLL. It can be seen that the output of the device (sense signal) is fed into the input of the lock-in amplifier (I/P port), where it is compared with the internal reference signal. Over a certain range, the phase and frequency of the input signal are followed by the digital data generated by the PLL block. Then the digital to analog converter (DAC) uses this data and generates a signal with the same phase and frequency as the input signal. However, the power of this signal can be set

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independently. This generated signal at O/P port of the PLL is used to drive the resonator (drive signal) allowing it to operate as an oscillator at a lower power level. The sense electrode has been connected with the spectrum analyzer (SA) to record the LO output.

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Fig. 7. Measurement setup for an oscillator with the resonator driven under lowstiffness configuration for the verification of phase noise reduction. 3.3. Mixler The measurement schematic of the mixler using both resonator and filter on the same chip is illustrated in Fig. 8(a) using a simplified block diagram. As shown, RF signal, vRF, is swept using NA and provided at the input of the filter. LO signal with oscillation amplitude, vLO, is simultaneously provided from the resonator operating as an oscillator through HF2LI without the need of an external function generator or quartz crystal oscillator, with the dc bias provided to both the resonator and filter beams. 12

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Fig. 8. (a). Simplified block diagram view for mixler + LO experimental setup with nonlinear resonator (LO) driven at optimum phase value. (b) Photo view with vacuum probe station, co-axial cables and associated instruments.

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Although HF2LI takes significant dc power and is an integral part of the experimental results, it can be replaced with the sustaining circuit or board level electronics in future. The primary focus in this work is to eliminate the need of the function generator, beneficial to realize miniaturization and potentially low power consumption. The

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generated output force with a frequency component containing the difference of RF and LO signal corresponds to the IF frequency (center frequency of the filter) which is monitored in the spectrum analyzer using MAX HOLD mode. The spectrum analyzer is set such that IF frequency falls within that frequency range. Fig. 8(b) shows the pictorial presentation of the setup with all the associated instruments.

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4. Fabrication Fabrication of the mixler obtained using the proposed resonator and filter design is done through the conventional polysilicon based surface micromachining process [3]. The process flow is defined for high-stiffness driven resonators with their cross-sectional view as shown in Fig. 9. 2µm thick oxide layer is grown over a phosphorus doped n+ Si wafer (100) using low-pressure chemical vapor deposition (LPCVD) to provide isolation between the device and the substrate followed by 0.35µm Nitride deposition. Again, using LPCVD, low-stress polysilicon layer is deposited incorporating PoCl3 doping to make it n++. Polysilicon is then patterned using dry etching which defines the electrodes for dc bias or drive/sense and interconnects. Next, 100nm thick hightemperature sacrificial oxide is deposited using LPCVD to define the gap spacing. The anchors are opened using reactive ion etching (RIE). Afterward, 2µm polysilicon is deposited as a structural layer and patterned using deep reactive ion etching (DRIE). Finally, sacrificial silicon dioxide is removed by releasing it in the Hydrofluoric (HF) 13

acid. After the release process, a beam structure which is free to vibrate can be obtained as shown in Fig. 9(d). Different width CC-beam resonators and filters have been fabricated with high-stiffness drive electrode design. Global Scanning Electron Microscope (SEM) images of the wide-width design resonator with both high-stiffness

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driving and low-stiffness driving configuration is represented in Fig. 10(a) and (b), respectively, whereas the view with both the resonator and the filter on the same chip is presented in Fig. 10(c).

Fig. 9. Detailed process flow for a CC-beam resonator with its cross-sectional view.

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5. Result and Discussion

5.1. Resonator In continuation of our previously reported resonator [6], it is well established that the larger width resonator will have better power handling. Table 1 summarizes the design and performance characteristics for 20µm and 60µm wide resonators verifying the same. Furthermore, Fig.11(a) presents the experimental frequency characteristics under various NA power levels for a 50µm wide CC-beam resonator with low-stiffness driving configuration in which driving and sensing is done at high-velocity and low14

velocity regions respectively under 8V dc bias at vacuum. It is observed that as the ac power is increased from -30dBm to 0dBm, the resonant peak keeps shifting to the lower frequency due to the softening effect and Duffing is observed at the higher input power.

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Fig. 10. SEM overview of a wide-width resonator with (a) high-stiffness driving configuration, (b) low-stiffness driving configuration, and (c) a MEMS mixer-filter + LO using the filter and resonator designed on a single chip. Also, the frequency shift is accompanied by lower resonance peaks leading to an

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increase of motional resistance, Rm. This is due to the existence of higher-order coefficients, which are attributed to the rise of both mechanical and electrostatic nonlinearity [12] [20] changing the motion equation to

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𝜕2𝑥 𝜕𝑥 (9) 𝑚 2 +𝑏 + (𝑘1𝑚 + 𝑘1𝑒 )𝑥 + (𝑘3𝑚 + 𝑘3𝑒 )𝑥 3 = 𝐹 cos(2𝜋𝑓𝑡) 𝜕𝑡 𝜕𝑡 Ignoring the quadratic nonlinear spring coefficient due to the symmetry of the device in the above equation, k1m and k3m are first and third-order mechanical spring constants. Whereas, k1e and k3e are first and third-order electrostatic spring coefficients. Individually considering mechanical and electrical nonlinearity, as the vibration amplitude is provided to the clamped-clamped beam, mechanical spring force corresponds to 𝐹𝑚 = −𝑘1𝑚 𝑥 − 𝑘3𝑚 𝑥 3 . 15

(10)

Parameters, k1m and k3m, here are greater than 0. Contrary to mechanical nonlinearity, for larger vibration amplitude, electrostatic nonlinearity reduces the stiffness of the resonator, thereby shifting the resonant frequency to lower frequency values. Net electrostatic force equation thus becomes, 𝐹𝑒 =

2𝜀𝑜 𝑤𝑒 ℎ 𝑑𝑜3

𝑉𝑃2 (𝑥 +

2 𝑑2

𝑥3 + ⋯ )

(11)

From the above equation, linear and cubic electrostatic nonlinear spring constants are defined as 𝑘1𝑒 = −

2𝜀𝑜 𝑤𝑒 ℎ 𝑑𝑜3

𝑉𝑃2

𝑘3𝑒 = −

4𝜀𝑜 𝑤𝑒 ℎ 𝑑𝑜5

𝑉𝑃2

(12)

The negative sign depicts the reduction of the resonant frequency. The above

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explanation describes that the shift in resonant frequency to the left side in Fig. 11 is

due to the existence of electrostatic nonlinearity where softening occurs. Next, Fig. 11(b) presents the measurement for the same device, under the same test conditions but

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with high-stiffness driving configuration.

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Fig. 11. Measured transmission versus frequency plots for a 50µm-wide clampedclamped beam resonator, showing Duffing nonlinearity as the input power increases under (a) low-stiffness driving and (b) high-stiffness driving setups.

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The same shift in frequency and Duffing at higher input power is seen in the plot. Since the Duffing is observed more in low-stiffness driving compared to high-stiffness driving for the same power (-10dBm) as shown in Fig. 12, it is clear that power handling of the device in the high-stiffness configuration is better than the other as the linear frequency characteristics remain preserved even at a higher NA power. This is because the maximum power that a CC-beam can handle while still preventing deleterious effects caused by nonlinearity is expressed as [18]

Pomax 

o kr  yd  a 2 do 2 Q 16

(13)

where a=0.56 (at resonance) is the fraction of the electrode-to-resonator gap beyond which the onset of strong nonlinearities ensues, do is the original gap spacing and kr(yd) is the effective stiffness of the resonator at its driving location, yd. It is clear from (13) that the power handling capability of the resonator mainly depends on the stiffness as

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the gap is limited by the low motional resistance design as well as the fabrication limits. Thus, it is definite that the power handling capability will be better for the high-stiffness driving configuration.

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Fig. 12. Measured transmission versus frequency plot for a 50µm-wide CC-beam resonator driven at the high-stiffness and low-stiffness locations under the same input power and dc bias.

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5.2. Oscillator The CC-beam resonator oscillator measurement is performed using the test setup depicted in Fig. 7. Fig. 13(a) presents the frequency spectrum plot for the oscillator using a 50µm wide CC-beam design under 8V dc bias and -5dBm input power (i.e., carrier power) for high-stiffness driving configuration which shows the output power of -36.12dBm. The performance of the superheterodyne receiver system is severely affected by the local oscillator’s performance, which is mainly characterized by its phase noise. Phase noise affects the signal integrity of overall frequency translation. Thus, it is very important to evaluate the phase noise performance of the local oscillator. Fig. 13(b) plots the phase noise variation (at 10kHz offset) with the carrier power (from HF2LI) for the same device under the same biasing conditions. The reduction of phase noise value is observed with the increase of carrier power. This can be validated from the Leeson’s model [21]. According to which, in steady-state, the phase noise at frequency offsets around the carrier frequency can be described by the expression L  fm 

2kT 1  FRamp   Rtot  Po  Rm

2    fo    1         2Ql f m  

(14)

where, fm is the offset from the carrier fo at which phase noise is being evaluated; k is 17

the Boltzmann’s constant, T is the temperature in Kelvin; FRamp is the sustaining amplifier’s noise factor; Po is the oscillator signal power and Ql is the loaded Q-factor. As indicated in (14), the phase noise performance of an oscillator is greatly influenced by the power handling capability of a micromechanical resonator referenced to it, which is better for the high-stiffness driving design (from Fig. 12). The best phase noise performance observed from Fig. 13(b) is -94dBc/Hz at 10kHz offset.

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5.3. Phase-Feedback Approach Another way to further improve the phase noise performance is by using the phase feedback approach. As mentioned earlier, this method tunes the resonant frequency by adjusting the phase of PLL inside the oscillation loop. The resonator can be made to oscillate at the highest amplitude value using the phase-adjusted closed-loop approach. To obtain the best phase noise performance, the oscillator is operated at discrete bifurcation points obtained within the hysteresis and varying the phase values in a

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particular phase span. Using the phase-feedback approach, an unstable region in the resonator can be made stable by making the frequency less sensitive to the change in phase at a particular phase value [22]. Also, the AM-to-PM noise conversion leading to (i) worst phase noise, (ii) instability at bifurcation point, (iii) change in frequency causing a change in amplitude [9], and (iv) generation of 1/f 3 component is significant in the nonlinear regime.

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But, by tracking the specific phase in the feedback loop, this noise is canceled resulting in an optimized phase noise value. This occurs when the coupling between frequencyphase and amplitude-phase becomes zero. For observing the hysteresis, the device has to be driven in the nonlinear region as it can only be seen with the Duffing behavior. A 50µm resonator oscillator is operated at 0dBm carrier power and 7.9V dc bias for this approach under low-stiffness driving configuration. To explore the nonlinearity of the resonator, low-stiffness driving is chosen. Since the device can be made to be operated in any of the driving state (high or low-stiffness), the designed configuration helps. Once the phase is locked at the biasing condition specified above, hysteresis is observed for the frequency phase curve on bidirectional sweep due to the intertwining of mechanical and electrical nonlinearity. As shown in Fig. 14(a), at the condition where third order and other higher-order terms co-exist, double-cycle (bidirectional sweep) hysteresis is seen. The amplitude and phase noise value are observed at all the phase points varying from -164.6° to 87.1° within the hysteresis loop. By operating the oscillator at 85°, the best phase noise plot is obtained as shown in Fig. 14(b). The shaded area (after 18kHz) shown in the figure is due to shaping as per the internal bandwidth setting of the lock-in and PLL. The purpose of finding the optimal point is for the mixler operation explained later. 18

(a)

(b)

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Fig. 13. (a) Fourier spectrum and (b) plot of phase noise (at 10kHz offset) versus PLL drive power (from O/P port of Zurich HF2LI) for a 50µm-wide CC-beam resonator oscillator driven at 8V dc-bias.

(b)

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Fig. 14. (a) Observation of hysteresis loop due to Duffing, (b) Best case phase noise performance plot using phase feedback control method.

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5.4. Filter The standalone filter measurements are first done in a vacuum. Just like the resonator, the filters with different widths are measured under different biasing conditions. Fig. 15 shows the measured transmission response for a filter made of two mechanically coupled 50µm wide beam resonators, driven at 8.5V dc bias and -35dBm ac power. The red curve shows the filter transmission plot with the center frequency of 9.707MHz, insertion loss of 47.947dB, unterminated jagged passband. The termination of the filter is then carried out using fixture simulator of the NA shown by the blue curve. The value of the termination resistor used here is RQ1=3.2kΩ and RQ2=2.2kΩ which is mainly determined by the electromechanical coupling of the resonator element, and resonator’s and filter’s Q as well [23]. For the same biasing conditions, the terminated filter shows the insertion loss of 5.736dB. By adjusting the dc bias using separate tuning electrodes 19

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(to decouple from I/O), frequency tuning can be done to achieve the required passband shape. Table II summarizes the designed and measured filter characteristics.

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Fig. 15. Transmission response for a 50µm filter driven at 8.5 V dc-bias and -35dBm ac power (with and without termination). 5.5. Mixler

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The 9.7MHz IF mixler function has been achieved using the resonator and filter on the same chip with the measurement setup shown in Fig. 8. Important input components for the mixler operation are LO and RF signal. Until now [14] [24] [25], the LO signal

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has always been obtained from an external source like a function generator. In this work, for the first time, the resonator has been utilized to attain the LO signal achieving onchip mixler operation. The performance of the mixler is characterized by its conversion

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loss which is defined as the ratio of output IF signal obtained for the given input RF signal through the use of LO signal. Thus, to have a good conversion loss, LO signal plays a critical role indicating the need of better oscillator, with the best phase noise

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performance. In order to achieve the best phase noise, the aforementioned phasefeedback technique has been adopted and the optimum phase point has been found.

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This is the reason the nonlinear resonator has been employed in the measurement setup shown in Fig. 8. The LO of the mixler is implemented by a 50µm wide resonator driven at 0dBm, 7.9V bias and locked using PLL for the oscillator operation. For the best phase noise performance in the nonlinear region, the oscillator has been chosen to operate at 85° phase point as mentioned earlier. This oscillator signal has then been provided at the filter biasing line. The filter used for mixler measurement is a 50µm wide CC-beam filter centered at 9.77MHz. Thus, VP is adjusted to have a LO frequency of 8.8MHz and 20

vLO=0.5VRMS. RF signal varying from 11-20MHz is provided using Agilent 4-port network analyzer. And, at the spectrum analyzer, with the frequency sweep from 9.2 to 10.2MHz using MAX HOLD mode, an IF signal corresponding to the targeted filter center frequency is obtained at the output as shown in Fig. 16(a). This is due to the fall of the RF-LO component in the required IF passband. The observed mixler’s output power is -77.176dBm and the corresponding frequency is 9.77MHz. To make the mixler passband region between two peaks flat, the tuning voltage can be adjusted. To allow a smooth spectrum, multiple sweeps are done during the measurement. The through-measurement result, as depicted by a red line in the

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figure, is done by bypassing the mixler device using short circuit. This allows to extract the combined mixer’s conversion loss and filter’s insertion loss (overall ~14.4dB). Subtracting the filter’s insertion loss (5.7dB) from the combined losses (14.4dB) using through gives the mixer’s conversion loss (8.7dB). Note that, the input and output

matching section will improve the flatness of the IF spectrum and will be taken care of

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as future work. Better IF to LO isolation is attributed to on-board LO signal and tuning

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capability. Fig. 16(b) shows RF-LO-IF isolation of better than 40dB.

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This proposed work demonstrates successful implementations of a MEMS-based RF mixler that is a combination of mixing and RF filtering. This technique overcomes the inter-stage matching section and its driver circuit requirement that entails added power consumption.

(a)

(b)

Fig. 16. Mixler (a) response using a 50µm resonator driven at optimum phase condition and a 50µm filter driven at 0.5V VLO, (b) isolation response for three stages; RF-IF, RF-LO and LO-IF, respectively.

6. Conclusions For better power handling and low motional resistance, a high-stiffness driven wide 21

CC-beam has been evaluated followed by the characterization of a standalone filter. Using a PLL, the resonator is made to perform as an oscillator. The nonlinear region of the resonator has been explored for improving the oscillator performance. To further improve the phase noise performance, phase-feedback approach has been followed to

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attain the optimum phase conditions. All the phase points are tracked in the hysteresis loop and the best phase point is chosen. At the same phase condition, the resonator has been driven to achieve the mixler operation to attain better conversion loss but using the low-stiffness driving configuration due to better nonlinear exploration property. This explains the importance of the device design configuration (high-stiffness and lowstiffness), which provides the freedom of operation. Mixler functionality has been finally achieved by fabricating a resonator and a filter together on the same chip thus eliminating the need of external function generator generating the LO signal. This leads to miniaturization and potentially low power consumption. Both filtering as well as down-conversion of RF signal from 18.57MHz to 9.77MHz has been achieved with a

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conversion loss of 8.7dB with isolation better than ~40dB. This work aims to have a fully integrated system for its future version.

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Author Statement

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It is to certify that all authors have seen and approved the final version of the manuscript being submitted. This work hasn’t received any prior publication and isn’t under consideration for publication elsewhere.

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Declaration of interests The authors declare that they have no known competing financial interests or personal relationships that could have appeared to influence the work reported in this paper.

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ACKNOWLEDGEMENTS The authors would like to thank the funding support from the Ministry of Science and Technology (MOST) of Taiwan under the grant of MOST-107-2218-E-007-020 and the Toward World-Class University Project. The authors also appreciate the Center for Nanotechnology, Materials Science and Microsystems (CNMM) of National Tsing Hua University for the support of measurement facilities.

22

REFERENCES [1] C. T.-C. Nguyen, “MEMS technology for timing and frequency control,” IEEE Transactions on Ultrasonics, Ferroelectrics, and Frequency. Control, vol. 54, no. 2, pp. 251-270, 2007.

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[2] B. Kim, R. N. Candler, M. A. Hopcraft, M. Agarwal, W.-T. Park, and T. W. Kenny, “Frequency stability of wafer-scale film encapsulated silicon based MEMS resonators,” Sensors and Actuators A: Physical, vol. 136, no. 1, pp. 125-131, 2007. [3] F. D. Bannon, J. R. Clark, and C. T.-C. Nguyen, “High-Q HF microelectromechanical filters,” IEEE J. Solid-State Circuits, vol. 35, no.4, pp. 512-526, April 2000. [4] Y. W. Lin, S. Lee, S.-S. Li, Y. Xie, Z. Ren, and C. T.-C. Nguyen, “Series-resonant VHF micromechanical resonator reference oscillators,” IEEE J. Solid-State Circuits, vol. 39, no. 12, pp. 2477-2491, 2004. [5] M. Akgul, “Oscillator far-from-carrier phase noise reduction via nano-scale gap

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Proceedings, 25th IEEE Int. Micro Electro Mechanical Systems Conf. (MEMS’12), Paris, 2012, pp. 700-703. S. Lee, M. U. Demirci, and C. T.-C. Nguyen, “A 10-MHz micromechanical resonator Pierce reference oscillator for communications,” Tech. Dig., 11th Int. Conf. on Solid-State Sensors & Actuators (Transducers’01), Munich. Germany, June 10-14, 2001, pp. 1094-1097. Discera DSC1001 Datasheets, Discera Inc., San Jose, CA, 2011. M. Agarwal, “Nonlinear characterization of electrostatic MEMS resonators,” Proceedings, 2006 IEEE Frequency Control Symposium and Exposition, Miami, FL, 2006, pp. 209-212. G. Sobreviela, “Parametric noise reduction in a high-order nonlinear MEMS resonator utilizing its bifurcation points,” IEEE/ASME J. Microelectromech. Syst., vol. 26, no. 6, pp. 1189-1195, 2017. M.-H. Li, C.-Y. Chen, C.-H. Chin, and S.-S. Li, “Optimizing the close-to-carrier phase noise of monolithic CMOS-MEMS oscillators using bias-dependent nonlinearity,” Tech. Dig., 2014 IEEE International Electron Devices Meeting

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[7]

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tuning of micromechanical resonators,” Tech. Dig., Int. Conf. on Solid-State Sensors, Actuators and Microsystems Conference (Transducers’09), Denver, CO, 2009, pp. 798-801. [6] L.-J. Hou, W.-C. Chen, C.-S. Li and S.-S. Li, “High-stiffness-driven micromechanical resonator oscillator with enhanced phase noise performance,”

[11]

(IEDM’14), San Francisco, CA, 2014, pp. 22.3.1-22.3.4. [12] L. C. Shao, M. Palaniapan, and W. W. Tan, “The nonlinearity cancellation phenomenon in micromechanical resonators,” J. Micromech. Microeng., vol. 18, no.6, pp. 0960-1317, 2008. 23

[13] R. M. C. Mestrom, R. H. B. Fey, and H. Nijmeijer, “Phase feedback for nonlinear MEM resonators in oscillator circuits,” IEEE/ASME Transactions on Mechatronics, vol. 14, no. 4, pp. 423-433, 2009. [14] A.-C. Wong, and C. T.-C. Nguyen, “Micromechanical mixer-filters (“mixlers”),”

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IEEE/ASME J. Microelectromech. Syst., vol. 13, no. 1, pp. 100-112, 2004. [15] R.-D. Blevins (1929). Formulas for natural frequency and mode shape. Retrieved from https://www.scribd.com/document/349869032/212225572-BlevinsFormulas-for-Natural-Frequency-and-Mode-Shape-pdf. [16] D.-A. Howe, “Frequency Domain Stability Measurements: A Tutorial Introduction,” NBS Tech Note 679, March 1976. [17] C.-Y. Chen, M.-H. Li, G. Pillai, J. M. L. Tsai, and S.-S. Li, “An innovative piezoMEMS channel-select filter design based on non-monotonic coupled modes,” Proceedings, 30th IEEE Int. Micro Electro Mechanical Systems Conf. (MEMS’17), Las Vegas, NV, Jan. 22-26, 2017, pp. 950-953.

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[18] Y. W. Lin, S.-S. Li, Z. Ren, and C. T.-C. Nguyen, “Low phase noise arraycomposite micromechanical wine-glass disk oscillator,” Tech. Dig., 2005 IEEE International Electron Devices Meeting, (IEDM’05), Washington, DC, 2005, pp. 287-290. [19] B. Razavi, RF Microelectronics, Prentice Hall.

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[20] E. Kenig, “Optimal operating points of oscillators using nonlinear resonators,” Physical Review E, Statistical, Nonlinear, and Soft Matter Physics, vol. 86, p. 056207, 2012. [21] D. B. Leeson, “A simple model of feedback oscillator noise spectrum,” Proceedings of the IEEE, vol. 54, no. 2, pp. 329-330, 1966. [22] S. Lee and C. T.-C. Nguyen, “Influence of automatic level control on micromechanical resonator oscillator phase noise,” Proceedings, 2003 Joint Conference of the IEEE International Frequency Control Symposium and 17th European Frequency and Time Forum (IFCS’03), Tampa, FL, USA, 2003, pp.341349. [23] J. L. Lopez, J. Verd, A. Uranga, J. Giner, G. Murillo, F. Torres, G. Abadal, and N. Barniol. “A CMOS-MEMS RF-tunable bandpass filter based on two high-22-MHz polysilicon clamped-clamped beam resonators,” Electron Dev. Lett., vol. 30, no. 7, pp. 718-720, 2009. [24] F. Chen, J. Brotz, U. Arslan, C.-C. Lo, T. Mukherjee, and G. K. Fedder, “CMOSMEMS resonant RF mixer-filters,” Proceedings, 18th IEEE Int. Micro Electro Mechanical Systems Conf. (MEMS’05), Miami Beach, FL, USA, 2005, pp. 24-27. [25] S. Ilyas, N. Jaber, and M. I. Younis, “A coupled resonator for highly tunable and amplified mixer/filter,” IEEE Transactions on Electron Devices, vol. 64, pp. 26592664, 2017. 24

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Jyoti Satija received the B.Tech. degree in electronics and communication engineering from N.C. College of Engineering, Haryana, India, in 2013 and the M.Tech. degree in VLSI from the Amity University, Noida, India, in 2015. She did her M.Tech. Dissertation at Semiconductor Device Research Laboratory, Department of Electronic Science, University of Delhi, India. Her M.Tech research interests focused mainly on junctionless nanowire transistors. She is currently pursuing the Ph.D. degree from the Institute of NanoEngineering and MicroSystems, National Tsing Hua University, Hsinchu, Taiwan. Her research interests include micromechanical resonators/oscillators for

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wireless communication and signal processing.

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Sukomal Dey (S’10, M’15) received the B.Tech degree in Electronics and Communication Engineering from the West

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Bengal University of Technology, Kolkata, India, in 2006, the M.Tech degree in Mechatronics Engineering from the Indian Institute of Engineering Science and Technology (IIEST), Shibpur, India, and did M.Tech dissertation (one year) from Central Electronics Engineering Research Institute, Pilani, India, in 2009. He completed his Ph.D. in Microwave Engineering from Centre for Applied Research in Electronics, Indian Institute of Technology Delhi in July, 2015. From, August, 2015 to July, 2016 he was working as a project scientist in Industrial Research and Development (IRD) centre, IIT Delhi. He also worked on a collaborative research project supported by Synergy Microwave Corp., NJ, USA during the same period. From August, 2016 to June-2018, he was working in the Radio Frequency Microsystem Lab (RFML), National Tsing Hua University, Taiwan, as a Post doctorate Research Fellow. Since June-2018 he is working as an assistant professor at Department of Electrical Engineering, Indian Institute of Technology, Palakkad. He is recipient of postgraduate student award from Institute of smart structure and system (2012), Bangalore, India, best industry relevant PhD thesis award from the Foundation for Innovation in Technology Transfer (FITT) and Distinction in Doctoral Thesis, both from IIT Delhi (2016), Postdoctoral Research Fellow scholarship from Ministry of Science and Technology (MOST), Taiwan, (2016, 2017). He has been awarded with Early Career Research Award (ECRA) from the Science and Engineering Research 25

Board (SERB), India (2019). His research interests include development of microwave devices, and components for future RF front ends.

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Shashwat Bhattacharya received the B.Tech. degree in electronics and communication engineering from the Amity University, Lucknow, India, in 2011. In 2011, he joined Accenture Services Private Limited, Bangalore, India, where he was an Associate Software Developer involved in the development of software framework for Big Data Analysis using Information Management Tools of Informatica and SAP. He received his M.Tech. degree in VLSI from the Amity University, Noida, India, in 2015. He did his M.Tech. Dissertation at Semiconductor Device Research Laboratory, Department of Electronic Science, University of Delhi, India. He is currently pursuing the Ph.D. degree from the Institute of NanoEngineering

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and MicroSystems, National Tsing Hua University, Hsinchu, Taiwan. His current research interests include nano/microelectromechanical systems, RF MEMS, SOIMEMS thermal piezoresistive oscillators for mass sensing.

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Gayathri Pillai (S’16) received the B.Tech. degree in electronics and communication engineering from the School of Engineering, Cochin University of Science and Technology, India, in 2013, and the M.S. degree from the Institute of Nanoengineering and Microsystems, National Tsing Hua University, Hsinchu, Taiwan, in 2015, where she is currently pursuing the Ph.D. degree. Her M.S. research interests focused mainly on Giga Hertz FBAR resonator and off-chip oscillator designs. Her current Ph.D. interests are mainly on the design of piezoelectric micromechanical resonators and filters for wireless communication systems.

Sheng-Shian Li (S’04–M’07–SM’17) received the B.S. and M.S. degrees in mechanical engineering from National Taiwan University, Taipei, Taiwan, in 1996 and 1998, respectively, and the M.S. and Ph.D. degrees in electrical engineering and computer

science from the University of Michigan at Ann Arbor, Ann Arbor, MI, USA, in 2004 and 2007, respectively. In 2007, he joined RF Micro Devices, Greensboro, NC, USA, where he was a Research and Development Senior Design Engineer for the development of MEMS resonators and filters. In 2008, 26

he joined the Institute of NanoEngineering and MicroSystems (iNEMS), National Tsing Hua University, Hsinchu, Taiwan, where he is currently a Professor and the Director of iNEMS. His current research interests include nano-/micro-electromechanical systems, integrated resonators and sensors, RF MEMS, CMOS-MEMS technology, front-end

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communication architectures, and integrated circuit design and technology. Dr. Li was a recipient of the Teaching Excellence Award and the Young Faculty Research Award from the College of Engineering, National Tsing Hua University, in 2011 and 2012, respectively. Together with his students, he received the Best Paper Awards at the 2011 IEEE International Frequency Control Symposium, the 2012 IEEE Sensors Conference, and the 2017 Transducers Conference. In 2013, he also received the Ta-Yu Wu Memorial Award from the National Science Council, Taiwan. He served as the TPC/ETPC for the IEEE International Frequency Control Symposium, the IEEE Sensors Conference, the Transducers Conference, and the IEEE MEMS Conference. He serves as an Associate Editor for the Journal of Micro/Nanolithography, MEMS,

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and MOEMS (SPIE).

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Table I. Resonator Data Summary. 60µm-Wide CC-Beam

Parameters

20µm-Wide CC-Beam

Units

150

GPa

Density, ρ

2,300

kg/m3

Beam Length, Lr

40

μm

Beam Thickness, h

2

μm

Driving Electrode Width, Wd

4

μm

Sensing Electrode Width, Ws

20

μm

100

nm

Electrode-to-Resonator Gap, do 9

9

V

Measured Resonant Frequency, fo

9.54

9.52

MHz

635

866

―

1.797×10-11

5.992×10-12

kg

5.665×104

1.888×104

N/m

Effective Damping, creff

1.589×10-6

3.884×10-7

m2

Drive-Port Electromech. Coupling Coeff., ηd

2.045×10-8

1.181×10-8

N/V

Sense-Port Electromech. Coupling Coeff., ηs

1.520×10-3

2.926×10-4

C/m

Measured Motional Resistance, Rm

50.12

56.2

kΩ

Calculated Motional Resistance, Rm

51.11

112.43

kΩ

Calculated Motional Inductance, Lm

0.578

1.734

H

Calculated Motional Capacitance, Cm

0.549

0.183

fF

Drive-Port Static Overlap Capacitance, Cod

42.50

14.17

fF

Sense-Port Static Overlap Capacitance, Cos

106.25

35.42

fF

Calculated High-Stiffness Driving Power Handling, Pomax

-5.69 (64.5µW)

-18.02 (15.8µW)

dBm

Calculated Low-Stiffness Driving Power Handling, Pomax

-24.17 (3.8µW)

-30.29 (0.9µW)

dBm

Layout Area

360×270

100×60

μm2

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DC-Bias Voltage, VP

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Young’s Modulus, E

Effective Mass, mreff

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Effective Stiffness, kreff

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Measured Quality Factor, Q

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Parameters

Value

Units

Structure Width, wr

50

µm

Structure Length, Lr

40

µm

Coupling Location

4.48

µm

Coupling Beam Length, Lc12

21.5

µm

Coupling Beam Width, wc12

1

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Design Parameters

Table II. Design and Performance Summary of 50µm wide CC-Beam Filter.

Electrode to Resonator Gap, do

100

nm

150

GPa

2300

kg/m3

8.5

V

47.9

dB

5.736

dB

9.707

MHz

Mixler’s Combined Loss

14.4

dB

Mixer’s Conversion Loss

8.7

dB

Motional Resistance, Rm

3.27⨯10-7

Ω

Equivalent Inductance, Lm

3.647⨯10-12

H

Equivalent Capacitance, Cm

8.663⨯10-5

F

Co1=Co2

3.727

fF

Coupling Beam Capacitance, Cs12a= Cs12b

-0.0081

F

Coupling Beam Capacitance, Cs12c

0.0081

F

Electro-mechanical Coupling Coefficient, ηe

6.55⨯10-6

C/m

Transformer Turns Ratio, ηc

6.8451

C/m

Termination Resistor, RQ1

3.2

kΩ

Termination Resistor, RQ2

2.2

kΩ

Young’s Modulus, E

Biasing Voltage, VP

Measured Results

Insertion Loss, I.L. @ unterminated

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Insertion Loss, I.L. @ terminated

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Density, ρ

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Calculated Results

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Center Frequency, fo

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