A micropower amperometric potentiostat

Sensors and Actuators B 97 (2004) 284–289

A micropower amperometric potentiostat Matthew D. Steinberg∗ , Christopher R. Lowe Institute of Biotechnology, University of Cambridge, Tennis Court Road, Cambridge, CB2 1QT, UK Received 1 July 2003; received in revised form 1 July 2003; accepted 2 September 2003

Abstract A micropower amperometric potentiostat, suitable for use in cost-sensitive analytical sensors has been built and characterised in our laboratory. The low-power, low-cost and miniature design makes this potentiostat potentially attractive for use in handheld, portable and battery powered electrochemical instruments and sensors where component cost, size, or power consumption, may be at a premium. The potentiostat uses a novel analogue to digital converter (ADC) circuit that allows the direct conversion of electrode currents in the nanoampere range to low-voltage CMOS logic levels using four operational amplifiers. The digital output of the potentiostat is suitable for direct interfacing to microprocessors, removing the need for expensive external, or on-chip, ADC’s. Specific characteristics of the embodiment we have evaluated are a current consumption of 260 ␮A, operation down to 2.5 V dc, a total component volume of less than 3 cm3 , and a component cost of approximately 5 Euros for prototype quantities. © 2003 Elsevier B.V. All rights reserved. Keywords: Amperometric; Potentiostat; Instrumentation; Micropower; Microsensor; ADC

1. Introduction Potentiostats are routinely employed in the study of electrochemical systems, and in particular, often form the instrumental component of electrochemical-based detection systems for chemical and biological sensors. There are many examples of sensors that require miniature, low-cost or low-power electronics for providing the interface between the chemically- or biologically-modified electrode and, for example, downstream signal-processing devices, such as microprocessors and microcontrollers. Specific examples of such products and devices include portable blood glucose [1] and lactate meters [2], toxic gas sensors [3], portable clinical analysers [4], percutaneous [5] and implantable electrode systems [6,7] and systems for total micro analysis (␮TAS) [8,9]. In this work, we attempt to address the need for a general purpose miniature amperometric potentiostat, having the specific properties of low-power and low-voltage operation, together with a low component and manufactured cost, and which is suitable for the direct interfacing of chemical and biosensor electrodes to four or eight-bit microcontrollers of limited processing power. Over the past 15 years, a number of successful potentiostat ∗ Corresponding author. Present address: Erasmus Technology LLP, 57A Moorfield Road, Duxford, Cambridge, CB2 4PP, UK. Tel.: +44-7980-803913; fax: +44-1223-561741. E-mail address: [email protected] (M.D. Steinberg).

0925-4005/$ – see front matter © 2003 Elsevier B.V. All rights reserved. doi:10.1016/j.snb.2003.09.002

designs have been miniaturised and implemented using integrated circuit (IC) techniques [10–12]. This approach has the attraction of very small size, low-power consumption, and even provides the opportunity of including microelectrodes on-chip [13]. Microcircuit engineering is a valid approach in addressing the instrumental requirements of highly integrated or sub-miniature sensor systems, as might be required for certain ␮TAS applications [14,15] and fully implanted medical systems [16], but for the majority of macro-scale electrochemical sensor applications (electrode areas ≥ 1 mm2 ), the move to integrated silicon devices is probably unnecessary, and may also be a prohibitively expensive step to undertake. We describe here a novel miniature potentiostat, which may be constructed from low-cost surface mount components. The design may readily be adapted to suit different amperometric chemical and biosensor applications, and in terms of size and power consumption, approaches the performance achieved by some IC potentiostats.

2. Circuit description Practical circuits and detailed reviews about the operation of analogue and digital amperometric potentiostats can be found in several reference source books [17,18], in novel research work [12,13] and in scientific journals [19–21]. Our micropower amperometric potentiostat is an adaptation of

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Vdd

R3 1M0 Buffer Vset

+ U2

2-electrode cell

R4 1M0

C3 100pF R6 1M0

AGND

Icell

U3

-

+

U4

R7 1M0

DATA OUT

+ Integrator

Q2 ZVN330rF

Schmitt Trigger

AGND

Vss

R5 1M0

AGND

Fig. 1. The micropower potentiostat makes use of a current-to-frequency converter to generate a CMOS compatible pulse output, DATA OUT, that varies in frequency with the current in the working electrode circuit, Icell .

the classic analogue transimpedance potentiostat (Fig. 1). We have substituted the transimpedance element, which is often purely resistive, for a capacitor, and additionally, have closed a feedback loop around the transimpedance stage to create a current controlled oscillator (CCO). The potentiostat works as follows: the potential of the working electrode with respect to the counter-reference electrode is set at the input of U2 by potential divider circuit R3R4. The output of U2 drives the counter electrode of the cell. The working electrode is connected at the inverting terminal of integrating amplifier U3, and is thus held at a potential near to analogue ground (AGND). Current flowing into the working electrode circuit charges capacitor C3, and as feedback capacitor C3 charges, the output voltage of amplifier U3 changes at a rate proportional to the magnitude of the working electrode current. Amplifier U4 is a Schmitt trigger, which toggles output state as the output of U3 reaches the threshold set by resistor network R5R6. With a high output state, U4 will turn on MOSFET Q2, thereby discharging capacitor C3 through R7 to Vss . As C3 discharges, the output of integrator U3 is reset, causing the output state of U4 to

return low. This removes gate drive from Q2, which allows the working electrode current to flow into C3 once again, thus beginning the oscillatory cycle over again. The rate at which the cycle repeats is determined by the magnitude of the working electrode current into C3, the capacitance of C3, and the time-constant of network R7C3. Since C3 and R7 are nominally constants, the frequency of the digital output signal, DATA OUT, is proportional to the current in the working electrode. The virtual analogue ground reference potential, AGND, is midway between Vdd and Vss , and is driven from the output of a buffer amplifier and potential divider circuit connected across the battery terminals (not shown).

3. Materials and methods The potentiostat circuit was designed rationally using conventional analogue circuit techniques, with due consideration devoted to the requirements for low-power consumption, low-voltage operation, and a minimum number

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of component parts. Printed circuit layouts were designed and drawn with commercial computer aided design (CAD) software (EASYPC Professional, Number One Systems Ltd., Huntingdon, Cambs., UK) running on a personal computer. Printed circuit boards were fabricated by a commercial manufacturer (RAK Printed Circuits Ltd., Saffron Walden, Essex, UK). All printed circuits were produced on 0.4 mm glass fibre board, clad with 30 oz/m2 copper. The 15 mm diameter double sided boards contained tin–lead alloy plated through hole vias, and were coated in a solder resist layer. Surface mount components for the potentiostat were obtained from a commercial electronics supplier (RS Components Ltd., Corby, Northants, UK) and manually soldered to the printed circuit boards. Assembled potentiostats were tested, and the following parameters experimentally determined: Current-to-frequency calibration, power consumption, voltage-clamp stability, electrode noise. All experimental measurements were made using certified calibrated laboratory instruments, comprising a digital multimeter/frequency counter (Philips, PM2525), a digital storage oscilloscope (Philips, PM3320A), a digital spectrum analyser (Hewlett Packard, HP4195A) and a multi-component bridge (Wayne Kerr, 6425).

4. Results 4.1. Calibration curve The current-to-frequency calibration curve obtained for the potentiostat is shown (Fig. 2). Different currents were sequentially set with high-ohmic thick-film resistors connected across the working and counter electrode terminals of the potentiostat. In each case, the current value was calculated from the ratio of the measured electrode potential difference to the measured value of the applied resistance.

The current-to-frequency transfer function is linear between approximately 4 and 150 nA, and extends up to currents of 625 nA. Although the response is non-linear above 150 nA, the calibration curves were consistent and reproducible over the full range tested. In the linear region for the circuit shown (Fig. 1), the output frequency was found by least squares fit to relate to cell current by the relationship Fout = 3.15 Icell + 2.15, where Icell is the electrode current in nA. The zero offset of the converter is 2.15 nA, and the sensitivity is 3.15 Hz/nA. 4.2. Randall electrode equivalent circuit The current-to-frequency calibration curve (Fig. 2) was based on connecting the potentiostat electrodes to resistive loads. A more reasonable approach is to use an equivalent circuit model of an electrode, that more closely accounts for cell capacitance and the effects of non-Faradaic charging at the electrode–electrolyte interface [22]. These phenomena can be approximated with a lumped equivalent circuit model, consisting of a resistance and parallel capacitance. The current-to-frequency calibration curve changes dramatically under capacitive loading, as shown (Fig. 3). For cell capacitances less than 100 pF or greater than 5000 pF, the calibration curve is flat (linear). However, at intermediate cell capacities, a non-linear region is encountered. By inspection of the new calibration curve (Fig. 3), shunting the electrode terminals with a 100 nF capacitor restores linear calibration, and increases the current sensitivity of the potentiostat from 3 to 7 Hz/nA. 4.3. Power consumption The current required to operate the potentiostat was determined experimentally as 260 ␮A, with a Vdd to Vss potential difference of 2.5 V. This yields a calculated power

Fig. 2. The current-to-frequency calibration curve of the micropower potentiostat with resistive loading of the electrode terminals. (a) Linear current range, 0–150 nA and (b) full current range, 0–625 nA.

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Fou t (Hz) 5000 4500 4000 3500 3000 2500 2000 1500 1000 500 0 1

10

100

1000

10000

100000

Cell cap acitan ce (p F) Rcell=100MOhm s

Rcell=10MOhm s

Rcell=1MOhm

Fig. 3. The current-to-frequency calibration curve of the micropower potentiostat with complex loading (parallel resistive and capacitive) of the electrode terminals.

consumption of 650 ␮W. Loading the electrode terminals with different complex impedances had no effect on overall power consumption. 4.4. Voltage-clamp regulation The following test was performed to determine the ability of the potentiostat to regulate the electrode potential difference. The mean dc cell potential was measured whilst the electrode terminals were loaded with impedances of 1, 10 or 100 M resistance and of capacitances between zero and 100,000 pF. The resulting cell potential as a function of complex cell impedance was found to be 634 + 9/ − 3 mV (mean, maximum, minimum), a regulation of better than 2% (Fig. 4). 4.5. Electrode noise The electrode terminals were loaded with a parallel impedance of 1 M and 10 nF. Spectral analysis was performed to investigate the harmonic content of the clamp voltage. The spectra thus obtained are shown (Fig. 5).

5. Discussion The micropower potentiostat exhibits good linearity in the current range up to 150 nA when calibrated with highly stable thick-film resistors, with a detection limit of approxi-

mately 2 nA. In the embodiment tested, a sensitivity of just over 3 Hz/nA was achieved. Theoretically, by altering the value of C3, the current sensitivity of the potentiostat may be either increased (smaller C3 value) or decreased (larger C3 value). However, with complex loads applied at the electrode terminals, comprising resistive and capacitive components, the calibration curve of the potentiostat was found to exhibit a non-linear characteristic in the capacitive load range of 100–5000 pF. Practically, this effect was overcome by shunting the electrode terminals with a 100 nF capacitor. Interestingly, this too had the additional effect of increasing the current sensitivity of the potentiostat from 3 to 7 Hz/nA. The potentiostat was able to control the working electrode potential with respect to the counter electrode with a regulation of better than ±2% under various complex load impedances. The circuit consumes 650 ␮W of power, achieved through the careful use of low-power CMOS amplifiers in the circuit design, and minimal quiescent current drain, and is therefore suitable for operation from standard 3 V Lithium coin cells. The general approach taken in the design of the potentiostat means that the embodiment presented here may be easily adapted for different sensor applications. For example, use of three electrodes is possible, for sensors requiring separate counter and reference electrodes, by breaking the feedback loop around amplifier U2, and using the inverting terminal of the amplifier for the reference electrode input. The working electrode clamp potential, Vset , may be varied by altering potential divider ratio R3R4, as required, and may be of negative polarity with respect to the

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645

640

Cell Potential (mv)

635

630

625

620

615

610

605

600 1

10

100

1000

10000

100000

1000000

Cell Capacitance (pF) PSU Rcell=100M

PSU Rcell=10M

PSU Rcell=1M0

Fig. 4. The electrode clamp (cell) potential as a function of applied complex load impedance.

Fig. 5. Signal power spectra measured at the electrode input terminals of the micropower potentiostat (a) in the low frequency band 0.001–10 Hz, and (b) across a wideband range of 1–25 kHz. The y-axis scale of both graphs extends from −100 to 0 dBm, in steps of 10 dBm.

reference electrode by tying R4 to Vss instead of AGND, as we have done.

6. Conclusions A novel current-to-frequency converter circuit has been designed and developed by us for implementing the function of an amperometric potentiostat with a minimum number of components. The specific unique qualities of this current-

to-frequency converter that make it particularly suited to this type of application are high input impedance, necessary to obtain accurate amperometric measurements, and the direct production of a pulsed digital output, compatible with lowvoltage CMOS logic gates, using only a few components. The converter exhibits good linearity in the nanoampere current range that is relevant for many chemical and biosensor electrodes. This particular design has a very low-power consumption, and may easily be adapted to meet the requirements of different electrode-based amperometric sensors.

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Acknowledgements This work was made possible through the financial support of the Biotechnology and Biological Sciences Research Council (BBSRC) in the U.K., who funded a graduate studentship at the Institute of Biotechnology for Matthew Steinberg.

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Biographies Matthew D. Steinberg received a BEng degree in electronic engineering from the University of Liverpool in 1988, and a PhD in biotechnology from the Institute of Biotechnology, at the University of Cambridge, in 1998. His career in biomedical engineering and biotechnology includes research conducted at St. Bartholomew’s Hospital on the peripheral circulatory system, and development of sensor-based instrumentation for microgravity space research at Brunel University, London. Between 1997 and 2002, he worked for Europe’s leading independent development company in Cambridge, where he specialised in medical, biotechnology and healthcare product and new business development. In 2002, he co-founded a science and technology consulting partnership, in which he is managing partner. Christopher R. Lowe received his BSc and PhD degrees in biochemistry from the University of Birmingham in 1967 and 1970, respectively. He has conducted post-doctoral research in Liverpool and Sweden and held a lectureship/senior lectureship at the University of Southampton. He is currently Director of the Institute of Biotechnology and professor of biotechnology at the University of Cambridge. He is a fellow of Trinity College. The principal focus of his biotechnology research programme over the last 30 years has been the high value-low volume sectors of pharmaceuticals, fine chemicals and diagnostics. The work not only covers aspects of biochemistry, microbiology, chemistry, electrochemistry, physics, electronics and chemical engineering, but also the entire range from pure science to strategic applied science, some of which has significant commercial applications and had led to the establishment of seven spin-out companies. He has 250 publications, 7 monographs and 40 patents. He is actively involved in many collaborations worldwide, is on the editorial boards of a number of academic journals, sits on a number of research council, grant awarding and government committees, and is active in various legal and entrepreneurial roles. He has supervised over 50 PhD students and won a number of prizes over the last two decades: the Pierce Award for Outstanding Contributions to the Field of Affinity Chromatography and Related Techniques (1989); David Curnow Prize in Clinical Chemistry for work on Biosensors (1991); Schlumberger Stichting Prize (1994); The Queen’s Award for Technological Achievement (1996); The Jubilee Medal of the Chromatographic Society (2002); Elected Russian Academy of Medical Sciences (2002).