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Bandpass filter based on asymmetric funnel shaped resonators with ultra wide upper stopband characteristics
Journal Pre-proofs Regular paper Bandpass filter based on asymmetric funnel shaped resonators with ultra wide upper stopband characteristics Ami Iqubal, P. Abdulla PII: DOI: Reference:
S1434-8411(19)30780-0 https://doi.org/10.1016/j.aeue.2020.153062 AEUE 153062
To appear in:
International Journal of Electronics and Communications
Received Date: Revised Date: Accepted Date:
3 April 2019 29 December 2019 1 January 2020
Please cite this article as: A. Iqubal, P. Abdulla, Bandpass filter based on asymmetric funnel shaped resonators with ultra wide upper stopband characteristics, International Journal of Electronics and Communications (2020), doi: https://doi.org/10.1016/j.aeue.2020.153062
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Title: Bandpass Filter Based on Asymmetric Funnel Shaped Resonators with Ultra Wide Upper Stopband Characteristics Authors: 1.
Ami Iqubal (Degree: pursuing Ph. D), email id: [email protected]
2.
Abdulla Parambil (Degree: Ph. D), email id: [email protected]
Affiliation: Division of Electronics, School of Engineering, Cochin University of Science and Technology, Kerala, India Corresponding Author: Ami Iqubal
BANDPASS FILTER BASED ON ASYMMETRIC FUNNEL SHAPED RESONATORS WITH ULTRA WIDE UPPER STOPBAND CHARACTERISTICS Ami Iqubal and Abdulla P. Division of Electronics, School of Engineering, CUSAT, Cochin-22, Kerala, India. Corresponding author: [email protected] Abstract — A novel microstrip bandpass filter with wide passband, good skirt selectivity and very good passband reflection characteristics using asymmetric funnel shaped resonators is presented in this paper. IE3D simulation software is used for the design and analysis of the proposed filter and the electric field intensities at various frequencies of interest are plotted and verified the role of even and odd mode resonances in achieving wide passband. There are six modes, including three even and three odd modes for the developed filter. Novel equations are derived for finding out the location of transmission zeros and the role of resonators in achieving sharp selectivity are verified. The out of band performance of filter is very good with a suppression level greater than 20 dB upto 24.2 GHz. The centre frequency of the filter is 5.68 GHz with a fractional bandwidth of 97.57 %. The fabrication process is carried out using MITS 11 Autolab fabrication machine with the low cost FR4 substrate and verified the performance improvement of the developed filter. Keywords – Bandpass Filter (BPF); Selectivity; Stopband; Transmission Zero; Wide Fractional Bandwidth
1. INTRODUCTION Wideband bandpass filters are the key components in wideband communication systems for removing the unwanted signals and noise over a large bandwidth. The main design considerations for developing an ideal bandpass filter are low cost, good in and out-of-band performance, sharp selectivity of the passband, very good relative stopband bandwidth and simple design procedures. Federal Communications Committee (FCC) in February 2002 granted the unlicensed use of ultra wideband (UWB) communications spectrum from 3.1 GHz to 10.6 GHz. Wide arrays of techniques were investigated to realize bandpass filter configurations over the last decade [2-15]. A wideband bandpass filter developed using fish spear shaped multiple mode resonator [2] wherein the transmission zeros were absent at the passband edges and hence the passband to stopband transition is very poor. Filter design using inductive via series [3] exhibits very good performance except its poor roll-off. Bandpass filter with cascaded linked hexagonal omega resonators [4] suffers from very small stopband bandwidth. Wideband filters consisting of three coupled lines including the cross coupling between adjacent resonators were discussed in [5]. The roll–off and relative stopband bandwidth shows very poor performance. The filters developed in [6-7] fail to achieve very good spurious suppression. UWB filter based on asymmetric parallel-coupled line (APCL) is discussed in [8], but it achieves stopband suppression only upto 12.8 GHz. Wideband bandpass filter design using complementary split ring resonators (CSRR) is developed in [9]. However maximum stopband achievable is very low for this filter. In [10], a folded multiple mode resonator is employed for the development of wideband bandpass filters. Even though a passband with wide bandwidth is obtained, the reflection characteristics of the filter show poor performance. Various filter topologies including meta material resonators [11], and double T- shaped resonators [14] suffers from very low upper outband skirt rejection
performance. UWB bandpass filter based on an objective function [12], structure based on microstrip to CPW transition [15] and filter using stub loaded multiple mode resonator [19] achieves upper stopband suppression only less than 18 GHz. The motivation of this paper is to develop an easily implementable wideband bandpass filter structure with very good stopband to passband transition and ultra wide upper stopband characteristics using a very low cost easily available FR4 substrate material with a thickness of 0.8 mm and a dielectric constant of 4.4. The design procedures and equations for finding out the transmission zero locations are explained in detail. Two high impedance stubs loaded symmetrically to the upper funnel shaped resonator played a great role in obtaining passband return loss characteristics better than 20 dB. Electric field intensities of various frequencies of interest are analyzed using IE3D simulation software and verified the role of even and odd mode resonances in achieving wide passband. MITS 11 Autolab fabrication machine is used for the prototype fabrication. The measurements were carried out using R&S ZVB 20 vector network analyzer and verified that the in and out-of-band performance of the developed filter is very good.
2. THE WIDEBAND BANDPASS FILTER DESIGN The bandpass filter having the desired wideband characteristics is designed as follows. 2.1.
THE DESIGN AND ANALYSIS OF THE BASIC FUNNEL SHAPED RESONATOR The filter design begins by allocating transmission zeros at lower and upper band edges. A
funnel shaped resonator consisting of a rectangular stub and a tapered portion is used for achieving sharp selectivity because the linear tapering provides better reflection characteristics since the
transition from high impedance line to low impedance line is gradual. The structure and equivalent circuit of the basic funnel shaped resonator is given in Fig. 1 (a) and (b) respectively.
Fig. 1. (a) Geometry (b) LC equivalent circuit
The structural dimensions of the funnel shaped resonator are as follows. The tapered section is having a width of Wuf = 4.78 mm, a length of Luf = 3.41 mm and an impedance of ZB. The rectangular region is characterized with a length of Llf = 9.15 mm, a width of Wlf = 1.1 mm and an impedance of ZA. The basic resonator is connected to a high impedance line of characteristic impedance ZL. The funnel shaped resonator can be approximately modeled as a series and shunt pairs of inductors and capacitors La, Ca and Lb, Cb respectively. Port 1 and Port 2 of the resonator are terminated with transmission lines of 50 Ω characteristic impedance having a width 1.5 mm for ensuring proper impedance matching since here we are using FR4 substrate with dielectric constant, εr = 4.4, thickness, h = 0.8 mm and a loss tangent of 0.02. The characteristics of the funnel shaped resonator is calculated by using the basic equations for εre and ZC [1, 16], where εre and ZC are the effective dielectric constant and the characteristic impedance of the lines respectively. The calculated values of ZA = 60.29 Ω and ZB = 22.52 Ω. The effective dielectric constant of rectangular region of funnel shaped resonator, εreA= 3.245 and the effective dielectric
constant of the tapered portion of the funnel shaped resonator, εreB = 3.68. The values of series and parallel combinations of the inductors and capacitors can be calculated by using the equations (1) and (2), where ‘c’ is the velocity of light in free space.
𝐿𝑎 =
𝐶𝑎 =
𝑍𝐴 𝜀𝑟𝑒𝐴
𝑍𝐵 𝜀𝑟𝑒𝐵
𝑐
𝑐
𝐿𝑙𝑓, 𝐿𝑏 =
𝐿𝑙𝑓 𝜀𝑟𝑒𝐴
𝐿𝑢𝑓 𝜀𝑟𝑒𝐵
𝑐𝑍𝐴
𝑐𝑍𝐵
, 𝐶𝑏 =
(1)
𝐿𝑢𝑓
(2)
The equivalent values of inductors and capacitors are obtained as follows. La = 3.312 nH, Lb=0.49 nH, Ca = 0.911 pF and Cb = 0.96 pF. The equivalent impedance of the lower rectangular portion of the funnel shaped resonator can be represented as a series combination of La and Ca and is given in equation (3). The equivalent impedance of the tapered region of the funnel shaped resonator is represented as the parallel combination of impedances Lb and Cb and is given in equation (4).
𝑍𝐴 =
1 ― 𝜔2𝐿𝑎𝐶𝑎
(3)
𝑗𝜔𝐶𝑎
𝑗𝜔𝐿𝑏
(4)
𝑍𝐵 = 1 ― 𝜔2𝐿 𝐶
𝑏 𝑏
The ABCD parameters of the funnel shaped resonator are given in the equations (5 - 7). 𝜔2𝐿𝐶𝑎(1 ― 𝜔2𝐿𝑏𝐶𝑏)
𝐴 = 𝐷 = 1 ― (1 ― 𝜔2𝐿 𝐶 )(1 ― 𝜔2𝐿 𝐶 ) ― 𝜔2𝐿 𝐶 𝑎 𝑎
𝑏 𝑏
𝑏 𝑎
𝑗𝜔3𝐿2𝐶𝑎(1 ― 𝜔2𝐿𝑏𝐶𝑏)
𝐵 = 2𝑗𝜔𝐿 ― (1 ― 𝜔2𝐿 𝐶 )(1 ― 𝜔2𝐿 𝐶 ) ― 𝜔2𝐿 𝐶 𝑎 𝑎
𝑏 𝑏
𝑗𝜔𝐶𝑎(1 ― 𝜔2𝐿𝑏𝐶𝑏)
𝐶 = (1 ― 𝜔2𝐿 𝐶 )(1 ― 𝜔2𝐿 𝐶 ) ― 𝜔2𝐿 𝐶 𝑎 𝑎
𝑏 𝑏
𝑏 𝑎
𝑏 𝑎
(5)
(6)
(7)
By substituting the values of A, B, C and D, we can find out the transmission and reflection characteristics S11 and S12. The transmission and reflection characteristics of the main resonator implies that the series inductor – capacitor pair [La, Ca] and the shunt inductor –capacitor pair [Lb, Cb] are responsible for the filter response. By varying the values of La, Lb, Ca and Cb, we can control the position of transmission zeros. By equating the value of transmission characteristics to zero, the lower and upper finite frequency attenuation poles at locations fz1 and fz2 can be determined as given in equations (8 – 11).
[𝜔4(𝐿𝑎𝐿𝑏𝐶𝑎𝐶𝑏) ― 𝜔2(𝐿𝑎𝐶𝑎 + 𝐿𝑏𝐶𝑏 + 𝐿𝑏𝐶𝑎) + 1] = 0
(8)
We can find out the resonant frequencies corresponding to the lower and upper passband transmission zeros by solving for 𝜔4 in the above equation (8) using the basic equation for finding out the roots of a quadratic equation and ω can be determined as given below. 𝜔= ±
∆1 ± ∆12 ― 4∆2
(9)
2∆2
where, ∆1 = 𝐿𝑎𝐶𝑎 + 𝐿𝑏𝐶𝑏 + 𝐿𝑏𝐶𝑎 , and ∆2 = 𝐿𝑎𝐿𝑏𝐶𝑎𝐶𝑏 The frequencies corresponding to the transmission zero locations are obtained as, 1
fz1 = 2𝜋
and,
1
fz2 = 2𝜋
∆1 + ∆12 ― 4∆2 2∆2
∆1 ― ∆12 ― 4∆2 2∆2
(10)
(11)
0
S-parameters (dB)
-5 -10 -15 -20 -25 -30
S 11
-35
S 12
-40 0
2
4
6
8
Frequency (GHz)
10
12
14
Fig. 2. Simulated frequency response of the basic funnel shaped resonator
Even though the funnel shaped resonator exhibits sharp skirt rejection and very good passband response, the resonator fails to suppress the spurious harmonics of the lower and upper stopband regions. The values of ∆1= 5.1105112 X 10-21 and ∆2= 1.4193 X 10-42 are obtained by substituting the values of La,Ca, Lb and Cb. The calculated values of resonant frequencies are fz1= 2.29 GHz and, fz2= 9.27 GHz obtained from equations (10) and (11). Fig. 2 shows the IE3D software simulated frequency response of the upper funnel shaped resonator having fz1 = 2.6 GHz and fz2 = 9.1 GHz which is in good accordance with the theory. Slight discrepancy in the theoretical and simulated values of transmission zeros are because of the fact that the dielectric loss of the substrate used is not considered in theoretical analysis. The attenuation zero of the resonator is located at 7.1 GHz with an attenuation level of -34 dB.
2.2.
ODD AND EVEN MODE ANALYSIS OF THE BASIC UPPER FUNNEL SHAPED FILTER STRUCTURE The basic structure of the upper funnel shaped resonator showing the plane of symmetry
A-A’ is shown in Fig. 3 and the equivalent transmission model of the funnel shaped resonator is given in Fig. 4.
Fig. 3. Basic upper funnel shaped resonator with plane of symmetry A-A'
Considering the plane A-A', the resonator is symmetrical in nature and the odd and even mode method can be applied for analysing it.
Fig. 4. Equivalent transmission line model of the funnel shaped resonator
When odd-mode excitation is applied, the potential difference along the symmetrical plane A-A' is zero and the central point of the resonator is short circuited. By neglecting the input and output port impedance, the odd mode equivalent transmission model is as shown below in Fig. 5.
Fig. 5. Odd mode equivalent circuit of the basic funnel shaped resonator
―𝑗
The input admittance for the odd mode, Yin, odd of the resonator = 𝑍1 cot 𝜃1
(12)
Thus the odd modes are fixed and do not depend on the upper funnel shaped resonator. Odd mode resonances can be calculated by equating Yin, odd = 0. Under even mode of excitation, the symmetrical plane A-A' should be open circuited as given in the references [17] and [18]. The even mode equivalent model of the basic upper funnel shaped resonator is shown in Fig. 6.
Fig. 6. Even mode equivalent circuit of the basic funnel shaped resonator tan 𝜃1 𝑍1
+∅
The input admittance for the even mode, Yin, even of the resonator = 1 ― tan 𝜃1𝑍1∅ where ∅ =
Yin, even =
tan 𝜃3 𝑍3
+
tan 𝜃2 𝑍2
𝑍1𝑍2tan 𝜃3 + 𝑍1𝑍3tan 𝜃2 + 𝑍2𝑍3tan 𝜃1 𝑍2𝑍3 ― 𝑍1 tan 𝜃1 tan 𝜃2 tan 𝜃3
(13) (14)
(15)
The even mode frequencies can be controlled by the funnel shaped resonator and the even mode resonances can be calculated by equating Yin, even = 0.
In order to improve the characteristics of the funnel shaped resonator, the spurious bands in the lower and upper stopband regions must be suppressed. For achieving a very good bandpass filter response with sharp outband spurious suppression characteristics, the basic funnel shaped resonator is loaded with electronic band gap (EBG) based suppression cells with proper coupling as described in the section 2.3. 2.3.
DESIGN OF EBG BASED SUPPRESSION CELLS For suppressing harmonics, four band gap structures having a width, We = 1 mm, Le = 2.45
mm are loaded symmetrically in side locations with respect to the central transmission line with a lateral spacing of Lwe= 3.64 mm. The basic structure and the frequency response of the EBG are given in Fig. 7 (a) and (b). The simulated frequency response shows that a filtering response is obtained by the loading of Electronic band gap structures to the main transmission line with a 3 dB cut-off frequency of 7.8 GHz. The EBG loaded filter structure exhibits the rejection of higher order harmonics as evident from the obtained filter response.
S -Parameters (dB)
0 -10 -20 -30 S11
-40
S12 -50 0
(a)
2
4
6
8
10
12
14
Frequency (GHz)
16
18
(b)
Fig. 7. (a) Basic EBG loaded structure (b) Frequency response of the EBG loaded structure
20
The wide stopband of the EBG loaded structure is used to suppress the higher order harmonics of the bandpass filter and thus it proves to be an ideal candidate for the rejection of higher order harmonics for the funnel shaped bandpass filter. The frequency response of the EBG loaded funnel shaped resonator is explained in the section 2.4. 2.4.
EBG LOADED FUNNEL SHAPED RESONATOR The electronic band gap loaded funnel shaped resonator structure and its frequency
response is given in Fig. 8 (a) and (b). Four identical EBG structures are placed symmetrically to the basic funnel shaped resonator for suppressing harmonics in the upper stopband region. The transmission zeros are located at 2.6 GHz and 9.1 GHz. The attenuation zeros are located at 0.1 GHz with a suppression level of -36.39 dB, at 4.925 GHz with a suppression level of -33.17 dB, at 7.337 GHz with a suppression level of -32 dB and at 8.543 GHz with a suppression level of 17.05 dB. Thus the EBG loaded funnel shaped resonator is having two transmission zeros at the lower and upper passband edges and four attenuation zeros.
S - Parameters (dB)
0
-10
-20
-30
S11 S12
-40 0
2
4
6
8
10
12
14
16
18
20
Frequency (GHz) (a)
(b)
Fig. 8. (a) EBG loaded funnel structure (b) response of the EBG loaded funnel structure
Three attenuation zeros are located at the passband region and one at lower stopband. The lower attenuation zero at 0.1 GHz must be shifted to the passband region for the suppression of lower harmonics in order to reject spurious responses at the lower stopband. 2.5.
DESIGN AND ANALYSIS OF THE GROUND PLANE APERTURE BASED TWO STAGE THREE LINE PARALLEL COUPLED RESONATORS For shifting the attenuation zeros present at the lower stopband, proper input output
coupling must be employed with tight coupling [5]. Comparing the conventional bandpass filter development methods like end-coupled and parallel coupled lines, the coupling intensity is more for parallel coupled lines [1]. The coupling intensity is limited by the tolerance introduced by the currently available fabrication methods. The minimum coupling gap that can be developed practically is only 0.2 mm in our laboratory. By keeping the minimum achievable coupling gap, an aperture is cut in to the ground plane exactly beneath the parallel coupled lines for attaining wide bandwidth. Two stage ground plane aperture based three line parallel coupled lines is introduced for shifting the lower stopband attenuation zeros to the passband. The basic layout of the coupling structure is shown in Fig. 9, black region shows the top view of the two stage three line coupled filter and the grey region represents the bottom view of the coupling structure.
Fig. 9. Layout of the basic coupling structure
The guided wavelength (λg) corresponding to the centre frequency of the proposed filter is 29 mm. The three line parallel coupled structures must be selected in such a way that the coupling length must be in the range of λg/4 for ensuring wide bandwidth. An optimized value of 7.67 mm is used in our design. The width of the three line parallel coupled structures are selected in order to accommodate three lines within 1.5 mm, the width of the input output port for ensuring impedance matching and by maintaining the minimum gap between lines as 0.2 mm. The dimensions of the basic structure are given as: L1 = 22.02 mm, W1= 0.5 mm, La= 7.67 mm, Wa= 2.4 mm and W2 = 0.3 mm.The frequency response of the filter is shown in Fig. 10.
S - Parameters (dB)
0 -10 -20 -30 -40
S11 S12
-50 0
2
4
6
8
10
12
14
16
18
20
Frequency (GHz) Fig. 10. Simulated response of the basic coupling structure for the range (0-20 GHz)
The reflection characteristics exhibits three attenuation zeros and are located at 3.6 GHz with - 28.9 dB suppression level, 8.4 GHz at -25.5 dB and at 16.38 GHz with -35.3 dB suppression level. From the frequency response, it is evident that the ground plane aperture based three line input output coupling is an ideal candidate for providing proper coupling. 2.6.
CASCADED STRUCTURE OF GROUND PLANE APERTURE BASED LINES AND EBG LOADED FUNNEL SHAPED RESONATORS For improving the out of band spurious rejection of the filter, three line parallel coupling
with proper ground plane aperture is fed to the EBG loaded funnel shaped resonator. The modified filter structure after the introduction of coupling is given in Fig. 11 (a). The frequency response of the coupled EBG loaded funnel shaped resonator for the frequency range from 0 to 20 GHz is given in Fig. 11 (b).
0
S - Parameters (dB)
-10 -20 -30 -40
S11 S12
-50 -60 0
2
4
6
8 10 12 14 Frequency (GHz)
(a)
16
18
20
(b)
Fig. 11. (a) Layout of the coupled EBG loaded funnel structure (b) Frequency response
The frequency response exhibits four attenuation zeros within the passband of the filter and are located at 3.3 GHz with -39 dB suppression level, at 5.29 GHz with -23 dB suppression level, at 7.7 GHz with -32 dB suppression level and at 8.4 GHz with -35 dB attenuation level. 2.7.
CASCADED STRUCTURE WITH FUNNEL SHAPED ATTENUATION CELL The structure and response of the suppressing cell are shown in Fig. 12 (a) and (b). The
dimensions of the attenuation cell are: Lfl = 0.73 mm, Wfl = 0.36 mm, Wbl= 1.4 mm and Lbl = 1.25 mm. The transmission zero at 15.8 GHz makes this suppressing cell a promising candidate for the suppression of spurious harmonics in the frequency range of interest.
S - Parameters (dB)
0
-10
-20
-30 S11 S12
-40 0
2
4
6
8
10
12
14
16
18
20
Frequency (GHz) (a)
(b) Fig. 12. (a) Layout (b) Frequency response
0
S - Parameters (dB)
-10 -20 -30 -40 -50
S11 S12
-60 0
(a)
2
4
6
8 10 12 14 Frequency (GHz)
16
18
20
(b)
Fig. 13. (a) Layout (b) Frequency response of the double funnel shaped stub loaded filter
The lower funnel shaped resonator which is used as the suppressing cell structure as shown in Fig. 13. (a) provides sufficient attenuation at higher frequencies and it contributes to the wide upper stopband rejection of the proposed filter structure. The attenuation zero of the lower funnel shaped resonator which is located at 8.04 GHz with -36.95 dB attenuation level interacts with the
transmission poles of the coupled ground plane aperture based EBG loaded upper funnel shaped resonator and as a result, the modified filter exhibits attenuation zeros at 3.03 GHz, 3.49 GHz, 4.62 GHz, 5.92 GHz, 7.43 GHz and at 8.12 GHz. The transmission zeros of the loaded structure are located at 2.6 GHz with -30.1 dB suppression level and at 9.1 GHz with -20.3 dB suppression level as shown in Fig. 13. (b). For improving the suppression level of upper cut off frequency and for further improving the return loss characteristics of the proposed filter, two ‘I’ shaped stubs are loaded symmetrically with respect to the funnel shaped resonators as discussed in the section 2.8. 2.8.
EFFORTS FOR IMPROVING THE BANDPASS FILTER CHARACTERISTICS The symmetrically placed ‘I’ shaped resonators interacts with the central funnel shaped
resonator and as a result, the upper transmission zero which is located at 9.1 GHz having the suppression level of 20.3 dB gets shifted into 23.1 dB and thus the selectivity of the filter got improved. Two ‘I’ shaped stubs are loaded symmetrically about the upper funnel shaped stub with a slot width, Si= 0.2 mm. The dimensions of the ‘I’ shaped stub is designed with a strip width, Wi = 0.3 mm and a strip length, Li = 3.25 mm. With the introduction of ‘ I’ shaped stubs, the positions of attenuation zeros are located at 3.03 GHz, 3.49 GHz, 4.62 GHz, 5.92 GHz, 7.43 GHz and at 8.12 GHz.
S - Parameters (dB)
0 -10 S11 S12
-20 -30 -40 -50 0
`
2
4
6
8
10 12 14 16 18 20 22 24 Frequency (GHz)
(a)
(b)
Fig. 14.(a)Layout of the final structure (b) Simulated response for the range (0-25 GHz)
0 Transmission characteristics (dB)
Transmission characteristics (dB)
0 -10 -20 -30 -40 -50 -60 -70
without coupling with tight coupling
-80 -90 2
4
6 8 Frequency (GHz)
(a)
10
12
-20 -30 -40
with gpa
-50
without gpa
-60
-100 0
-10
14
0
2
4
6 8 10 Frequency (GHz)
12
14
(b)
Fig. 15. Response (a) with and without coupling (b) with and without aperture on ground (gpa)
The ‘I’ shaped stub loaded structure and the frequency response is given in Fig. 14 (a) and (b) respectively. The coupling plays an important role in achieving wide passband characteristics. The funnel shaped main resonator without ground plane aperture based parallel coupling shows a very good passband response but fails to meet the stopband requirements. For suppressing the spurious harmonics at lower band edge of the filter, the coupling based on two stage three line parallel coupled lines with ground plane aperture is used. Even though, the input output coupling is used, the filter exhibits desired characteristics only when the coupling applied is optimum. The filter response with and without proper input-output coupling is given in Fig. 15 (a). The dotted line shows the filter response under zero coupling and the solid line shows the filter response under strong coupling. Under zero and strong coupling the transmission zero frequency locations remains unchanged because of the upper funnel shaped resonator which is the contributing element for the transmission zeros at 2.6 GHz and 9.1 GHz. Under zero coupling, the resonator exhibits resonant frequencies at 3.01 GHz, 4.02 GHz, 5.92 GHz, 7.73 GHz and at 8.74 GHz. The suppression levels
corresponding to the various frequency modes under zero coupling are at -40.3 dB, -21.1 dB, -16.6 dB, -19.24 dB and at -34.1 dB. As the optimum input output coupling with ground plane aperture based parallel coupling is applied, the transmission characteristics moves upward and forms a very good wideband filter response. Even though, the two stage three line parallel coupling is used for coupling, the dimensions of the aperture plays a very important role in achieving spurious free passband as given in Fig. 15 (b). Without aperture on ground plane (gpa), the passband shows fluctuations and the filtering performance is not linear.
3. THE STUDY OF FIELD INTENSITY CHARACTERISTICS FOR VARIOUS FREQUENCIES OF INTEREST For verifying the correlation between the simulated results and the field behavior of the proposed filter, a study of field intensities is carried out using simulation software IE3D and is given in Fig. 16 (a) – (k).
Fig. 16. Electric field intensities at: (a) upper stopband @ 15 GHz (b) lower stopband @ 1 GHz (c) transmission zero 1 @ 2.6 GHz (d) transmission zero 2 @ 9.1 GHz (e) centre frequency @ 5.68 GHz (f) mode 1 @ 3.03 GHz (g) mode 2 @ 3.49 GHz (h) mode 3 @ 4.62 GHz (i) mode 4 @ 5.92 GHz (j) mode 5 @ 7.43 GHz (k) mode 6 @ 8.12 GHz
The various frequencies of interest includes lower stopband, upper stopband, centre frequency and the multiple modes in the passband. The frequencies upto the lower cut-off frequency are the lower stopband and the range of frequencies above the upper cut-off frequency are termed as upper stopband. The electric field intensity at 1 GHz and 15 GHz are taken here for visualizing the field intensities at lower and upper stopband ranges respectively as shown in Fig.
16 (a) and (b). At stopband, the field distribution shows that almost all energy is locked near the input port and it is not coupled to the output. The electric field distribution at transmission zeros 1 and 2, that is at 2.6 GHz and 9.1 GHz shows that the signal from input port is not reaching the output port and it is crowded near the upper funnel shaped resonator as given in Fig. 16 (c) and (d). The electric field intensity at the passband centre frequency 5.68 GHz is plotted in Fig. 16 (e) and it implies that full signal power is transmitted from port 1 to port 2. The electric field distribution at various transmission poles are analyzed for determining the even and odd mode frequencies in Fig. 16 (f – k). The transmission poles are known as the modes and there are 6 modes at frequencies 3.03 GHz (mode 1), 3.49 GHz (mode 2), 4.62 GHz (mode 3), 5.92 GHz (mode 4), 7.43 GHz (mode 5) and at 8.12 GHz (mode 6). The electric field distribution analysis shows that among the six modes, there are three even modes and three odd modes. The even mode frequencies are at 3.03 GHz, 4.62 GHz and 7.43 GHz and the distribution is symmetrical with respect to the central funnel shaped resonator. The odd mode frequencies occurs at 3.49 GHz, 5.92 GHz and at 8.12 GHz in which the electric field is distributed along the main transmission line only and it do not depend on the central funnel shaped resonator.
4. PARAMETRIC STUDY The parametric effects on the performance of the filter with the parameters ‘Wlf’ width of the rectangular portion of upper funnel shaped resonator, ‘Llf’ length of the rectangular portion of upper funnel shaped resonator, ‘Wuf’ width of the tapered portion of upper funnel shaped resonator and ‘Luf’ length of the tapered portion of upper funnel shaped resonator are conducted since these parameters are responsible for the upper and lower transmission zeros and is given below.
4.1.
THE EFFECT OF ‘Luf’ LENGTH OF THE TAPERED PORTION OF UPPER FUNNEL SHAPED RESONATOR IN TRANSMISSION AND REFLECTION CHARACTERISTICS OF THE PROPOSED FILTER. As the length of tapered portion of upper funnel shaped resonator increases from Luf = 2.41
mm to 4.41 mm, the location of both lower and upper transmission zeros decreases as shown in Fig. 17. Here, the ideal value used is Luf = 3.41 mm, since for this case, both lower and upper cut-off points show suppression better than 20 dB. Also, it is seen that by increasing the length of tapered portion of upper funnel shaped resonator, the lower out of band suppression level gets improved and the upper out of band suppression level attenuates. Similarly, by decreasing the length of tapered portion of upper funnel shaped resonator, the upper out of band suppression level gets improved and the lower out of band suppression level attenuates. The reflection characteristics shows better passband performance greater than 20 dB for Luf = 3.41 mm. As the value of Luf varies from 2.41 mm to 4.41 mm, the passband shifts towards the lower frequency region as depicted in Fig. 18.
Transmission Characterstics (dB)
0 -5 -10 -15 -20 -25 Luf = 2.41 mm
-30
Luf = 3.41 mm
-35
Luf = 4.41 mm
-40 0
2
4
6 8 Frequency (GHz)
10
12
14
Fig. 17. Simulated transmission characteristics of the proposed filter for various values of Luf.
Reflection Characteristics (dB)
0 -10 -20 -30 -40 Luf = 2.41 mm
-50
Luf = 3.41 mm Luf = 4.41 mm
-60 0
2
4
6 8 Frequency (GHz)
10
12
14
Fig. 18. Simulated reflection characteristics of the proposed filter for various values of Luf.
4.2.
THE EFFECT OF ‘Wuf’ WIDTH OF THE TAPERED PORTION OF UPPER FUNNEL SHAPED RESONATOR IN TRANSMISSION AND REFLECTION CHARACTERISTICS OF THE PROPOSED FILTER. As the width of tapered portion of upper funnel shaped resonator increases from Wuf = 3.78
mm to 5.78 mm, the location of both lower and upper transmission zeros decreases as shown in Fig. 9. Here, the ideal value used is Wuf = 4.78 mm, since for this case, both lower and upper cutoff points show suppression better than 20 dB. Also, it is seen that by increasing the width of tapered portion of upper funnel shaped resonator, the lower out of band suppression level gets improved and the upper out of band suppression level attenuates. Similarly, by decreasing the width of tapered portion of upper funnel shaped resonator, the upper out of band suppression level gets improved and the lower out of band suppression level attenuates.
Transmission Characteristics (dB)
0 -5 -10 -15 -20 -25 Wuf = 3.78 mm
-30
Wuf = 4.78 mm
-35
Wuf = 5.78 mm
-40 0
2
4
6 8 Frequency (GHz)
10
12
14
Fig. 19. Simulated transmission characteristics of the proposed filter for various values of Wuf.
The reflection characteristics shows better passband performance greater than 20 dB for Wuf = 4.78 mm. As the value of Wuf varies from 3.78 mm to 5.78 mm, the passband shifts towards the lower frequency region as depicted in Fig. 20.
Reflection Characteristics (dB)
0 -5 -10 -15 -20 -25 Wuf = 3.78 mm
-30
Wuf = 4.78 mm
-35
Wuf = 5.78 mm
-40 0
2
4
6 8 Frequency (GHz)
10
12
14
Fig. 20. Simulated reflection characteristics of the proposed filter for various values of Wuf.
4.3.
THE EFFECT OF ‘Wlf’ WIDTH OF THE RECTANGULAR PORTION OF UPPER FUNNEL SHAPED RESONATOR IN TRANSMISSION AND REFLECTION CHARACTERISTICS OF THE PROPOSED FILTER. We can see that as the width of rectangular portion of upper funnel shaped resonator
increases from Wlf = 0.3 mm to 1.1 mm, the location of lower transmission zero positions shifts towards higher value. Also, it is seen that by increasing the width of rectangular portion of upper funnel shaped resonator, the passband insertion loss near the lower stopband gets improved. Here, the ideal value used is Wlf = 1.1 mm, since for this case, the passband insertion loss is minimum as
shown in Fig. 21. The variation of reflection characteristics of the proposed filter is depicted in Fig. 22. As the value of Wlf varies from 0.3 mm to 1.1 mm, the passband return loss gets improved and shows greater than 20 dB performance for Wlf = 1.1 mm.
Transmission Characteristics (dB)
0 -10 -20 -30 -40 Wlf = 0.3 mm
-50
Wlf = 0.7 mm Wlf = 1.1 mm
-60 0
2
4
6 8 Frequency (GHz)
10
12
Fig. 21. Simulated transmission characteristics of the proposed filter for various values of Wlf.
Reflection Characteristics (dB)
0 -10 -20 -30 -40 Wlf = 0.3 mm
-50
Wlf = 0.7 mm Wlf = 1.1 mm
-60 0
2
4
6 8 Frequency (GHz)
10
12
Fig. 22. Simulated reflection characteristics of the proposed filter for various values of Wlf.
THE EFFECT OF ‘Llf’ LENGTH OF THE RECTANGULAR PORTION OF UPPER
FUNNEL
SHAPED
RESONATOR
IN
TRANSMISSION
AND
REFLECTION CHARACTERISTICS OF THE PROPOSED FILTER. 0 Transmission Characteristics (dB)
4.4.
-5 -10 -15 -20 -25 -30
Llf = 8.15 mm Llf = 9.15 mm
-35
Llf = 10.15 mm
-40 0
2
4
6 8 Frequency (GHz)
10
12
14
Fig. 23. Simulated transmission characteristics of the proposed filter for various values of Llf.
Fig. 24. Simulated reflection characteristics of the proposed filter for various values of Llf.
As the length of the rectangular portion of the upper funnel shaped resonator increases from Llf = 8.15 mm to 10.15 mm, the upper and lower transmission zero positions got shifted to the right side as shown in Fig. 23. In the proposed design, we selected Llf = 9.15 mm because the stopband suppression level achieved is better than 20 dB for 9.15 mm. The variation of reflection characteristics with respect to Llf is shown in Fig. 24 and from this analysis, we concluded that the return loss performance obtained shows better performance for Llf = 9.15 mm.
5. RESULTS AND DISCUSSIONS The filter is simulated and optimized using the electromagnetic simulation software IE3D and fabricated using MITS 11 Auto lab fabrication machine. The measurements are taken using R&S ZVB 20 Vector Network Analyzer. For ensuring impedance matching, the input and output ports are properly matched. Since we are using the FR4 substrate with thickness 0.8 mm, the width of both ports must be 1.5 mm. The optimized design parameters of the proposed filter are as follows: L1 = 22.02, W1= 0.5, La = 7.67, Wa = 2.4, W2 = 0.3, Lwe = 3.64, Le= 2.45, We = 1, Wlf = 1.1, Llf = 9.15, Wuf = 4.78, Luf = 3.41, Duf = 3.85, Lfl= 0.73, Wbl = 1.4, Lbl = 1.25 and Wfl = 0.36. (All dimensions are in mm). The 3 dB centre frequency of the developed filter is 5.68 GHz. The 3 dB pass band has a fractional bandwidth of 97.57 %. The insertion loss (IL) at centre frequency is less than 1.62 dB. The minimum value of return loss (RL) obtained is better than 20 dB within the pass band, and sharp selectivity is observed because of the two transmission zeros obtained at the lower and upper passband edges. The simulated and measured transmission and reflection characteristics of the developed filter are shown in Fig. 25 (a).
10 S1
S21
S-Parameters (dB)
0 -10 -20 -30 -40
SL : Measured DL : Simulated
-50 0
2
4
6 8 Frequency (GHz)
10
12
14
Fig. 25. (a) Simulated and measured responses of the proposed filter [SL: solid line and D L: dotted line]
Transmission characteristics (dB)
10 S21
0 -10 -20 -30 -40 -50 0
2
4
6
8 10 12 14 Frequency (GHz)
16
18
20
Fig. 25. (b) Measured Wideband transmission characteristics of the developed filter
Even though the proposed filter exhibits upper stopband with greater than 20 dB suppression level upto 24.2 GHz, the measured wideband response is taken only upto 20 GHz because the maximum frequency range that can be measured using the available R&S ZVB 20 vector network analyzer is 20 GHz and the measured wideband response is given in Fig. 25 (b). TABLE. I. Comparison of the Proposed Filter with Existing Filters Dielectric
Centre
3- dB
frequency
FBW
(GHz)
(%)
[2]
5.96
85
10
[5]
4
67
[6]
4.175
[7]
Ref.
RL
constant, εr/
(dB)
thickness
Loss
Maximum
Upper
tangent,
IL
stopband
stopband
tan δ
(dB)
attenuation (dB)
(GHz)
4.4/1.6
0.02
2
15
20
<20
2.2/1.574
0.0009
NA
20
8.5
28.02
10
3.2/1.14
0.004
NA
20
7
3
80
30
2.65/1.5
0.0005
NA
20
7.5
[10]
5.3
-
<10
2.2/0.508
0.0009
NA
20
10
[13]
3.52
62.75
13
4.4/1.6
0.02
0.69
20
9.72
[14]
4.5
69.1
12.3
2.55/0.8
0.0029
1.4
13
12.5
Proposed
5.68
97.57
>20
4.4/0.8
0.02
1.62
>20
24.2
(mm)
TABLE. I shows the comparison between existing filters and the developed, the obtained parameters of the fabricated filter are flat group delay, very good reflection characteristics and a suppression level greater than 20 dB up to 24.2 GHz.
(a)
(b)
Fig. 26. Photograph of the fabricated filter prototype (a) Top view (b) Bottom view
The photographs of the fabricated filter prototype are given in Fig. 26 (a) and (b). The top view shows the asymmetric funnel shaped resonators and side stubs with proper input output coupling. The bottom view shows the two symmetric apertures cut on the ground plane of the structure for the enhancement of coupling. The normalized circuit size, NCS of the developed filter is 0.3157 λg2. The relative stopband bandwidth of the proposed filter is 79.87 %. The roll – off rate for the lower and upper passband of the filter is 72.24 dB/GHz and 19.01 dB/GHz respectively at 20 dB suppression level.
6. CONCLUSION In this paper, a low cost wideband bandpass filter with wide pass band, good skirt selectivity with very good passband reflection characteristics is proposed and developed. The main resonator consists of a symmetrical funnel shaped resonator loaded centrally to the uniform impedance line which contributes two transmission zeros at the passband edges. Innovative and simple equations are derived for finding out the transmission zero locations and it can be applied for the realization of sharp skirt selectivity filter designs. The developed structure is compact with a normalized circuit size of 0.3157 in terms of λg2. The out of band performance is very good with a suppression level greater than 20 dB up to 24.2 GHz. The low cost realization, very good reflection characteristics, wide passband and very good spurious suppression makes this filter a promising candidate for wideband wireless communication systems. REFERENCES 1. Hong J. S., Lancaster M. J. Microstrip Filters for RF/Microwave Applications. Wiley, New
York, 2001.
2. Jhariya D., Azad A. R., Mohan A., and Sinha M. Compact wideband bandpass filter using fish
spear shaped multimode resonator. Microwave and Optical Technology Letters; Vol. 57, No. 12, 2833-2837, 2015. https://doi.org/10.1002/mop.29437 3. Dey R., Sarkar M., Rana B. A novel band pass filter design and analysis using inductive via
series.
AEU
–
Int
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Electron
and
Commun
2019;
99:315-324.
https://doi.org/10.1016/j.aeue.2018.12.009 4. Sassi I., Talbi L., Hettak K. Compact bandpass filters based on linked hexagonal-omega
resonators. Microwave and Optical Technology Letters; Vol. 58. No. 5, 1049-1052, 2016. https://doi.org/10.1002/mop.29720 5. Chen C. P., Kato N., Anada T. Synthesis scheme for wideband filters consisting of three-
coupled- lines including the cross coupling between non-adjacent lines. IET Microw. Antennas Propag; Vol. 9, 1558-1566, 2015. https://doi.org/10.1049/iet-map.2014.0467 6. Ma X. B., Jiang T. Wideband bandpass filter with controllable bandwidth and high selectivity
using two different types of resonators. Microwave and Optical Technology Letters; Vol. 57, 1319 - 1323, Vol. 57, 2015. https://doi.org/10.1002/mop.29085 7. Li L., Li Z. F. Side coupled shorted microstrip line for compact quasi elliptic wideband
bandpass filter design.IEEE Microwave and Wireless Components Letters; Vol. 20, No. 6, 322-324, 2010. http://dx.doi.org/10.1109%2FLMWC.2010.2047516 8. Xia X., Cheng X., Chen F. and Deng X. Compact UWB bandpass filter with sharp roll-off
using APCL structure. IET Electronics Letters; Vol. 54, No. 13, 837-839, 2018. https://doi.org/10.1049/el.2018.1151
9. Luo X., Qian H., Ma J. G. and Li E. P., Wideband bandpass filter with excellent selectivity
using new CSRR based resonators. IET Electronics Letters; Vol. 46, No. 20, 2010. https://doi.org/10.1049/el.2010.1817 10. Wang H., Chu Q. X., Gong J. Q., A compact wideband microstrip filter using folded multiple
mode resonator.IEEE Microwave and Wireless Components Letters; Vol. 19, No. 5, 287-289, 2009. https://doi.org/10.1109/LMWC.2009.2017591 11. Mayani M. G., Asadi S., Pirhadi A., Mahani S. M., Design and analysis of a super compact
wide-band bandpass filter based on meta-material resonators; AEU – Int J Electron and Commun 2018. 97:79-84. https://doi.org/10.1016/j.aeue.2018.10.010 12. Taibi A., Trabelsi M., Slimane A., Belaroussi M. T., Raskin J. P., A novel design method for
compact UWB bandpass filters.IEEE Microwave and Wireless Components Letters; Vol. 25, No. 1, 4-6, 2015. https://doi.org/10.1109/LMWC.2014.2363016 13. Yechou L., Tribak A., Kacim M., Zbitou J., Sanchez AM. A novel wideband bandpass filter
using coupled lines and T- shaped transmission lines with wide stopband on low cost substrate. Progress
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doi:10.2528/PIERC16062204 14. Zhang Z. C., Liu H. W., Wong S. W., Compact wideband bandpass filter based on double -T-
shaped stub loaded resonator and loading technique for zero -voltage point. Int. J RF Microw Comput Aided Eng.;2018 https://doi.org/10.1002/mmce.21197 15. Ghazali A. N., Sazid M., Virdee B., A compact UWB-BPF based on microstrip to cpw
transition with multiple transmission zeros. Microwave and Optical Technology Letters; Vol. 60, 1925-1928, 2018. https://doi.org/10.1002/mop.31274 16. Pozar D. M., Microwave Engineering, 4th ed. USA, Wiley, 2011.
17. Li Y., Li W., Liu C. and Ye Q., A Compact UWB Bandpass Filter with ultra - narrow tri- notch
band characteristic. ACES journal, Vol. 29, No. 2, February 2014 18. Li Y., Li W., Yu W. and Liu C., A miniaturization Band-pass Filter with ultra-narrow multi-
notch band characteristic for ultra wideband communication applications. ACES journal, vol. 29, No. 4, April 2014 19. Chu Q. X., Wu X. H. and Tian X. K., Novel UWB Bandpass Filter Using Stub-
Loaded Multiple-Mode Resonator. IEEE Microwave and Wireless Components Letters, vol. 21, no. 8, pp. 403-405, Aug 2011. https://doi.org/10.1109/LMWC.2011.2160526 AUTHOR BIOGRAPHIES
Ami Iqubal was born in Kerala, India in May,1988. She received the B. Tech in Electronics and Communication Engineering from CUSAT,Kerala in 2010; received M.Tech in Applied Electronics from Mahatma Gandhi University, Kerala in 2013 and currently working towards Ph. D degree in Electronics Engineering at CUSAT. Her research interests include the design ,analysis and development of microwave filters for wideband applications.
Abdulla P received the B. Tech in Eletrical and Electronics Engineering from University of Calicut, Kerala in 1991; received M.Tech in Electronics from CUSAT, Kerala in1993 and Ph.D degree in Electronics and Communication Engineering from IIT Kharagpur in 2009. He is currently working as a Professor in Division of Electronics Engineering, School of Engineering, Cochin University of Science and Technology. His areas of interst include design and development of microwave filters, Antennas and Computational Electromagnetics. He served as the Secretary IEEE Antenna Prpagation Society (APS), Kerala Section and currently the Vice Chair of APS, Kerala Section. He has more than 60 publications in his credit both international journals and conferences.
7. BANDPASS FILTER BASED ON ASYMMETRIC FUNNEL SHAPED RESONATORS WITH ULTRA WIDE UPPER STOPBAND CHARACTERISTICS Ami Iqubal and Abdulla P. Division of Electronics, School of Engineering, CUSAT, Cochin-22, Kerala, India. Corresponding author: [email protected] LIST OF FIGURES AND TABLES
Fig. 2. (a) Geometry (b) LC equivalent circuit
0
S-parameters (dB)
-5 -10 -15 -20 -25 -30
S 11
-35
S 12
-40 0
2
4
6
8
Frequency (GHz)
10
12
Fig. 2. Simulated frequency response of the basic funnel shaped resonator
Fig. 3. Basic upper funnel shaped resonator with plane of symmetry A-A'
14
Fig. 4. Equivalent transmission line model of the funnel shaped resonator
Fig. 5. Odd mode equivalent circuit of the basic funnel shaped resonator
Fig. 6. Even mode equivalent circuit of the basic funnel shaped resonator
S -Parameters (dB)
0 -10 -20 -30 S11
-40
S12 -50 0
(b)
2
4
6
8
10
12
14
Frequency (GHz)
16
18
(b)
Fig. 7. (a) Basic EBG loaded structure (b) Frequency response of the EBG loaded structure
20
S - Parameters (dB)
0
-10
-20
-30
S11 S12
-40 0
2
4
6
8
10
12
14
16
18
20
Frequency (GHz) (b)
(b)
Fig. 8. (a) EBG loaded funnel structure (b) response of the EBG loaded funnel structure
Fig. 9. Layout of the basic coupling structure
S - Parameters (dB)
0 -10 -20 -30 -40
S11 S12
-50 0
2
4
6
8
10
12
14
16
18
20
Frequency (GHz) Fig. 10. Simulated response of the basic coupling structure for the range (0-20 GHz)
0
S - Parameters (dB)
-10 -20 -30 -40
S11 S12
-50 -60 0
(b)
2
4
6
8 10 12 14 Frequency (GHz)
16
18
(b)
Fig. 11. (a) Layout of the coupled EBG loaded funnel structure (b) Frequency response
20
S - Parameters (dB)
0
-10
-20
-30 S11 S12
-40 0
2
4
6
8
10
12
14
16
18
20
Frequency (GHz) (b)
(b) Fig. 12. (a) Layout (b) Frequency response
0
S - Parameters (dB)
-10 -20 -30 -40 -50
S11 S12
-60 0
(b)
2
4
6
8 10 12 14 Frequency (GHz)
16
18
(b)
Fig. 13. (a) Layout (b) Frequency response of the double funnel shaped stub loaded filter
20
S - Parameters (dB)
0 -10 S11 S12
-20 -30 -40 -50 0
2
4
6
8
`
10 12 14 16 18 20 22 24 Frequency (GHz)
(b)
(b)
Fig. 14.(a)Layout of the final structure (b) Simulated response for the range (0-25GHz)
0 Transmission characteristics (dB)
Transmission characteristics (dB)
0 -10 -20 -30 -40 -50 -60 -70
without coupling with tight coupling
-80 -90 2
4
6 8 Frequency (GHz)
(b)
10
12
-20 -30 -40
with gpa
-50
without gpa
-60
-100 0
-10
14
0
2
4
6 8 10 Frequency (GHz)
12
14
(b)
Fig. 15. Response (a) with and without coupling (b) with and without aperture on ground (gpa)
Fig. 16. Electric field intensities at: (a) upper stopband @ 15 GHz (b) lower stopband @ 1 GHz (c) transmission zero 1@ 2.6 GHz (d) transmission zero 2 @ 9.1 GHz (e) centre frequency @ 5.68 GHz (f)
mode 1 @ 3.03 GHz (g) mode 2 @ 3.49 GHz(h) mode 3 @ 4.62 GHz (i) mode 4 @ 5.92 GHz (j)mode 5 @ 7.43 GHz (k) mode 6 @ 8.12 GHz
Transmission Characterstics (dB)
0 -5 -10 -15 -20 -25 Luf = 2.41 mm
-30
Luf = 3.41 mm
-35
Luf = 4.41 mm
-40 0
2
4
6 8 Frequency (GHz)
10
12
14
Fig. 17. Simulated transmission characteristics of the proposed filter for various values of Luf.
Reflection Characteristics (dB)
0 -10 -20 -30 -40 Luf = 2.41 Luf = 3.41
-50
Luf = 4.41
-60 0
2
4
6 8 Frequency (GHz)
10
12
14
Fig. 18. Simulated reflection characteristics of the proposed filter for various values of Luf.
Transmission Characteristics (dB)
0 -5 -10 -15 -20 -25
Wuf = 3.78 mm
-30
Wuf = 4.78 mm
-35
Wuf = 5.78 mm
-40 0
2
4
6 8 Frequency (GHz)
10
12
14
Fig. 19. Simulated transmission characteristics of the proposed filter for various values of Wuf.
Reflection Characteristics (dB)
0 -5 -10 -15 -20 -25
Wuf = 3.78smm
-30
Wuf = 4.78 mm
-35
Wuf = 5.78 mm
-40 0
2
4
6 8 Frequency (GHz)
10
12
14
Fig. 20. Simulated reflection characteristics of the proposed filter for various values of Wuf.
Transmission Characteristics (dB)
0 -10 -20 -30 -40
Wlf = 0.3mm Wlf = 0.7mm
-50
Wlf = 1.1mm
-60 0
2
4 6 8 Frequency (GHz)
10
12
Fig. 21. Simulated transmission characteristics of the proposed filter for various values of Wlf.
Reflection Characteristics (dB)
0 -10 -20 -30 -40 Wlf = 0.3 mm
-50
Wlf = 0.7 mm Wlf = 1.1 mm
-60 0
2
4
6 8 Frequency (GHz)
10
12
Fig. 22. Simulated reflection characteristics of the proposed filter for various values of Wlf.
Transmission Characteristics (dB)
0 -5 -10 -15 -20 -25 Llf = 8.15 mm
-30
Llf = 9.15 mm
-35
Llf = 10.15 mm
-40 0
2
4
6 8 Frequency (GHz)
10
12
14
Fig. 23. Simulated transmission characteristics of the proposed filter for various values of Llf.
Fig. 24. Simulated reflection characteristics of the proposed filter for various values of Llf.
10 S1
S21
S-Parameters (dB)
0 -10 -20 -30 -40
SL : Measured DL : Simulated
-50 0
2
4
6 8 Frequency (GHz)
10
12
14
Fig. 25. (a) Simulated and measured responses of the proposed filter [SL: solid line and D L: dotted line]
Transmission characteristics (dB)
10 S21
0 -10 -20 -30 -40 -50 0
2
4
6
8 10 12 14 Frequency (GHz)
16
18
20
Fig. 25. (b) Measured Wideband transmission characteristics of the developed filter
TABLE. I. Comparison of the Proposed Filter with Existing Filters
Dielectric
Centre
3- dB
frequency
FBW
(GHz)
(%)
[2]
5.96
85
10
[5]
4
67
[6]
4.175
[7]
RL
constant, εr/
(dB)
thickness
Loss
Maximum
Upper
tangent,
IL
stopband
stopband
tan δ
(dB)
attenuation (dB)
(GHz)
4.4/1.6
0.02
2
15
20
<20
2.2/1.574
0.0009
NA
20
8.5
28.02
10
3.2/1.14
0.004
NA
20
7
3
80
30
2.65/1.5
0.0005
NA
20
7.5
[10]
5.3
-
<10
2.2/0.508
0.0009
NA
20
10
[13]
3.52
62.75
13
4.4/1.6
0.02
0.69
20
9.72
[14]
4.5
69.1
12.3
2.55/0.8
0.0029
1.4
13
12.5
Proposed
5.68
97.57
>20
4.4/0.8
0.02
1.62
>20
24.2
Ref.
(a)
(mm)
(b)
Fig. 26. Photograph of the fabricated filter prototype (a) Top view (b) Bottom view
Declaration of interests
☒ The authors declare that they have no known competing financial interests or personal relationships that could have appeared to influence the work reported in this paper.
☐The authors declare the following financial interests/personal relationships which may be considered as potential competing interests:
S1434-8411(19)30780-0 https://doi.org/10.1016/j.aeue.2020.153062 AEUE 153062
To appear in:
International Journal of Electronics and Communications
Received Date: Revised Date: Accepted Date:
3 April 2019 29 December 2019 1 January 2020
Please cite this article as: A. Iqubal, P. Abdulla, Bandpass filter based on asymmetric funnel shaped resonators with ultra wide upper stopband characteristics, International Journal of Electronics and Communications (2020), doi: https://doi.org/10.1016/j.aeue.2020.153062
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Title: Bandpass Filter Based on Asymmetric Funnel Shaped Resonators with Ultra Wide Upper Stopband Characteristics Authors: 1.
Ami Iqubal (Degree: pursuing Ph. D), email id: [email protected]
2.
Abdulla Parambil (Degree: Ph. D), email id: [email protected]
Affiliation: Division of Electronics, School of Engineering, Cochin University of Science and Technology, Kerala, India Corresponding Author: Ami Iqubal
BANDPASS FILTER BASED ON ASYMMETRIC FUNNEL SHAPED RESONATORS WITH ULTRA WIDE UPPER STOPBAND CHARACTERISTICS Ami Iqubal and Abdulla P. Division of Electronics, School of Engineering, CUSAT, Cochin-22, Kerala, India. Corresponding author: [email protected] Abstract — A novel microstrip bandpass filter with wide passband, good skirt selectivity and very good passband reflection characteristics using asymmetric funnel shaped resonators is presented in this paper. IE3D simulation software is used for the design and analysis of the proposed filter and the electric field intensities at various frequencies of interest are plotted and verified the role of even and odd mode resonances in achieving wide passband. There are six modes, including three even and three odd modes for the developed filter. Novel equations are derived for finding out the location of transmission zeros and the role of resonators in achieving sharp selectivity are verified. The out of band performance of filter is very good with a suppression level greater than 20 dB upto 24.2 GHz. The centre frequency of the filter is 5.68 GHz with a fractional bandwidth of 97.57 %. The fabrication process is carried out using MITS 11 Autolab fabrication machine with the low cost FR4 substrate and verified the performance improvement of the developed filter. Keywords – Bandpass Filter (BPF); Selectivity; Stopband; Transmission Zero; Wide Fractional Bandwidth
1. INTRODUCTION Wideband bandpass filters are the key components in wideband communication systems for removing the unwanted signals and noise over a large bandwidth. The main design considerations for developing an ideal bandpass filter are low cost, good in and out-of-band performance, sharp selectivity of the passband, very good relative stopband bandwidth and simple design procedures. Federal Communications Committee (FCC) in February 2002 granted the unlicensed use of ultra wideband (UWB) communications spectrum from 3.1 GHz to 10.6 GHz. Wide arrays of techniques were investigated to realize bandpass filter configurations over the last decade [2-15]. A wideband bandpass filter developed using fish spear shaped multiple mode resonator [2] wherein the transmission zeros were absent at the passband edges and hence the passband to stopband transition is very poor. Filter design using inductive via series [3] exhibits very good performance except its poor roll-off. Bandpass filter with cascaded linked hexagonal omega resonators [4] suffers from very small stopband bandwidth. Wideband filters consisting of three coupled lines including the cross coupling between adjacent resonators were discussed in [5]. The roll–off and relative stopband bandwidth shows very poor performance. The filters developed in [6-7] fail to achieve very good spurious suppression. UWB filter based on asymmetric parallel-coupled line (APCL) is discussed in [8], but it achieves stopband suppression only upto 12.8 GHz. Wideband bandpass filter design using complementary split ring resonators (CSRR) is developed in [9]. However maximum stopband achievable is very low for this filter. In [10], a folded multiple mode resonator is employed for the development of wideband bandpass filters. Even though a passband with wide bandwidth is obtained, the reflection characteristics of the filter show poor performance. Various filter topologies including meta material resonators [11], and double T- shaped resonators [14] suffers from very low upper outband skirt rejection
performance. UWB bandpass filter based on an objective function [12], structure based on microstrip to CPW transition [15] and filter using stub loaded multiple mode resonator [19] achieves upper stopband suppression only less than 18 GHz. The motivation of this paper is to develop an easily implementable wideband bandpass filter structure with very good stopband to passband transition and ultra wide upper stopband characteristics using a very low cost easily available FR4 substrate material with a thickness of 0.8 mm and a dielectric constant of 4.4. The design procedures and equations for finding out the transmission zero locations are explained in detail. Two high impedance stubs loaded symmetrically to the upper funnel shaped resonator played a great role in obtaining passband return loss characteristics better than 20 dB. Electric field intensities of various frequencies of interest are analyzed using IE3D simulation software and verified the role of even and odd mode resonances in achieving wide passband. MITS 11 Autolab fabrication machine is used for the prototype fabrication. The measurements were carried out using R&S ZVB 20 vector network analyzer and verified that the in and out-of-band performance of the developed filter is very good.
2. THE WIDEBAND BANDPASS FILTER DESIGN The bandpass filter having the desired wideband characteristics is designed as follows. 2.1.
THE DESIGN AND ANALYSIS OF THE BASIC FUNNEL SHAPED RESONATOR The filter design begins by allocating transmission zeros at lower and upper band edges. A
funnel shaped resonator consisting of a rectangular stub and a tapered portion is used for achieving sharp selectivity because the linear tapering provides better reflection characteristics since the
transition from high impedance line to low impedance line is gradual. The structure and equivalent circuit of the basic funnel shaped resonator is given in Fig. 1 (a) and (b) respectively.
Fig. 1. (a) Geometry (b) LC equivalent circuit
The structural dimensions of the funnel shaped resonator are as follows. The tapered section is having a width of Wuf = 4.78 mm, a length of Luf = 3.41 mm and an impedance of ZB. The rectangular region is characterized with a length of Llf = 9.15 mm, a width of Wlf = 1.1 mm and an impedance of ZA. The basic resonator is connected to a high impedance line of characteristic impedance ZL. The funnel shaped resonator can be approximately modeled as a series and shunt pairs of inductors and capacitors La, Ca and Lb, Cb respectively. Port 1 and Port 2 of the resonator are terminated with transmission lines of 50 Ω characteristic impedance having a width 1.5 mm for ensuring proper impedance matching since here we are using FR4 substrate with dielectric constant, εr = 4.4, thickness, h = 0.8 mm and a loss tangent of 0.02. The characteristics of the funnel shaped resonator is calculated by using the basic equations for εre and ZC [1, 16], where εre and ZC are the effective dielectric constant and the characteristic impedance of the lines respectively. The calculated values of ZA = 60.29 Ω and ZB = 22.52 Ω. The effective dielectric constant of rectangular region of funnel shaped resonator, εreA= 3.245 and the effective dielectric
constant of the tapered portion of the funnel shaped resonator, εreB = 3.68. The values of series and parallel combinations of the inductors and capacitors can be calculated by using the equations (1) and (2), where ‘c’ is the velocity of light in free space.
𝐿𝑎 =
𝐶𝑎 =
𝑍𝐴 𝜀𝑟𝑒𝐴
𝑍𝐵 𝜀𝑟𝑒𝐵
𝑐
𝑐
𝐿𝑙𝑓, 𝐿𝑏 =
𝐿𝑙𝑓 𝜀𝑟𝑒𝐴
𝐿𝑢𝑓 𝜀𝑟𝑒𝐵
𝑐𝑍𝐴
𝑐𝑍𝐵
, 𝐶𝑏 =
(1)
𝐿𝑢𝑓
(2)
The equivalent values of inductors and capacitors are obtained as follows. La = 3.312 nH, Lb=0.49 nH, Ca = 0.911 pF and Cb = 0.96 pF. The equivalent impedance of the lower rectangular portion of the funnel shaped resonator can be represented as a series combination of La and Ca and is given in equation (3). The equivalent impedance of the tapered region of the funnel shaped resonator is represented as the parallel combination of impedances Lb and Cb and is given in equation (4).
𝑍𝐴 =
1 ― 𝜔2𝐿𝑎𝐶𝑎
(3)
𝑗𝜔𝐶𝑎
𝑗𝜔𝐿𝑏
(4)
𝑍𝐵 = 1 ― 𝜔2𝐿 𝐶
𝑏 𝑏
The ABCD parameters of the funnel shaped resonator are given in the equations (5 - 7). 𝜔2𝐿𝐶𝑎(1 ― 𝜔2𝐿𝑏𝐶𝑏)
𝐴 = 𝐷 = 1 ― (1 ― 𝜔2𝐿 𝐶 )(1 ― 𝜔2𝐿 𝐶 ) ― 𝜔2𝐿 𝐶 𝑎 𝑎
𝑏 𝑏
𝑏 𝑎
𝑗𝜔3𝐿2𝐶𝑎(1 ― 𝜔2𝐿𝑏𝐶𝑏)
𝐵 = 2𝑗𝜔𝐿 ― (1 ― 𝜔2𝐿 𝐶 )(1 ― 𝜔2𝐿 𝐶 ) ― 𝜔2𝐿 𝐶 𝑎 𝑎
𝑏 𝑏
𝑗𝜔𝐶𝑎(1 ― 𝜔2𝐿𝑏𝐶𝑏)
𝐶 = (1 ― 𝜔2𝐿 𝐶 )(1 ― 𝜔2𝐿 𝐶 ) ― 𝜔2𝐿 𝐶 𝑎 𝑎
𝑏 𝑏
𝑏 𝑎
𝑏 𝑎
(5)
(6)
(7)
By substituting the values of A, B, C and D, we can find out the transmission and reflection characteristics S11 and S12. The transmission and reflection characteristics of the main resonator implies that the series inductor – capacitor pair [La, Ca] and the shunt inductor –capacitor pair [Lb, Cb] are responsible for the filter response. By varying the values of La, Lb, Ca and Cb, we can control the position of transmission zeros. By equating the value of transmission characteristics to zero, the lower and upper finite frequency attenuation poles at locations fz1 and fz2 can be determined as given in equations (8 – 11).
[𝜔4(𝐿𝑎𝐿𝑏𝐶𝑎𝐶𝑏) ― 𝜔2(𝐿𝑎𝐶𝑎 + 𝐿𝑏𝐶𝑏 + 𝐿𝑏𝐶𝑎) + 1] = 0
(8)
We can find out the resonant frequencies corresponding to the lower and upper passband transmission zeros by solving for 𝜔4 in the above equation (8) using the basic equation for finding out the roots of a quadratic equation and ω can be determined as given below. 𝜔= ±
∆1 ± ∆12 ― 4∆2
(9)
2∆2
where, ∆1 = 𝐿𝑎𝐶𝑎 + 𝐿𝑏𝐶𝑏 + 𝐿𝑏𝐶𝑎 , and ∆2 = 𝐿𝑎𝐿𝑏𝐶𝑎𝐶𝑏 The frequencies corresponding to the transmission zero locations are obtained as, 1
fz1 = 2𝜋
and,
1
fz2 = 2𝜋
∆1 + ∆12 ― 4∆2 2∆2
∆1 ― ∆12 ― 4∆2 2∆2
(10)
(11)
0
S-parameters (dB)
-5 -10 -15 -20 -25 -30
S 11
-35
S 12
-40 0
2
4
6
8
Frequency (GHz)
10
12
14
Fig. 2. Simulated frequency response of the basic funnel shaped resonator
Even though the funnel shaped resonator exhibits sharp skirt rejection and very good passband response, the resonator fails to suppress the spurious harmonics of the lower and upper stopband regions. The values of ∆1= 5.1105112 X 10-21 and ∆2= 1.4193 X 10-42 are obtained by substituting the values of La,Ca, Lb and Cb. The calculated values of resonant frequencies are fz1= 2.29 GHz and, fz2= 9.27 GHz obtained from equations (10) and (11). Fig. 2 shows the IE3D software simulated frequency response of the upper funnel shaped resonator having fz1 = 2.6 GHz and fz2 = 9.1 GHz which is in good accordance with the theory. Slight discrepancy in the theoretical and simulated values of transmission zeros are because of the fact that the dielectric loss of the substrate used is not considered in theoretical analysis. The attenuation zero of the resonator is located at 7.1 GHz with an attenuation level of -34 dB.
2.2.
ODD AND EVEN MODE ANALYSIS OF THE BASIC UPPER FUNNEL SHAPED FILTER STRUCTURE The basic structure of the upper funnel shaped resonator showing the plane of symmetry
A-A’ is shown in Fig. 3 and the equivalent transmission model of the funnel shaped resonator is given in Fig. 4.
Fig. 3. Basic upper funnel shaped resonator with plane of symmetry A-A'
Considering the plane A-A', the resonator is symmetrical in nature and the odd and even mode method can be applied for analysing it.
Fig. 4. Equivalent transmission line model of the funnel shaped resonator
When odd-mode excitation is applied, the potential difference along the symmetrical plane A-A' is zero and the central point of the resonator is short circuited. By neglecting the input and output port impedance, the odd mode equivalent transmission model is as shown below in Fig. 5.
Fig. 5. Odd mode equivalent circuit of the basic funnel shaped resonator
―𝑗
The input admittance for the odd mode, Yin, odd of the resonator = 𝑍1 cot 𝜃1
(12)
Thus the odd modes are fixed and do not depend on the upper funnel shaped resonator. Odd mode resonances can be calculated by equating Yin, odd = 0. Under even mode of excitation, the symmetrical plane A-A' should be open circuited as given in the references [17] and [18]. The even mode equivalent model of the basic upper funnel shaped resonator is shown in Fig. 6.
Fig. 6. Even mode equivalent circuit of the basic funnel shaped resonator tan 𝜃1 𝑍1
+∅
The input admittance for the even mode, Yin, even of the resonator = 1 ― tan 𝜃1𝑍1∅ where ∅ =
Yin, even =
tan 𝜃3 𝑍3
+
tan 𝜃2 𝑍2
𝑍1𝑍2tan 𝜃3 + 𝑍1𝑍3tan 𝜃2 + 𝑍2𝑍3tan 𝜃1 𝑍2𝑍3 ― 𝑍1 tan 𝜃1 tan 𝜃2 tan 𝜃3
(13) (14)
(15)
The even mode frequencies can be controlled by the funnel shaped resonator and the even mode resonances can be calculated by equating Yin, even = 0.
In order to improve the characteristics of the funnel shaped resonator, the spurious bands in the lower and upper stopband regions must be suppressed. For achieving a very good bandpass filter response with sharp outband spurious suppression characteristics, the basic funnel shaped resonator is loaded with electronic band gap (EBG) based suppression cells with proper coupling as described in the section 2.3. 2.3.
DESIGN OF EBG BASED SUPPRESSION CELLS For suppressing harmonics, four band gap structures having a width, We = 1 mm, Le = 2.45
mm are loaded symmetrically in side locations with respect to the central transmission line with a lateral spacing of Lwe= 3.64 mm. The basic structure and the frequency response of the EBG are given in Fig. 7 (a) and (b). The simulated frequency response shows that a filtering response is obtained by the loading of Electronic band gap structures to the main transmission line with a 3 dB cut-off frequency of 7.8 GHz. The EBG loaded filter structure exhibits the rejection of higher order harmonics as evident from the obtained filter response.
S -Parameters (dB)
0 -10 -20 -30 S11
-40
S12 -50 0
(a)
2
4
6
8
10
12
14
Frequency (GHz)
16
18
(b)
Fig. 7. (a) Basic EBG loaded structure (b) Frequency response of the EBG loaded structure
20
The wide stopband of the EBG loaded structure is used to suppress the higher order harmonics of the bandpass filter and thus it proves to be an ideal candidate for the rejection of higher order harmonics for the funnel shaped bandpass filter. The frequency response of the EBG loaded funnel shaped resonator is explained in the section 2.4. 2.4.
EBG LOADED FUNNEL SHAPED RESONATOR The electronic band gap loaded funnel shaped resonator structure and its frequency
response is given in Fig. 8 (a) and (b). Four identical EBG structures are placed symmetrically to the basic funnel shaped resonator for suppressing harmonics in the upper stopband region. The transmission zeros are located at 2.6 GHz and 9.1 GHz. The attenuation zeros are located at 0.1 GHz with a suppression level of -36.39 dB, at 4.925 GHz with a suppression level of -33.17 dB, at 7.337 GHz with a suppression level of -32 dB and at 8.543 GHz with a suppression level of 17.05 dB. Thus the EBG loaded funnel shaped resonator is having two transmission zeros at the lower and upper passband edges and four attenuation zeros.
S - Parameters (dB)
0
-10
-20
-30
S11 S12
-40 0
2
4
6
8
10
12
14
16
18
20
Frequency (GHz) (a)
(b)
Fig. 8. (a) EBG loaded funnel structure (b) response of the EBG loaded funnel structure
Three attenuation zeros are located at the passband region and one at lower stopband. The lower attenuation zero at 0.1 GHz must be shifted to the passband region for the suppression of lower harmonics in order to reject spurious responses at the lower stopband. 2.5.
DESIGN AND ANALYSIS OF THE GROUND PLANE APERTURE BASED TWO STAGE THREE LINE PARALLEL COUPLED RESONATORS For shifting the attenuation zeros present at the lower stopband, proper input output
coupling must be employed with tight coupling [5]. Comparing the conventional bandpass filter development methods like end-coupled and parallel coupled lines, the coupling intensity is more for parallel coupled lines [1]. The coupling intensity is limited by the tolerance introduced by the currently available fabrication methods. The minimum coupling gap that can be developed practically is only 0.2 mm in our laboratory. By keeping the minimum achievable coupling gap, an aperture is cut in to the ground plane exactly beneath the parallel coupled lines for attaining wide bandwidth. Two stage ground plane aperture based three line parallel coupled lines is introduced for shifting the lower stopband attenuation zeros to the passband. The basic layout of the coupling structure is shown in Fig. 9, black region shows the top view of the two stage three line coupled filter and the grey region represents the bottom view of the coupling structure.
Fig. 9. Layout of the basic coupling structure
The guided wavelength (λg) corresponding to the centre frequency of the proposed filter is 29 mm. The three line parallel coupled structures must be selected in such a way that the coupling length must be in the range of λg/4 for ensuring wide bandwidth. An optimized value of 7.67 mm is used in our design. The width of the three line parallel coupled structures are selected in order to accommodate three lines within 1.5 mm, the width of the input output port for ensuring impedance matching and by maintaining the minimum gap between lines as 0.2 mm. The dimensions of the basic structure are given as: L1 = 22.02 mm, W1= 0.5 mm, La= 7.67 mm, Wa= 2.4 mm and W2 = 0.3 mm.The frequency response of the filter is shown in Fig. 10.
S - Parameters (dB)
0 -10 -20 -30 -40
S11 S12
-50 0
2
4
6
8
10
12
14
16
18
20
Frequency (GHz) Fig. 10. Simulated response of the basic coupling structure for the range (0-20 GHz)
The reflection characteristics exhibits three attenuation zeros and are located at 3.6 GHz with - 28.9 dB suppression level, 8.4 GHz at -25.5 dB and at 16.38 GHz with -35.3 dB suppression level. From the frequency response, it is evident that the ground plane aperture based three line input output coupling is an ideal candidate for providing proper coupling. 2.6.
CASCADED STRUCTURE OF GROUND PLANE APERTURE BASED LINES AND EBG LOADED FUNNEL SHAPED RESONATORS For improving the out of band spurious rejection of the filter, three line parallel coupling
with proper ground plane aperture is fed to the EBG loaded funnel shaped resonator. The modified filter structure after the introduction of coupling is given in Fig. 11 (a). The frequency response of the coupled EBG loaded funnel shaped resonator for the frequency range from 0 to 20 GHz is given in Fig. 11 (b).
0
S - Parameters (dB)
-10 -20 -30 -40
S11 S12
-50 -60 0
2
4
6
8 10 12 14 Frequency (GHz)
(a)
16
18
20
(b)
Fig. 11. (a) Layout of the coupled EBG loaded funnel structure (b) Frequency response
The frequency response exhibits four attenuation zeros within the passband of the filter and are located at 3.3 GHz with -39 dB suppression level, at 5.29 GHz with -23 dB suppression level, at 7.7 GHz with -32 dB suppression level and at 8.4 GHz with -35 dB attenuation level. 2.7.
CASCADED STRUCTURE WITH FUNNEL SHAPED ATTENUATION CELL The structure and response of the suppressing cell are shown in Fig. 12 (a) and (b). The
dimensions of the attenuation cell are: Lfl = 0.73 mm, Wfl = 0.36 mm, Wbl= 1.4 mm and Lbl = 1.25 mm. The transmission zero at 15.8 GHz makes this suppressing cell a promising candidate for the suppression of spurious harmonics in the frequency range of interest.
S - Parameters (dB)
0
-10
-20
-30 S11 S12
-40 0
2
4
6
8
10
12
14
16
18
20
Frequency (GHz) (a)
(b) Fig. 12. (a) Layout (b) Frequency response
0
S - Parameters (dB)
-10 -20 -30 -40 -50
S11 S12
-60 0
(a)
2
4
6
8 10 12 14 Frequency (GHz)
16
18
20
(b)
Fig. 13. (a) Layout (b) Frequency response of the double funnel shaped stub loaded filter
The lower funnel shaped resonator which is used as the suppressing cell structure as shown in Fig. 13. (a) provides sufficient attenuation at higher frequencies and it contributes to the wide upper stopband rejection of the proposed filter structure. The attenuation zero of the lower funnel shaped resonator which is located at 8.04 GHz with -36.95 dB attenuation level interacts with the
transmission poles of the coupled ground plane aperture based EBG loaded upper funnel shaped resonator and as a result, the modified filter exhibits attenuation zeros at 3.03 GHz, 3.49 GHz, 4.62 GHz, 5.92 GHz, 7.43 GHz and at 8.12 GHz. The transmission zeros of the loaded structure are located at 2.6 GHz with -30.1 dB suppression level and at 9.1 GHz with -20.3 dB suppression level as shown in Fig. 13. (b). For improving the suppression level of upper cut off frequency and for further improving the return loss characteristics of the proposed filter, two ‘I’ shaped stubs are loaded symmetrically with respect to the funnel shaped resonators as discussed in the section 2.8. 2.8.
EFFORTS FOR IMPROVING THE BANDPASS FILTER CHARACTERISTICS The symmetrically placed ‘I’ shaped resonators interacts with the central funnel shaped
resonator and as a result, the upper transmission zero which is located at 9.1 GHz having the suppression level of 20.3 dB gets shifted into 23.1 dB and thus the selectivity of the filter got improved. Two ‘I’ shaped stubs are loaded symmetrically about the upper funnel shaped stub with a slot width, Si= 0.2 mm. The dimensions of the ‘I’ shaped stub is designed with a strip width, Wi = 0.3 mm and a strip length, Li = 3.25 mm. With the introduction of ‘ I’ shaped stubs, the positions of attenuation zeros are located at 3.03 GHz, 3.49 GHz, 4.62 GHz, 5.92 GHz, 7.43 GHz and at 8.12 GHz.
S - Parameters (dB)
0 -10 S11 S12
-20 -30 -40 -50 0
`
2
4
6
8
10 12 14 16 18 20 22 24 Frequency (GHz)
(a)
(b)
Fig. 14.(a)Layout of the final structure (b) Simulated response for the range (0-25 GHz)
0 Transmission characteristics (dB)
Transmission characteristics (dB)
0 -10 -20 -30 -40 -50 -60 -70
without coupling with tight coupling
-80 -90 2
4
6 8 Frequency (GHz)
(a)
10
12
-20 -30 -40
with gpa
-50
without gpa
-60
-100 0
-10
14
0
2
4
6 8 10 Frequency (GHz)
12
14
(b)
Fig. 15. Response (a) with and without coupling (b) with and without aperture on ground (gpa)
The ‘I’ shaped stub loaded structure and the frequency response is given in Fig. 14 (a) and (b) respectively. The coupling plays an important role in achieving wide passband characteristics. The funnel shaped main resonator without ground plane aperture based parallel coupling shows a very good passband response but fails to meet the stopband requirements. For suppressing the spurious harmonics at lower band edge of the filter, the coupling based on two stage three line parallel coupled lines with ground plane aperture is used. Even though, the input output coupling is used, the filter exhibits desired characteristics only when the coupling applied is optimum. The filter response with and without proper input-output coupling is given in Fig. 15 (a). The dotted line shows the filter response under zero coupling and the solid line shows the filter response under strong coupling. Under zero and strong coupling the transmission zero frequency locations remains unchanged because of the upper funnel shaped resonator which is the contributing element for the transmission zeros at 2.6 GHz and 9.1 GHz. Under zero coupling, the resonator exhibits resonant frequencies at 3.01 GHz, 4.02 GHz, 5.92 GHz, 7.73 GHz and at 8.74 GHz. The suppression levels
corresponding to the various frequency modes under zero coupling are at -40.3 dB, -21.1 dB, -16.6 dB, -19.24 dB and at -34.1 dB. As the optimum input output coupling with ground plane aperture based parallel coupling is applied, the transmission characteristics moves upward and forms a very good wideband filter response. Even though, the two stage three line parallel coupling is used for coupling, the dimensions of the aperture plays a very important role in achieving spurious free passband as given in Fig. 15 (b). Without aperture on ground plane (gpa), the passband shows fluctuations and the filtering performance is not linear.
3. THE STUDY OF FIELD INTENSITY CHARACTERISTICS FOR VARIOUS FREQUENCIES OF INTEREST For verifying the correlation between the simulated results and the field behavior of the proposed filter, a study of field intensities is carried out using simulation software IE3D and is given in Fig. 16 (a) – (k).
Fig. 16. Electric field intensities at: (a) upper stopband @ 15 GHz (b) lower stopband @ 1 GHz (c) transmission zero 1 @ 2.6 GHz (d) transmission zero 2 @ 9.1 GHz (e) centre frequency @ 5.68 GHz (f) mode 1 @ 3.03 GHz (g) mode 2 @ 3.49 GHz (h) mode 3 @ 4.62 GHz (i) mode 4 @ 5.92 GHz (j) mode 5 @ 7.43 GHz (k) mode 6 @ 8.12 GHz
The various frequencies of interest includes lower stopband, upper stopband, centre frequency and the multiple modes in the passband. The frequencies upto the lower cut-off frequency are the lower stopband and the range of frequencies above the upper cut-off frequency are termed as upper stopband. The electric field intensity at 1 GHz and 15 GHz are taken here for visualizing the field intensities at lower and upper stopband ranges respectively as shown in Fig.
16 (a) and (b). At stopband, the field distribution shows that almost all energy is locked near the input port and it is not coupled to the output. The electric field distribution at transmission zeros 1 and 2, that is at 2.6 GHz and 9.1 GHz shows that the signal from input port is not reaching the output port and it is crowded near the upper funnel shaped resonator as given in Fig. 16 (c) and (d). The electric field intensity at the passband centre frequency 5.68 GHz is plotted in Fig. 16 (e) and it implies that full signal power is transmitted from port 1 to port 2. The electric field distribution at various transmission poles are analyzed for determining the even and odd mode frequencies in Fig. 16 (f – k). The transmission poles are known as the modes and there are 6 modes at frequencies 3.03 GHz (mode 1), 3.49 GHz (mode 2), 4.62 GHz (mode 3), 5.92 GHz (mode 4), 7.43 GHz (mode 5) and at 8.12 GHz (mode 6). The electric field distribution analysis shows that among the six modes, there are three even modes and three odd modes. The even mode frequencies are at 3.03 GHz, 4.62 GHz and 7.43 GHz and the distribution is symmetrical with respect to the central funnel shaped resonator. The odd mode frequencies occurs at 3.49 GHz, 5.92 GHz and at 8.12 GHz in which the electric field is distributed along the main transmission line only and it do not depend on the central funnel shaped resonator.
4. PARAMETRIC STUDY The parametric effects on the performance of the filter with the parameters ‘Wlf’ width of the rectangular portion of upper funnel shaped resonator, ‘Llf’ length of the rectangular portion of upper funnel shaped resonator, ‘Wuf’ width of the tapered portion of upper funnel shaped resonator and ‘Luf’ length of the tapered portion of upper funnel shaped resonator are conducted since these parameters are responsible for the upper and lower transmission zeros and is given below.
4.1.
THE EFFECT OF ‘Luf’ LENGTH OF THE TAPERED PORTION OF UPPER FUNNEL SHAPED RESONATOR IN TRANSMISSION AND REFLECTION CHARACTERISTICS OF THE PROPOSED FILTER. As the length of tapered portion of upper funnel shaped resonator increases from Luf = 2.41
mm to 4.41 mm, the location of both lower and upper transmission zeros decreases as shown in Fig. 17. Here, the ideal value used is Luf = 3.41 mm, since for this case, both lower and upper cut-off points show suppression better than 20 dB. Also, it is seen that by increasing the length of tapered portion of upper funnel shaped resonator, the lower out of band suppression level gets improved and the upper out of band suppression level attenuates. Similarly, by decreasing the length of tapered portion of upper funnel shaped resonator, the upper out of band suppression level gets improved and the lower out of band suppression level attenuates. The reflection characteristics shows better passband performance greater than 20 dB for Luf = 3.41 mm. As the value of Luf varies from 2.41 mm to 4.41 mm, the passband shifts towards the lower frequency region as depicted in Fig. 18.
Transmission Characterstics (dB)
0 -5 -10 -15 -20 -25 Luf = 2.41 mm
-30
Luf = 3.41 mm
-35
Luf = 4.41 mm
-40 0
2
4
6 8 Frequency (GHz)
10
12
14
Fig. 17. Simulated transmission characteristics of the proposed filter for various values of Luf.
Reflection Characteristics (dB)
0 -10 -20 -30 -40 Luf = 2.41 mm
-50
Luf = 3.41 mm Luf = 4.41 mm
-60 0
2
4
6 8 Frequency (GHz)
10
12
14
Fig. 18. Simulated reflection characteristics of the proposed filter for various values of Luf.
4.2.
THE EFFECT OF ‘Wuf’ WIDTH OF THE TAPERED PORTION OF UPPER FUNNEL SHAPED RESONATOR IN TRANSMISSION AND REFLECTION CHARACTERISTICS OF THE PROPOSED FILTER. As the width of tapered portion of upper funnel shaped resonator increases from Wuf = 3.78
mm to 5.78 mm, the location of both lower and upper transmission zeros decreases as shown in Fig. 9. Here, the ideal value used is Wuf = 4.78 mm, since for this case, both lower and upper cutoff points show suppression better than 20 dB. Also, it is seen that by increasing the width of tapered portion of upper funnel shaped resonator, the lower out of band suppression level gets improved and the upper out of band suppression level attenuates. Similarly, by decreasing the width of tapered portion of upper funnel shaped resonator, the upper out of band suppression level gets improved and the lower out of band suppression level attenuates.
Transmission Characteristics (dB)
0 -5 -10 -15 -20 -25 Wuf = 3.78 mm
-30
Wuf = 4.78 mm
-35
Wuf = 5.78 mm
-40 0
2
4
6 8 Frequency (GHz)
10
12
14
Fig. 19. Simulated transmission characteristics of the proposed filter for various values of Wuf.
The reflection characteristics shows better passband performance greater than 20 dB for Wuf = 4.78 mm. As the value of Wuf varies from 3.78 mm to 5.78 mm, the passband shifts towards the lower frequency region as depicted in Fig. 20.
Reflection Characteristics (dB)
0 -5 -10 -15 -20 -25 Wuf = 3.78 mm
-30
Wuf = 4.78 mm
-35
Wuf = 5.78 mm
-40 0
2
4
6 8 Frequency (GHz)
10
12
14
Fig. 20. Simulated reflection characteristics of the proposed filter for various values of Wuf.
4.3.
THE EFFECT OF ‘Wlf’ WIDTH OF THE RECTANGULAR PORTION OF UPPER FUNNEL SHAPED RESONATOR IN TRANSMISSION AND REFLECTION CHARACTERISTICS OF THE PROPOSED FILTER. We can see that as the width of rectangular portion of upper funnel shaped resonator
increases from Wlf = 0.3 mm to 1.1 mm, the location of lower transmission zero positions shifts towards higher value. Also, it is seen that by increasing the width of rectangular portion of upper funnel shaped resonator, the passband insertion loss near the lower stopband gets improved. Here, the ideal value used is Wlf = 1.1 mm, since for this case, the passband insertion loss is minimum as
shown in Fig. 21. The variation of reflection characteristics of the proposed filter is depicted in Fig. 22. As the value of Wlf varies from 0.3 mm to 1.1 mm, the passband return loss gets improved and shows greater than 20 dB performance for Wlf = 1.1 mm.
Transmission Characteristics (dB)
0 -10 -20 -30 -40 Wlf = 0.3 mm
-50
Wlf = 0.7 mm Wlf = 1.1 mm
-60 0
2
4
6 8 Frequency (GHz)
10
12
Fig. 21. Simulated transmission characteristics of the proposed filter for various values of Wlf.
Reflection Characteristics (dB)
0 -10 -20 -30 -40 Wlf = 0.3 mm
-50
Wlf = 0.7 mm Wlf = 1.1 mm
-60 0
2
4
6 8 Frequency (GHz)
10
12
Fig. 22. Simulated reflection characteristics of the proposed filter for various values of Wlf.
THE EFFECT OF ‘Llf’ LENGTH OF THE RECTANGULAR PORTION OF UPPER
FUNNEL
SHAPED
RESONATOR
IN
TRANSMISSION
AND
REFLECTION CHARACTERISTICS OF THE PROPOSED FILTER. 0 Transmission Characteristics (dB)
4.4.
-5 -10 -15 -20 -25 -30
Llf = 8.15 mm Llf = 9.15 mm
-35
Llf = 10.15 mm
-40 0
2
4
6 8 Frequency (GHz)
10
12
14
Fig. 23. Simulated transmission characteristics of the proposed filter for various values of Llf.
Fig. 24. Simulated reflection characteristics of the proposed filter for various values of Llf.
As the length of the rectangular portion of the upper funnel shaped resonator increases from Llf = 8.15 mm to 10.15 mm, the upper and lower transmission zero positions got shifted to the right side as shown in Fig. 23. In the proposed design, we selected Llf = 9.15 mm because the stopband suppression level achieved is better than 20 dB for 9.15 mm. The variation of reflection characteristics with respect to Llf is shown in Fig. 24 and from this analysis, we concluded that the return loss performance obtained shows better performance for Llf = 9.15 mm.
5. RESULTS AND DISCUSSIONS The filter is simulated and optimized using the electromagnetic simulation software IE3D and fabricated using MITS 11 Auto lab fabrication machine. The measurements are taken using R&S ZVB 20 Vector Network Analyzer. For ensuring impedance matching, the input and output ports are properly matched. Since we are using the FR4 substrate with thickness 0.8 mm, the width of both ports must be 1.5 mm. The optimized design parameters of the proposed filter are as follows: L1 = 22.02, W1= 0.5, La = 7.67, Wa = 2.4, W2 = 0.3, Lwe = 3.64, Le= 2.45, We = 1, Wlf = 1.1, Llf = 9.15, Wuf = 4.78, Luf = 3.41, Duf = 3.85, Lfl= 0.73, Wbl = 1.4, Lbl = 1.25 and Wfl = 0.36. (All dimensions are in mm). The 3 dB centre frequency of the developed filter is 5.68 GHz. The 3 dB pass band has a fractional bandwidth of 97.57 %. The insertion loss (IL) at centre frequency is less than 1.62 dB. The minimum value of return loss (RL) obtained is better than 20 dB within the pass band, and sharp selectivity is observed because of the two transmission zeros obtained at the lower and upper passband edges. The simulated and measured transmission and reflection characteristics of the developed filter are shown in Fig. 25 (a).
10 S1
S21
S-Parameters (dB)
0 -10 -20 -30 -40
SL : Measured DL : Simulated
-50 0
2
4
6 8 Frequency (GHz)
10
12
14
Fig. 25. (a) Simulated and measured responses of the proposed filter [SL: solid line and D L: dotted line]
Transmission characteristics (dB)
10 S21
0 -10 -20 -30 -40 -50 0
2
4
6
8 10 12 14 Frequency (GHz)
16
18
20
Fig. 25. (b) Measured Wideband transmission characteristics of the developed filter
Even though the proposed filter exhibits upper stopband with greater than 20 dB suppression level upto 24.2 GHz, the measured wideband response is taken only upto 20 GHz because the maximum frequency range that can be measured using the available R&S ZVB 20 vector network analyzer is 20 GHz and the measured wideband response is given in Fig. 25 (b). TABLE. I. Comparison of the Proposed Filter with Existing Filters Dielectric
Centre
3- dB
frequency
FBW
(GHz)
(%)
[2]
5.96
85
10
[5]
4
67
[6]
4.175
[7]
Ref.
RL
constant, εr/
(dB)
thickness
Loss
Maximum
Upper
tangent,
IL
stopband
stopband
tan δ
(dB)
attenuation (dB)
(GHz)
4.4/1.6
0.02
2
15
20
<20
2.2/1.574
0.0009
NA
20
8.5
28.02
10
3.2/1.14
0.004
NA
20
7
3
80
30
2.65/1.5
0.0005
NA
20
7.5
[10]
5.3
-
<10
2.2/0.508
0.0009
NA
20
10
[13]
3.52
62.75
13
4.4/1.6
0.02
0.69
20
9.72
[14]
4.5
69.1
12.3
2.55/0.8
0.0029
1.4
13
12.5
Proposed
5.68
97.57
>20
4.4/0.8
0.02
1.62
>20
24.2
(mm)
TABLE. I shows the comparison between existing filters and the developed, the obtained parameters of the fabricated filter are flat group delay, very good reflection characteristics and a suppression level greater than 20 dB up to 24.2 GHz.
(a)
(b)
Fig. 26. Photograph of the fabricated filter prototype (a) Top view (b) Bottom view
The photographs of the fabricated filter prototype are given in Fig. 26 (a) and (b). The top view shows the asymmetric funnel shaped resonators and side stubs with proper input output coupling. The bottom view shows the two symmetric apertures cut on the ground plane of the structure for the enhancement of coupling. The normalized circuit size, NCS of the developed filter is 0.3157 λg2. The relative stopband bandwidth of the proposed filter is 79.87 %. The roll – off rate for the lower and upper passband of the filter is 72.24 dB/GHz and 19.01 dB/GHz respectively at 20 dB suppression level.
6. CONCLUSION In this paper, a low cost wideband bandpass filter with wide pass band, good skirt selectivity with very good passband reflection characteristics is proposed and developed. The main resonator consists of a symmetrical funnel shaped resonator loaded centrally to the uniform impedance line which contributes two transmission zeros at the passband edges. Innovative and simple equations are derived for finding out the transmission zero locations and it can be applied for the realization of sharp skirt selectivity filter designs. The developed structure is compact with a normalized circuit size of 0.3157 in terms of λg2. The out of band performance is very good with a suppression level greater than 20 dB up to 24.2 GHz. The low cost realization, very good reflection characteristics, wide passband and very good spurious suppression makes this filter a promising candidate for wideband wireless communication systems. REFERENCES 1. Hong J. S., Lancaster M. J. Microstrip Filters for RF/Microwave Applications. Wiley, New
York, 2001.
2. Jhariya D., Azad A. R., Mohan A., and Sinha M. Compact wideband bandpass filter using fish
spear shaped multimode resonator. Microwave and Optical Technology Letters; Vol. 57, No. 12, 2833-2837, 2015. https://doi.org/10.1002/mop.29437 3. Dey R., Sarkar M., Rana B. A novel band pass filter design and analysis using inductive via
series.
AEU
–
Int
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Electron
and
Commun
2019;
99:315-324.
https://doi.org/10.1016/j.aeue.2018.12.009 4. Sassi I., Talbi L., Hettak K. Compact bandpass filters based on linked hexagonal-omega
resonators. Microwave and Optical Technology Letters; Vol. 58. No. 5, 1049-1052, 2016. https://doi.org/10.1002/mop.29720 5. Chen C. P., Kato N., Anada T. Synthesis scheme for wideband filters consisting of three-
coupled- lines including the cross coupling between non-adjacent lines. IET Microw. Antennas Propag; Vol. 9, 1558-1566, 2015. https://doi.org/10.1049/iet-map.2014.0467 6. Ma X. B., Jiang T. Wideband bandpass filter with controllable bandwidth and high selectivity
using two different types of resonators. Microwave and Optical Technology Letters; Vol. 57, 1319 - 1323, Vol. 57, 2015. https://doi.org/10.1002/mop.29085 7. Li L., Li Z. F. Side coupled shorted microstrip line for compact quasi elliptic wideband
bandpass filter design.IEEE Microwave and Wireless Components Letters; Vol. 20, No. 6, 322-324, 2010. http://dx.doi.org/10.1109%2FLMWC.2010.2047516 8. Xia X., Cheng X., Chen F. and Deng X. Compact UWB bandpass filter with sharp roll-off
using APCL structure. IET Electronics Letters; Vol. 54, No. 13, 837-839, 2018. https://doi.org/10.1049/el.2018.1151
9. Luo X., Qian H., Ma J. G. and Li E. P., Wideband bandpass filter with excellent selectivity
using new CSRR based resonators. IET Electronics Letters; Vol. 46, No. 20, 2010. https://doi.org/10.1049/el.2010.1817 10. Wang H., Chu Q. X., Gong J. Q., A compact wideband microstrip filter using folded multiple
mode resonator.IEEE Microwave and Wireless Components Letters; Vol. 19, No. 5, 287-289, 2009. https://doi.org/10.1109/LMWC.2009.2017591 11. Mayani M. G., Asadi S., Pirhadi A., Mahani S. M., Design and analysis of a super compact
wide-band bandpass filter based on meta-material resonators; AEU – Int J Electron and Commun 2018. 97:79-84. https://doi.org/10.1016/j.aeue.2018.10.010 12. Taibi A., Trabelsi M., Slimane A., Belaroussi M. T., Raskin J. P., A novel design method for
compact UWB bandpass filters.IEEE Microwave and Wireless Components Letters; Vol. 25, No. 1, 4-6, 2015. https://doi.org/10.1109/LMWC.2014.2363016 13. Yechou L., Tribak A., Kacim M., Zbitou J., Sanchez AM. A novel wideband bandpass filter
using coupled lines and T- shaped transmission lines with wide stopband on low cost substrate. Progress
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doi:10.2528/PIERC16062204 14. Zhang Z. C., Liu H. W., Wong S. W., Compact wideband bandpass filter based on double -T-
shaped stub loaded resonator and loading technique for zero -voltage point. Int. J RF Microw Comput Aided Eng.;2018 https://doi.org/10.1002/mmce.21197 15. Ghazali A. N., Sazid M., Virdee B., A compact UWB-BPF based on microstrip to cpw
transition with multiple transmission zeros. Microwave and Optical Technology Letters; Vol. 60, 1925-1928, 2018. https://doi.org/10.1002/mop.31274 16. Pozar D. M., Microwave Engineering, 4th ed. USA, Wiley, 2011.
17. Li Y., Li W., Liu C. and Ye Q., A Compact UWB Bandpass Filter with ultra - narrow tri- notch
band characteristic. ACES journal, Vol. 29, No. 2, February 2014 18. Li Y., Li W., Yu W. and Liu C., A miniaturization Band-pass Filter with ultra-narrow multi-
notch band characteristic for ultra wideband communication applications. ACES journal, vol. 29, No. 4, April 2014 19. Chu Q. X., Wu X. H. and Tian X. K., Novel UWB Bandpass Filter Using Stub-
Loaded Multiple-Mode Resonator. IEEE Microwave and Wireless Components Letters, vol. 21, no. 8, pp. 403-405, Aug 2011. https://doi.org/10.1109/LMWC.2011.2160526 AUTHOR BIOGRAPHIES
Ami Iqubal was born in Kerala, India in May,1988. She received the B. Tech in Electronics and Communication Engineering from CUSAT,Kerala in 2010; received M.Tech in Applied Electronics from Mahatma Gandhi University, Kerala in 2013 and currently working towards Ph. D degree in Electronics Engineering at CUSAT. Her research interests include the design ,analysis and development of microwave filters for wideband applications.
Abdulla P received the B. Tech in Eletrical and Electronics Engineering from University of Calicut, Kerala in 1991; received M.Tech in Electronics from CUSAT, Kerala in1993 and Ph.D degree in Electronics and Communication Engineering from IIT Kharagpur in 2009. He is currently working as a Professor in Division of Electronics Engineering, School of Engineering, Cochin University of Science and Technology. His areas of interst include design and development of microwave filters, Antennas and Computational Electromagnetics. He served as the Secretary IEEE Antenna Prpagation Society (APS), Kerala Section and currently the Vice Chair of APS, Kerala Section. He has more than 60 publications in his credit both international journals and conferences.
7. BANDPASS FILTER BASED ON ASYMMETRIC FUNNEL SHAPED RESONATORS WITH ULTRA WIDE UPPER STOPBAND CHARACTERISTICS Ami Iqubal and Abdulla P. Division of Electronics, School of Engineering, CUSAT, Cochin-22, Kerala, India. Corresponding author: [email protected] LIST OF FIGURES AND TABLES
Fig. 2. (a) Geometry (b) LC equivalent circuit
0
S-parameters (dB)
-5 -10 -15 -20 -25 -30
S 11
-35
S 12
-40 0
2
4
6
8
Frequency (GHz)
10
12
Fig. 2. Simulated frequency response of the basic funnel shaped resonator
Fig. 3. Basic upper funnel shaped resonator with plane of symmetry A-A'
14
Fig. 4. Equivalent transmission line model of the funnel shaped resonator
Fig. 5. Odd mode equivalent circuit of the basic funnel shaped resonator
Fig. 6. Even mode equivalent circuit of the basic funnel shaped resonator
S -Parameters (dB)
0 -10 -20 -30 S11
-40
S12 -50 0
(b)
2
4
6
8
10
12
14
Frequency (GHz)
16
18
(b)
Fig. 7. (a) Basic EBG loaded structure (b) Frequency response of the EBG loaded structure
20
S - Parameters (dB)
0
-10
-20
-30
S11 S12
-40 0
2
4
6
8
10
12
14
16
18
20
Frequency (GHz) (b)
(b)
Fig. 8. (a) EBG loaded funnel structure (b) response of the EBG loaded funnel structure
Fig. 9. Layout of the basic coupling structure
S - Parameters (dB)
0 -10 -20 -30 -40
S11 S12
-50 0
2
4
6
8
10
12
14
16
18
20
Frequency (GHz) Fig. 10. Simulated response of the basic coupling structure for the range (0-20 GHz)
0
S - Parameters (dB)
-10 -20 -30 -40
S11 S12
-50 -60 0
(b)
2
4
6
8 10 12 14 Frequency (GHz)
16
18
(b)
Fig. 11. (a) Layout of the coupled EBG loaded funnel structure (b) Frequency response
20
S - Parameters (dB)
0
-10
-20
-30 S11 S12
-40 0
2
4
6
8
10
12
14
16
18
20
Frequency (GHz) (b)
(b) Fig. 12. (a) Layout (b) Frequency response
0
S - Parameters (dB)
-10 -20 -30 -40 -50
S11 S12
-60 0
(b)
2
4
6
8 10 12 14 Frequency (GHz)
16
18
(b)
Fig. 13. (a) Layout (b) Frequency response of the double funnel shaped stub loaded filter
20
S - Parameters (dB)
0 -10 S11 S12
-20 -30 -40 -50 0
2
4
6
8
`
10 12 14 16 18 20 22 24 Frequency (GHz)
(b)
(b)
Fig. 14.(a)Layout of the final structure (b) Simulated response for the range (0-25GHz)
0 Transmission characteristics (dB)
Transmission characteristics (dB)
0 -10 -20 -30 -40 -50 -60 -70
without coupling with tight coupling
-80 -90 2
4
6 8 Frequency (GHz)
(b)
10
12
-20 -30 -40
with gpa
-50
without gpa
-60
-100 0
-10
14
0
2
4
6 8 10 Frequency (GHz)
12
14
(b)
Fig. 15. Response (a) with and without coupling (b) with and without aperture on ground (gpa)
Fig. 16. Electric field intensities at: (a) upper stopband @ 15 GHz (b) lower stopband @ 1 GHz (c) transmission zero 1@ 2.6 GHz (d) transmission zero 2 @ 9.1 GHz (e) centre frequency @ 5.68 GHz (f)
mode 1 @ 3.03 GHz (g) mode 2 @ 3.49 GHz(h) mode 3 @ 4.62 GHz (i) mode 4 @ 5.92 GHz (j)mode 5 @ 7.43 GHz (k) mode 6 @ 8.12 GHz
Transmission Characterstics (dB)
0 -5 -10 -15 -20 -25 Luf = 2.41 mm
-30
Luf = 3.41 mm
-35
Luf = 4.41 mm
-40 0
2
4
6 8 Frequency (GHz)
10
12
14
Fig. 17. Simulated transmission characteristics of the proposed filter for various values of Luf.
Reflection Characteristics (dB)
0 -10 -20 -30 -40 Luf = 2.41 Luf = 3.41
-50
Luf = 4.41
-60 0
2
4
6 8 Frequency (GHz)
10
12
14
Fig. 18. Simulated reflection characteristics of the proposed filter for various values of Luf.
Transmission Characteristics (dB)
0 -5 -10 -15 -20 -25
Wuf = 3.78 mm
-30
Wuf = 4.78 mm
-35
Wuf = 5.78 mm
-40 0
2
4
6 8 Frequency (GHz)
10
12
14
Fig. 19. Simulated transmission characteristics of the proposed filter for various values of Wuf.
Reflection Characteristics (dB)
0 -5 -10 -15 -20 -25
Wuf = 3.78smm
-30
Wuf = 4.78 mm
-35
Wuf = 5.78 mm
-40 0
2
4
6 8 Frequency (GHz)
10
12
14
Fig. 20. Simulated reflection characteristics of the proposed filter for various values of Wuf.
Transmission Characteristics (dB)
0 -10 -20 -30 -40
Wlf = 0.3mm Wlf = 0.7mm
-50
Wlf = 1.1mm
-60 0
2
4 6 8 Frequency (GHz)
10
12
Fig. 21. Simulated transmission characteristics of the proposed filter for various values of Wlf.
Reflection Characteristics (dB)
0 -10 -20 -30 -40 Wlf = 0.3 mm
-50
Wlf = 0.7 mm Wlf = 1.1 mm
-60 0
2
4
6 8 Frequency (GHz)
10
12
Fig. 22. Simulated reflection characteristics of the proposed filter for various values of Wlf.
Transmission Characteristics (dB)
0 -5 -10 -15 -20 -25 Llf = 8.15 mm
-30
Llf = 9.15 mm
-35
Llf = 10.15 mm
-40 0
2
4
6 8 Frequency (GHz)
10
12
14
Fig. 23. Simulated transmission characteristics of the proposed filter for various values of Llf.
Fig. 24. Simulated reflection characteristics of the proposed filter for various values of Llf.
10 S1
S21
S-Parameters (dB)
0 -10 -20 -30 -40
SL : Measured DL : Simulated
-50 0
2
4
6 8 Frequency (GHz)
10
12
14
Fig. 25. (a) Simulated and measured responses of the proposed filter [SL: solid line and D L: dotted line]
Transmission characteristics (dB)
10 S21
0 -10 -20 -30 -40 -50 0
2
4
6
8 10 12 14 Frequency (GHz)
16
18
20
Fig. 25. (b) Measured Wideband transmission characteristics of the developed filter
TABLE. I. Comparison of the Proposed Filter with Existing Filters
Dielectric
Centre
3- dB
frequency
FBW
(GHz)
(%)
[2]
5.96
85
10
[5]
4
67
[6]
4.175
[7]
RL
constant, εr/
(dB)
thickness
Loss
Maximum
Upper
tangent,
IL
stopband
stopband
tan δ
(dB)
attenuation (dB)
(GHz)
4.4/1.6
0.02
2
15
20
<20
2.2/1.574
0.0009
NA
20
8.5
28.02
10
3.2/1.14
0.004
NA
20
7
3
80
30
2.65/1.5
0.0005
NA
20
7.5
[10]
5.3
-
<10
2.2/0.508
0.0009
NA
20
10
[13]
3.52
62.75
13
4.4/1.6
0.02
0.69
20
9.72
[14]
4.5
69.1
12.3
2.55/0.8
0.0029
1.4
13
12.5
Proposed
5.68
97.57
>20
4.4/0.8
0.02
1.62
>20
24.2
Ref.
(a)
(mm)
(b)
Fig. 26. Photograph of the fabricated filter prototype (a) Top view (b) Bottom view
Declaration of interests
☒ The authors declare that they have no known competing financial interests or personal relationships that could have appeared to influence the work reported in this paper.
☐The authors declare the following financial interests/personal relationships which may be considered as potential competing interests: