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Design and performance of the readout electronics for the CsI(Tl) detector
NUCLEAR lNSTRUMENlS 8METNoDS IN PNYSBCS RESEARCH
ELSEWIER
Nuclear
Instruments
and Methods
in Physics
Research
A 411 (1998) 437-448
Section A
Design and performance of the readout electronics for the CsI(T1) detector Yu.G. Kudenkoa**, O.V. Mineev”, J. Imazatob aInstitute,for Nuclear Research RAS, 6li-th October Rerol. pr. 7a. MOSCOUI 17312, Russia hln.stitute o/‘Particle and Nuclear Studies. High Enera Accelerutor Research Orgunixtion (K&Y), Tsukuha. Iboruki 305. Jupan Received 29 October
I997
Abstract
This paper describes a readout electronics system constructed for the CsI(TI) calorimetry of neutral pions in experiment E246 at KEK. The instrumentation was designed to cover a wide range of energies (from 300 keV to 300 MeV) with high measurement accuracy under a counting rate of a few ten kHz per channel. The readout is provided by one 18 x 18mm’ (or 28 x 28mm’) Hamamatsu PIN-photodiode coupled directly to the rear end of each Csf(T1) crystal. A charge-sensitive preamplifier was developed specifically to have low noise for a high photodiode capacitance. The equivalent noise charge of the photodiode-preamplifier system at a 2 ps shaping time was found to be 640 and 840 electrons for 18 x 18 and 28 x 28 mm’ photodiodes, respectively. A pulse processor consists of a shaping amplifier with two outputs to cover the large dynamic range and a timing filter amplifier. A baseline restorer incorporated into the shaping amplifier retains good energy resolution under rates up to 50 kHz. A multichannel constant-fraction discriminator was designed to accept long risetime signals. A time resolution of 11.4 ns (fwhm) has been achieved in beam runs for the photon energy range from 10 to 200MeV. I,? 1998 Elsevier Science B.V. All rights reserved. PAC.!? 39.4O.Vj; 07.50.Q~ Kejjwor-ds: Photodiodes:
Constant-fraction
discriminator;
Baseline restorer
no and
1. Introduction
regions
The goal of the KEK-E246 collaboration is the search for T-violating muon polarization (PT) in the decay K -+ x”p+v [l], The unique feature of this experiment is a complete detection of the
*Correspondingauthor. Tel.: + 7095 3340184; 095 33401 84: e-mail: kudenko~wocup.inr.troitsk.ru.
fax:
+ 7
0168-9002~98/$19.00 ii;) 1998 Elsevier Science B.V. All rights reserved PII: SO1 5x-9002(98)00273-3
p+ kinematics. By using the kinematical producing opposite effects in PI., a double
ratio between the forward- and backward-going x0 events significantly reduces systematic errors. The basic principles of the experiment and set-up are described in Refs. [1,2]. The energy and direction of 7~’ are measured by a highly segmented electromagnetic calorimeter consisting of 768 CsI(T1) crystals assembled in a tight barrel [2]. The calorimeter is placed in the central region of the
toroidal magnet, where a magnetic field and limited space stipulate using large-area silicon PIN photodiodes to detect the scintillation light. A CsI(TI)/ photodiode system has some advantages: (1) immunity from magnetic fields; (2) compact size; (3) matched spectral response; (4) low operating volrage; (5) inherent stability and long lifetime of PINdiodes. However, it also leads to complications d electronics readout and proWSsing the connected with the noise problem C3,4-jand poor timing. In general, the electronics for the photon detector must provide high energy resolution over a wide range of 0.3-3OOMeV and m~intaiu the resolution under a counting rate up to 30 kHz. per crystal. Electrical noise is the main factor which deteriorates the resolution in the low~e~er~y region ( -c tOMeV). Taking into account that an electron-photon shower spreads over several crystals, the naise charge is summed by an algorithm which depends on the origin of the noise. Thermal and current shot noise are added ~uad~tically~ white pick-up noise co~tr~b~t~s IinearIy in the total noise. For the optimum sig~aI-tortoise ratio the signal should be processed by a pulse~shapin~ circuit which, however, has to constrain the pulse to a finite duration in order to avoid any pile-up, In order to achieve good energy resolution in the photon detector, a low-noise preamplifier and shaping amplifier have been designed and tested. The time resolution of the detector needs to be
better than 15 ns (fwhm) within the energy range of 10-300 MeV to suppress any accidental hits, which make the identification of 7~’ difficult under an intensive background. The timing performance is provided by an appropriate shaping in the timing filter amplifier, and then discriminating by the CFD module, which was specially designed to accept slow risetime signals.
The scheme of the crystal readout is shown in Fig. 1. A silicon PIE-photodiodc (I’D), glued on the rear face of each crystal by an elastic silicone glue, converts the scintillation light into an clectrical signal. The system for each channel consists of a low-noise preampIi~er mounted directly bebind the crystal, the main amplifier with low- and highgain outputs, a eonsta~t-fraction disc~minat~~r (CFD), a common stop TDC (0.7 ns resol~tio~)~ a time digitizer (TD) and a yak-sensitive AIX. The 32-channel FASTBUS TD module is based on a switched capacitor array integrated circuit, which allows waveform recording to be performed with 12-bit accuracy at a sampling rate of 10 MHz. A detailed description of the TD schematic is given in Ref. [SJ. The photodiode performances, such as the junction capacitance and Leakage current, dominate the
Fig. X, Electronics block diagram for the Csl detector.
+ bias voltage
-12v Fig. 2. Preamplifier scheme.
overall noise in the case of the large-area photodiodes: HAMAMATSU S3584-05 (28 x 28mm’ sensitive area, C,, = 200pF, fn < 100nA) and S3204-03 ( 18 x 18 mm’, Cpn = 140 pF, In < 20 nA). Although the larger S3.584-05 gives a better signalto-noise ratio [3], and also reduces the pick-up noise interference, they are applied only for the largest crystals due to the cost considerations. A charge-sensitive preamplifier (Fig. 2) has been developed for our purpose by Clear Pulse Co. [6]. basing on a commercial product CS.507. It is assembled with two hybrid ICs (a charge-sensitive preamp and a buffer amplifier) into a compact Al-box of 33 x 40 x 20mm”, thereby minimizing the possibility of noise pick-up. To keep a small size and save deficient space, no connector was used in the preamplifier. A &wire shielded cable was soldered directly inside the preamp case and drawn out of the detector to a distributor board mounted on the front and rear end plates of the barrel crystal holder. The cable length varies from 250 to 800mm. depending on the crystal position in the barrel. The preamp rise time of 150ns does not affect the output signal, since the signal rise time is dominated by the 900ns decay constant of CsI(T1). The anticipated event rate and the average signal amplitude dictated the choice of the feedback constants in order to avoid overloading caused by the effect of pile-up. The preamplifier features ac-coupling to a photodiode, an integration time constant of 660~s (330 MQ x 2pF) and the ability to drive a 100 R twisted-pair cable. A pole-zero cancellation
network (PZC) shortens the long fall time of the integrated pulse to 49ps just before the output amplifier (gain of 8). The output is capacitively coupled and has a saturation limit of - 2.2 V. The overall gain of the preamplifier is 2 V/PC at 1OOfz load. resulting in a conversion coefficient of about 2.4mV/MeV for a single CsI(TI) module. A moderate power dissipation of 0.25 W per a single preamplifier (200 W for the whole detector) causes a temperature gradient of 8°C over the ambient air temperature in confined space around a photodiode. Relatively small heat cycles may lead to a degradation of the mechanical and optical contact between a PD and the crystal surface in the case of the hard optical glues used. An elastic silicone glue, which provides a more temperaturestable contact, was utilized to provide the crystal-PD contact. Nevertheless. cool dry air is pumped through the crystal barrel to keep the temperature and humidity at a safe level. The noise of the photodiode-preamplifier system was measured by irradiating the diodes with 59.5 keV y-rays from a “I Am source. The spread of the total absorption peak (1.64 x 10” e-h pairs) indicates the noise level directly in a photoelectron number [4]. Using the Ortec 573 amplifier with a shaping time of 2 us. the equivalent noise charge (ENC) at room temperature has been found to be 640 i 10 and 840 + 10 electrons for S3204-03 and S3584-05 photodiodes, respectively. The noise rises to 690 electrons for PD (18 x I8 mm’) at a reduced shaping time of 1 vs. A typical Csl(T1) large crystal
Yu.G. Kudenko et al. /Nucl. Instr. and Meth. in Phys. Res. A 411 (19981 437-448
440
(photoyield of lO”ph-e/MeV), attached to the PD(18 x 18 mm2) and the preamplifier, displays an excellent noise level of crE= 70 keV, even at such a shorter than optimum shaping time.
3. Main amplifier 3.1. Concept of the main amplifier The concept of the main amplifier was devetoped and implemented in the hybrid circuit design to meet the demands imposed by the photon detector. A 16-channel NIM module incorporates the shaping amplitier (SA) with two unipolar outputs and a timing filter amplifier (TFA). The low- and highgain outputs of SA allow it to work within a large dynamic range terminated by the highest energy of 300 MeV, deposited in a CsIfTl) crystal. The lowest border of the range is defined by the signal-to-noise ratio. High quality crystals coupled with the lownoise preamplifiers retain noise at the 300 keV level (minimal energy we are able to separate from the noise fluctuations). In this case, we have a dynamic range of 1000, which is not covered by the available
12-bit ADC. The high-gain output facilitates a calibration procedure with low-energy y-sources without any gain adjustment. The main parts of the SA are a differential receiver, a variable PZC circuit, a second-order active filter for the low-gain output and a fourth-order filter for the high-gain output, baseline restorer and output buffers. The baseline restorer (BLR) is employed to compensate the effect of a baseline shift caused by uncancelled ACinterstages in the preamplifier and undercompensation of the PZC network. In comparison with previously published circuits [7-93, the new BLR features an extremely simple design and fast restoration to the base level. The TFA forms a timing pulse in an optimum way to attain the best available rise time while relaxing the effect of the pulse front jitter caused by noise ffuctuations. 3.2. Shaping amplijer
A circuit diagram of the SA is shown in Fig. 3. The input to the main amplifier is a differential receiver stage, after which the output to the TFA branches off. Shaping is done in the next two stages
2Ok
ALL
design
OF AMPS: MX1197
Fig. 3. Circuit diagram of the shaping amplifier.
1Ok
by differentiation with an adjustable PZC network and integration with an active second-order lowpass filter. The filter parameters were chosen in an empirical way to optimize the energy resolution while keeping the pulse width as short as possible in order to handle high counting rate. The corresponding shaping time is 1 ps, and the jumpers on the circuit board provide an option to reduce the shaping time to 0.5 ps. The output waveforms of the CsI signal for several shaping times are shown in Fig. 4. The filter is followed by a variable-gain stage. The gain is adjusted by a fine-gain resistor through the front panel and by jumpers located on the printed board. A 1OOOpF capacitor separates the gain stage from the low-gain output section that consists of a buffer amplifier and the BLR. The BLR circuit is set as a negative feedback over the amplifier, and includes a fast BLR op amp, switching diodes and a separating 1OOOpFcapacitor. A 1 MR resistor provides a bias current to the non-inverting input of the buffer op amp. The separating capacitor transmits a negative polarity signal with a time constant of 1 ms exhibiting negligible distortion. Positive overshoot. inverted and ampli~ed by the BLR op amp, causes an opening of
the diode set in parallel to the 1 MR resistor so that the time constant drops to a value of less than 0,5~.ls, determined by the resistance of the open diode. Thus. negative signals pass through to the output being unaffected. while a positive level is compelled to return to the baseline level. The diodes in the BLR op amp feedback and a 1 kSZ resistor define the slew rate of this returning. The components were selected to provide fast BLR response while retaining a satisfactory resolution and linearity for small ~~mplitudes, which are sensitive to the BLR bypass effects. The low-gain output section is followed by a high-gain output section. which combines an amplifier with a fixed gain of 12 and a second-order lowpass filter. The filter makes the actual gain to be dependent on the pulse shape: for CsI(T1) signals the high-to-low gain ratio is 9.8. The buffers for both high- and low-gain outputs are designed to drive a twisted pair 1OOfl cable. The maximum output voltage is limited to 2 V by a + 6 V power supply used to match the input range of the peak-hold ADC. The high speed dual op amps CX20197(Sony) were used for each stage. The op amps are unity gain stable and have GBW = 40 MHz.
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YuG. Kudenko et al. /Nucl. Instr. and Meth. in Phys Res. A 41 I (1998) 437-448
3.3. Timing Jilter ampli$er
The TFA forms a timing signal for subsequent discrimination by a CFD module. A circuit diagram is shown in Fig. 5. Shaping is done by double differentiation (rdirr = 7SOns) and double integration (rint = 112 ns). Such a solution allows the noise influence to be minimized while keeping the timing pe~o~ances of the leading edge. While the risetime of the signal after preamplifier is about 2 us, TFA shortens it to 450ns. The gain adjustment is carried out through the front panel by a variable resistor. A jumper located on the printed board gives the option to change the gain by a factor of 2. The output resistors (22Q) restrict the maximum output voltage to - 6.9 V so as to fit it for the CFD input range.
4. Constant-fraction discriminator 4.1. Time derivation methods The CsI(T1) detector identifies rr” using the time coincidence between two photons from the no decay. A good time resolution provides a suppression of accidental events, which can seriously deteriorate pion identification under an intense
background. The estimated background level in E246 dictates the time resolution to be better than 15 ns (fwhm) for an energy range of 10-250 MeV. The time spread is caused by such effects as time walk - dependence on a signal amplitude variation; time jitter - statistical fluctuations of a photoelectron yield; noise-induced jitter of a signal front; dependence of comparator response time on input signal slope. The last source can dominate for those signals with a risetime stretching to a few hundred nanoseconds, as in our case of 450 ns. To minimize this effect, the trigger point should be arranged at the high slope of the signal through the zero crossing. The optimum trigger level chosen at a certain fraction of the pulse height results in the best accuracy. The noise influence is relaxed by filtering in the TFA. Zero-crossing discriminators fix the triggering point at the top of the pulse [lo]. While the zero-crossing technique overcomes the walk problem, this method is inherently subject to problems of statistical fluctuations and noise, especially for signals with long rise times. The extrapolated leading edge timing (ELET) method records the times at which the input front crosses two different
MA704A
Fig. 5. Circuit diagram of the timing filter amplifier.
+12 v
Yu.G. Kudenkff et al. iNm2
Instr. and Seth.
threshold levels, and then extrapolates them to the start point [ll]. Although the ELET technique is suitable to handle slow-rising pulses with varying risetimes, it requires an empirical correction for the real waveform signals. An ELET discriminator prototype has been designed and tested. Although there have been satisfactory results, the discriminator had been rejected because of the complexity and insufficient dynamic range. Taking into account everything reported above, the constant-fraction method [12] was chosen to design the CFD. 4.2. CFD design The l&channel CFD module was designed in the TKO, a standard package accepted at KEK for large-scale experiments [13]. Fig. 6 shows the details of the basic circuit. The input stage is a differential receiver-inverter (C)(20197), after which the signal is faned out into 3 parts. The first part is fed to a threshold comparator. The signal polarity is reversed to positive to conform the input range of the dual comparator AD96687 ( - 2.5 to + 5 V). The other parts are furnished on the timing comparator with one part delayed for 300ns by a distributed passive delay line ZMIO-300 (Showa Electric Wire&Cable Co.). The delay line features
Fig. 6. Circuit
diagram
in Phm. Rex A 411 {I~~~~ 437-448
443
a DIP-case, lOOn impedance, a risetime of 35ns, and a temperature coefficient of 1OOppm. The last part is differentiated with rRc = 300 ns before being sent to the inverting input of the timing comparator. Thereby, both of the compared voltages become equal in the crossing point at high relative slope (Fig. 7). In a more detailed photo (Fig. 7(a)) one can see the small spires generated by a hysteresis network. The hysteresis loop through C,,, and Rhys prevents possible oscillations of the comparator output state at the small slew rate signal presented on the inputs. Since the input range is over the maximum ratings for AD96687, protective Zener diodes are set on the non-inverting inputs. Such an arrangement extends the dynamic range of the CFD as long as the trigger point occurs below the damped region. The variable resistors in the differentiation network allow to tune the trigger point and output delay within the range of 60 ns. The common threshold for all 16 channels is adjusted at the local mode by a trimmer through the front panel. In the common mode the threshold vohage is supplied from the TKO backplane. The output signal width is variable from 20 to 3001-1s. Two separate ECL outputs per channel are available via 40-pin connectors.
of the constant-fraction
discriminator
Fig. 7. Attenuated and delayed voltage signals at the comparator inputs: (a) detailed view of a small pulse: (b) total view of a large pulse. The crossing point occurs at the constant fraction of a delayed Isignal and defines the timing of an output logic pulse.
5. Test results 5.1. Shaping ampli$er
The optimization of light collection in the CsI(T1) module by the crystal surface treatment, the choice of reflector, and the obtained uniformity of the light yield along the crystal length are described in detail in Refs. [3,14]. Here, we employ the typical CsI module (25 cm length. 18 x I8 mm2 PIN diode, CPpreamplifier) for measurements of the SA response. A noncollimated “Na y-ray source (1.275 MeV) was used to measure the photoyield, energy resolution and noise characteristics. Table 1 summarizes the results, as well as the results for the Ortec572 amplifier, which are shown for comparison. The linearity of both outputs was checked with an Ortec Research Pulser. The low-gain output was linear within the measurement accuracy of a 12-bit ADC. The high-gain output showed a light nonlinearity in an area of less than 5% of its full range, which corresponds to 1 MeV in the energy scale. The nonlinearity is caused by an inadequate response (bypass feedback through stray elements) of the BLR-circuit, and does not evoke a noticeable error in the energy measurement. A random pulse generator was applied to reproduce the real waveform input signals for the main amplifier. The spectrum distortion of the low-gain output pulses was monitored as a function of the
Table 1 Shaping amplifier performances Amplifier
Ortec572. ssh = 0.5 ps Ortec572, T,~= 1ps Ortec.571. r,,, = I! I*S High gain output Low gain output
ENC Photoyield FWHM (p.e.) (p.e./MeV) at 1.27 MeV
770 680 650 670 700
8600 10200 12300
11000 10~0
Pulse width Ar,, I
(X)
(W)
12.3 14.7 11.3 14.1
6 8 13 s 7
counting rate. It has been found that the effect of pile-up was negligible for the peak resolution ( - 0.8% fwhm) at a random rate up to 50 kHz. Pile-up leads to losses of useful events. Under losses we consider the signals, whose shift from the generator pulse peak due to piIe-up was more than 5%. This limit was chosen rather arbitrary to evaluate the BLR performance. The fraction of lost pulses versus the counting rate is presented in Fig. 8. It is worth noting that the losses exceeded 40% even at 1OkHz rate in the tested amplifier without a RLR. 5.2. Constant-fraction
discriminator
An artificial pulse with a rise time of 430ns was formed to measure the time walk over the dynamic
Yu.G. Kudenko et al./Nucl.
0
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20
30
40
Ins@. and Meth. in Phw. Res. A 4/I
50
Random frequency,
1
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I
0.5
III!
/ / 1.0 INPUT
II
5.0 PULSE
Table 2 Time resolution
obtained
Energy
window
(MeVt
Integral 559 9913 15-19 2428 32 36 4&44 49953
(5100)
in the cosmic ray test FWHM
(ns)
12.6 59 40 ‘5.3 16.7
13.x Il.0 9.6
kHz
Fig. 8. Relative pile-up losses of a random pulse generator vs. counting rate. The losses are counted if the pulse height is shifted more than 5% of the peak value.
‘-1
445
(1998) 437-448
VOLTAGE,
V
Fig. 9. Output pulse time walk vs. input voltage in the CFD. The input pulse reproduced the real waveform of the CsI signal after TFA.
range of 1: 70. The maximum voltage was limited by the input range of the CFD. Since amplitudes below 100mV drastically deteriorate the time resolution, 90mV was fixed to be the low limit of the tested region. The time walk versus the input voltage is shown in Fig. 9. The walk is confined within 6 ns, contributing a relatively small part in the total time spread, as will be seen from a beam test. The raise of the right part of the curve is caused by waveform nonlinearity. The comparator dependence on the input slew rate is considered to be responsible for the gradual slope of the remaining part. Cosmic rays were employed to measure the time resolution in the appropriate energy range. A plas-
tic scintillator (100 x 200mm) provided the start pulse, while the stop one was generated by the CsI(T1) crystal. A time resolution of 12.6 ns (fwhm) has been achieved in the integral range from 4 MeV up to an energy extended over 100MeV with the electronic set: CP-preamplifier + TFA + CFD. The differential time resolution was measured within the energy windows, and the results are listed in Table 2. Shifting of the peak in the time spectrum was observed at small energies below 1S MeV. It is supposed to be caused not by the time walk, but by the difference in the ground potentials between the CFD and the TFA. A few millivolts, usually not noticed, resulted in the relative large time shift ( - 10 ns) due to the long rise time of discriminated signals. That kind of walk is suppressed by the alignment of the ground potentials.
6. Performance
in the beam
The CsI electronics system went into operation in October, 1995, when E246 first took data. During the subsequent 2 year period of operation, none of the 768 photodiodes failed. One preamplifier had become dead soon after detector operation, presumably due to bad assembly, thus confirming the fact that it is the most unreliable part of the readout electronics. The electronic noise was determined from the widths of the pedestals, and can be separated into coherent (correlated) and incoherent contributions. The incoherent contributions sum in quadrature, whereas the coherent ones combine linearly. The
446
Yu.G. Kudenko et al. jNuc1. Instr. and Meth. in Phys. Res. A 411 (1998) 437-448
3000
3500
4000
ADC channel Fig. 10. Typical pulse height spectrum of the Xe-lamp for a single. CsI crystal under a high intensity beam. The trigger was started by the Xe-puker.
noise contribution to the energy resolution (rr) obtained for a cluster of nine CsI crystals is 250 keV. The contribution to the noise from each crystal (incoherent) does not exceed 75 keV (a), and the coherent noise was found to be about 11 keV (a) per crystal. The correlated deposit increases the noise of a cluster by only 10%. A Xe-lamp was employed to test the baseline stability at high counting rates. The Xe-lamp peak position was monitored at different beam intensities. A typical pulse height spectrum for one crystal at a counting rate of about 20 kHz is shown in Fig. 10. The energy spectrum is distorted from the true Gaussian shape by piling up of the Xe-pulse with randomly occurring scintillating pulses from the beam. The peak position and resolution were preserved at a maximum measured counting rate of 32 kHz, where the counts were registered if the energy deposit in a crystal exceeded 10 MeV. We
calculated the losses by counting the Xe-lamp events whose energy shift from the peak was more than 8 MeV. Comparing this number with the total number of triggers generated by the Xe-pulser, we obtained the fraction of real events lost due to pile-up (Fig. it). It should be noted that a part of pile-up events will be saved in off-line analysis by fitting the signal waveform digitized by the TD. Since the average counting rate per crystal in our experiment is under 10 kHz, the pile-up does not create any significant loss of data during the final analysis. We consider a good time resolution as an important factor to provide reliable identification of neutral particles and to suppress the background. The time distribution for all crystals and the corresponding photon energy distribution from the rr” decay for the KP3 events recorded during beam runs are shown in Fig. 12(a) and (b), respectively.
Yu.G. Kudenko et al. /Nucl. Instr. and Meth. in Phvs. Res. .4 4/l
I
&@.&f+
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(1998) 437-448
447
I
I I
L-.-e..I
s
10
I5
21)
25
30
Counting rate,
35
kHz
Fig. 11. Relative piie-up losses of the Xe-lamp vs. the counting rate in a crystal. The Xe-lamp rate was fixed at 10 Hz. The losses are counted if the Xe-pulse height shift from the peak value is more than 8 MeV.
Shown in Fig. 12(a) is the time spectrum for the crystals which are the centers of the gamma clusters. Only events with energy deposited in a central crystal above a threshold of 10MeV are shown. The time peak is well fitted by a Gaussian with CI= 4.8 ns. Some asymmetry of the peak shape is explained by the time shift observed at small pulse amplitudes (low energies). As seen in Fig. 13, a time resolution of 3.5 ns at 1OOMeV deteriorates to 16.8ns in the low energy range of 10-20MeV. The cosmic ray test described in the previous chapter showed a better resolution at a similar energy deposited in the crystal. It stems from the high quality (large light yield) of the selected crystal under the cosmic test, while the data obtained in the beam combines all 768 CsI modules of the calorimeter. An incorrect peak alignment for different crystals aiso widens the total time distribution. Exploiting the obtained timing performance, the Csl signals from all CFD outputs were inserted in the trigger logic as the OR strobe. A 1501~swidth of a discriminator pulse guarantees the acceptance of all valuable events at a level of about loo%, while reducing the trigger rate by a factor of 2,
7. Conclusions The readout electronics for the photon detector has been developed specifically for the needs of the
1500
: 250
750
500 150
c (b)
,_h!TRGY, MeV
Fig. It. (a) Times~ctrum of all Csl crystals with aligned peak positions. One TDC channel = 0,711s. (b) DisIribu~~on of the energy deposited in a crystal (center of cluster) during beam runs.
experiment E246. The electronic system consists of the low-noise preamplifier, a main amplifier, a CFD, a time digitizer, a peak-sensitive ADC and a commercial TDC. The preamplifier in combination with a large-area PIN-photodiode provides excellent noise characteristics and also features a compact size. A typical noise level of 640 (840) electrons (rms) at a shaping time of 2l.t~ was
448
Yu.G. K%denko et al. /Nucl. Ins&. and Meth. in Phys. Res. A 411 (199S) 437-448
Acknowledgements s 3
16.0 -
The authors would like to thank K. Mori of Clear Pulse Co. for valuable technical suggestions, M. Aoki for helpful discussions, 0. Sasaki for cooperation in the CFD production. We would like also to thank all members of the E246 collaboration who participated in the beam runs. This work was done during a stay of one of the authors (O.M.) at KEK as a foreign researcher of the Japan Ministry of Education, Culture, Science and Sport. This work was supported in part by the Russian Foundation for Basic Research Grant No. 96-02-16081.
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Energy, MeV Fig. 13. Time resolution (rms) per energy window, AE = 10 MeV, as a function of the photon energy.
achieved with S3204-03 (S3584-05) photodiodes at room temperature. The 16ch. main amplifier combines the shaping amplifier with two outputs to cover the large dynamic range and the timing filter amplifier. The performances of the shaping amplifier are similar to that of an Ortec 572 amplifier with a shaping time of 1 us. Intrinsic jumpers give an option to halve the shaping time. A random rate test was performed in order to verify the new design of a baseline restorer. No baseline shift was observed under a counting rate of up to 50 kHz. A CFD unit adapted to input pulses with a long rise time has been designed. A time resolution of 11.4ns (fwhm) has been obtained in beam runs for a photon energy range from 10 to 200 MeV.
References [l] J. Imazato et al., KEK Report 91-S. November 1991. [2] D.V. Dementyev et al., Nucl. Instc. and Meth. A 379 (1996) 499. [3] Yu.G. Kudenko. J. Imazato, KEK Report 92-15. October 1992. [4] E. Gramsch et al., Nucl. In&r. and Meth. A 311 (1992) 529. [S] D.R. Marlow et al., Preprint, Princeton/HEP/95-10. [6] Clear Pulse Co., Ohta-ku, Tokyo, 143 Japan. [7] T. Taniguchi et al., IEEE Trans. Nucl. Sci. NS-36 (1)(1989) 657. [8] Clear Pulse Co., Technical manual to 4040 Filter Amplifier. [P] S. Ohkawa, K. Husimi, IEEE Trans. Nucl. Sei. NS-33 (1) (1986) 415. [lo] B.T. Turko, R.C. Smith, IEEE Trans. Nucl. Sci. NS-39 (5) (1992) 1311. [ll] B. Gottschalk, Nucl. Instr. and Meth. 190 (1981) 67. [12] EG%G ORTEC Detectors & Instruments for Nuclear Spectroscopy Catalog (1991/1992) 3-138. [I33 TKO specification. KEK Report 85-10, 1985. [14] M.P. Grigoryev et al., Instr. Exp. Tech. 39 (2) (1996) 164.
ELSEWIER
Nuclear
Instruments
and Methods
in Physics
Research
A 411 (1998) 437-448
Section A
Design and performance of the readout electronics for the CsI(T1) detector Yu.G. Kudenkoa**, O.V. Mineev”, J. Imazatob aInstitute,for Nuclear Research RAS, 6li-th October Rerol. pr. 7a. MOSCOUI 17312, Russia hln.stitute o/‘Particle and Nuclear Studies. High Enera Accelerutor Research Orgunixtion (K&Y), Tsukuha. Iboruki 305. Jupan Received 29 October
I997
Abstract
This paper describes a readout electronics system constructed for the CsI(TI) calorimetry of neutral pions in experiment E246 at KEK. The instrumentation was designed to cover a wide range of energies (from 300 keV to 300 MeV) with high measurement accuracy under a counting rate of a few ten kHz per channel. The readout is provided by one 18 x 18mm’ (or 28 x 28mm’) Hamamatsu PIN-photodiode coupled directly to the rear end of each Csf(T1) crystal. A charge-sensitive preamplifier was developed specifically to have low noise for a high photodiode capacitance. The equivalent noise charge of the photodiode-preamplifier system at a 2 ps shaping time was found to be 640 and 840 electrons for 18 x 18 and 28 x 28 mm’ photodiodes, respectively. A pulse processor consists of a shaping amplifier with two outputs to cover the large dynamic range and a timing filter amplifier. A baseline restorer incorporated into the shaping amplifier retains good energy resolution under rates up to 50 kHz. A multichannel constant-fraction discriminator was designed to accept long risetime signals. A time resolution of 11.4 ns (fwhm) has been achieved in beam runs for the photon energy range from 10 to 200MeV. I,? 1998 Elsevier Science B.V. All rights reserved. PAC.!? 39.4O.Vj; 07.50.Q~ Kejjwor-ds: Photodiodes:
Constant-fraction
discriminator;
Baseline restorer
no and
1. Introduction
regions
The goal of the KEK-E246 collaboration is the search for T-violating muon polarization (PT) in the decay K -+ x”p+v [l], The unique feature of this experiment is a complete detection of the
*Correspondingauthor. Tel.: + 7095 3340184; 095 33401 84: e-mail: kudenko~wocup.inr.troitsk.ru.
fax:
+ 7
0168-9002~98/$19.00 ii;) 1998 Elsevier Science B.V. All rights reserved PII: SO1 5x-9002(98)00273-3
p+ kinematics. By using the kinematical producing opposite effects in PI., a double
ratio between the forward- and backward-going x0 events significantly reduces systematic errors. The basic principles of the experiment and set-up are described in Refs. [1,2]. The energy and direction of 7~’ are measured by a highly segmented electromagnetic calorimeter consisting of 768 CsI(T1) crystals assembled in a tight barrel [2]. The calorimeter is placed in the central region of the
toroidal magnet, where a magnetic field and limited space stipulate using large-area silicon PIN photodiodes to detect the scintillation light. A CsI(TI)/ photodiode system has some advantages: (1) immunity from magnetic fields; (2) compact size; (3) matched spectral response; (4) low operating volrage; (5) inherent stability and long lifetime of PINdiodes. However, it also leads to complications d electronics readout and proWSsing the connected with the noise problem C3,4-jand poor timing. In general, the electronics for the photon detector must provide high energy resolution over a wide range of 0.3-3OOMeV and m~intaiu the resolution under a counting rate up to 30 kHz. per crystal. Electrical noise is the main factor which deteriorates the resolution in the low~e~er~y region ( -c tOMeV). Taking into account that an electron-photon shower spreads over several crystals, the naise charge is summed by an algorithm which depends on the origin of the noise. Thermal and current shot noise are added ~uad~tically~ white pick-up noise co~tr~b~t~s IinearIy in the total noise. For the optimum sig~aI-tortoise ratio the signal should be processed by a pulse~shapin~ circuit which, however, has to constrain the pulse to a finite duration in order to avoid any pile-up, In order to achieve good energy resolution in the photon detector, a low-noise preamplifier and shaping amplifier have been designed and tested. The time resolution of the detector needs to be
better than 15 ns (fwhm) within the energy range of 10-300 MeV to suppress any accidental hits, which make the identification of 7~’ difficult under an intensive background. The timing performance is provided by an appropriate shaping in the timing filter amplifier, and then discriminating by the CFD module, which was specially designed to accept slow risetime signals.
The scheme of the crystal readout is shown in Fig. 1. A silicon PIE-photodiodc (I’D), glued on the rear face of each crystal by an elastic silicone glue, converts the scintillation light into an clectrical signal. The system for each channel consists of a low-noise preampIi~er mounted directly bebind the crystal, the main amplifier with low- and highgain outputs, a eonsta~t-fraction disc~minat~~r (CFD), a common stop TDC (0.7 ns resol~tio~)~ a time digitizer (TD) and a yak-sensitive AIX. The 32-channel FASTBUS TD module is based on a switched capacitor array integrated circuit, which allows waveform recording to be performed with 12-bit accuracy at a sampling rate of 10 MHz. A detailed description of the TD schematic is given in Ref. [SJ. The photodiode performances, such as the junction capacitance and Leakage current, dominate the
Fig. X, Electronics block diagram for the Csl detector.
+ bias voltage
-12v Fig. 2. Preamplifier scheme.
overall noise in the case of the large-area photodiodes: HAMAMATSU S3584-05 (28 x 28mm’ sensitive area, C,, = 200pF, fn < 100nA) and S3204-03 ( 18 x 18 mm’, Cpn = 140 pF, In < 20 nA). Although the larger S3.584-05 gives a better signalto-noise ratio [3], and also reduces the pick-up noise interference, they are applied only for the largest crystals due to the cost considerations. A charge-sensitive preamplifier (Fig. 2) has been developed for our purpose by Clear Pulse Co. [6]. basing on a commercial product CS.507. It is assembled with two hybrid ICs (a charge-sensitive preamp and a buffer amplifier) into a compact Al-box of 33 x 40 x 20mm”, thereby minimizing the possibility of noise pick-up. To keep a small size and save deficient space, no connector was used in the preamplifier. A &wire shielded cable was soldered directly inside the preamp case and drawn out of the detector to a distributor board mounted on the front and rear end plates of the barrel crystal holder. The cable length varies from 250 to 800mm. depending on the crystal position in the barrel. The preamp rise time of 150ns does not affect the output signal, since the signal rise time is dominated by the 900ns decay constant of CsI(T1). The anticipated event rate and the average signal amplitude dictated the choice of the feedback constants in order to avoid overloading caused by the effect of pile-up. The preamplifier features ac-coupling to a photodiode, an integration time constant of 660~s (330 MQ x 2pF) and the ability to drive a 100 R twisted-pair cable. A pole-zero cancellation
network (PZC) shortens the long fall time of the integrated pulse to 49ps just before the output amplifier (gain of 8). The output is capacitively coupled and has a saturation limit of - 2.2 V. The overall gain of the preamplifier is 2 V/PC at 1OOfz load. resulting in a conversion coefficient of about 2.4mV/MeV for a single CsI(TI) module. A moderate power dissipation of 0.25 W per a single preamplifier (200 W for the whole detector) causes a temperature gradient of 8°C over the ambient air temperature in confined space around a photodiode. Relatively small heat cycles may lead to a degradation of the mechanical and optical contact between a PD and the crystal surface in the case of the hard optical glues used. An elastic silicone glue, which provides a more temperaturestable contact, was utilized to provide the crystal-PD contact. Nevertheless. cool dry air is pumped through the crystal barrel to keep the temperature and humidity at a safe level. The noise of the photodiode-preamplifier system was measured by irradiating the diodes with 59.5 keV y-rays from a “I Am source. The spread of the total absorption peak (1.64 x 10” e-h pairs) indicates the noise level directly in a photoelectron number [4]. Using the Ortec 573 amplifier with a shaping time of 2 us. the equivalent noise charge (ENC) at room temperature has been found to be 640 i 10 and 840 + 10 electrons for S3204-03 and S3584-05 photodiodes, respectively. The noise rises to 690 electrons for PD (18 x I8 mm’) at a reduced shaping time of 1 vs. A typical Csl(T1) large crystal
Yu.G. Kudenko et al. /Nucl. Instr. and Meth. in Phys. Res. A 411 (19981 437-448
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(photoyield of lO”ph-e/MeV), attached to the PD(18 x 18 mm2) and the preamplifier, displays an excellent noise level of crE= 70 keV, even at such a shorter than optimum shaping time.
3. Main amplifier 3.1. Concept of the main amplifier The concept of the main amplifier was devetoped and implemented in the hybrid circuit design to meet the demands imposed by the photon detector. A 16-channel NIM module incorporates the shaping amplitier (SA) with two unipolar outputs and a timing filter amplifier (TFA). The low- and highgain outputs of SA allow it to work within a large dynamic range terminated by the highest energy of 300 MeV, deposited in a CsIfTl) crystal. The lowest border of the range is defined by the signal-to-noise ratio. High quality crystals coupled with the lownoise preamplifiers retain noise at the 300 keV level (minimal energy we are able to separate from the noise fluctuations). In this case, we have a dynamic range of 1000, which is not covered by the available
12-bit ADC. The high-gain output facilitates a calibration procedure with low-energy y-sources without any gain adjustment. The main parts of the SA are a differential receiver, a variable PZC circuit, a second-order active filter for the low-gain output and a fourth-order filter for the high-gain output, baseline restorer and output buffers. The baseline restorer (BLR) is employed to compensate the effect of a baseline shift caused by uncancelled ACinterstages in the preamplifier and undercompensation of the PZC network. In comparison with previously published circuits [7-93, the new BLR features an extremely simple design and fast restoration to the base level. The TFA forms a timing pulse in an optimum way to attain the best available rise time while relaxing the effect of the pulse front jitter caused by noise ffuctuations. 3.2. Shaping amplijer
A circuit diagram of the SA is shown in Fig. 3. The input to the main amplifier is a differential receiver stage, after which the output to the TFA branches off. Shaping is done in the next two stages
2Ok
ALL
design
OF AMPS: MX1197
Fig. 3. Circuit diagram of the shaping amplifier.
1Ok
by differentiation with an adjustable PZC network and integration with an active second-order lowpass filter. The filter parameters were chosen in an empirical way to optimize the energy resolution while keeping the pulse width as short as possible in order to handle high counting rate. The corresponding shaping time is 1 ps, and the jumpers on the circuit board provide an option to reduce the shaping time to 0.5 ps. The output waveforms of the CsI signal for several shaping times are shown in Fig. 4. The filter is followed by a variable-gain stage. The gain is adjusted by a fine-gain resistor through the front panel and by jumpers located on the printed board. A 1OOOpF capacitor separates the gain stage from the low-gain output section that consists of a buffer amplifier and the BLR. The BLR circuit is set as a negative feedback over the amplifier, and includes a fast BLR op amp, switching diodes and a separating 1OOOpFcapacitor. A 1 MR resistor provides a bias current to the non-inverting input of the buffer op amp. The separating capacitor transmits a negative polarity signal with a time constant of 1 ms exhibiting negligible distortion. Positive overshoot. inverted and ampli~ed by the BLR op amp, causes an opening of
the diode set in parallel to the 1 MR resistor so that the time constant drops to a value of less than 0,5~.ls, determined by the resistance of the open diode. Thus. negative signals pass through to the output being unaffected. while a positive level is compelled to return to the baseline level. The diodes in the BLR op amp feedback and a 1 kSZ resistor define the slew rate of this returning. The components were selected to provide fast BLR response while retaining a satisfactory resolution and linearity for small ~~mplitudes, which are sensitive to the BLR bypass effects. The low-gain output section is followed by a high-gain output section. which combines an amplifier with a fixed gain of 12 and a second-order lowpass filter. The filter makes the actual gain to be dependent on the pulse shape: for CsI(T1) signals the high-to-low gain ratio is 9.8. The buffers for both high- and low-gain outputs are designed to drive a twisted pair 1OOfl cable. The maximum output voltage is limited to 2 V by a + 6 V power supply used to match the input range of the peak-hold ADC. The high speed dual op amps CX20197(Sony) were used for each stage. The op amps are unity gain stable and have GBW = 40 MHz.
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3.3. Timing Jilter ampli$er
The TFA forms a timing signal for subsequent discrimination by a CFD module. A circuit diagram is shown in Fig. 5. Shaping is done by double differentiation (rdirr = 7SOns) and double integration (rint = 112 ns). Such a solution allows the noise influence to be minimized while keeping the timing pe~o~ances of the leading edge. While the risetime of the signal after preamplifier is about 2 us, TFA shortens it to 450ns. The gain adjustment is carried out through the front panel by a variable resistor. A jumper located on the printed board gives the option to change the gain by a factor of 2. The output resistors (22Q) restrict the maximum output voltage to - 6.9 V so as to fit it for the CFD input range.
4. Constant-fraction discriminator 4.1. Time derivation methods The CsI(T1) detector identifies rr” using the time coincidence between two photons from the no decay. A good time resolution provides a suppression of accidental events, which can seriously deteriorate pion identification under an intense
background. The estimated background level in E246 dictates the time resolution to be better than 15 ns (fwhm) for an energy range of 10-250 MeV. The time spread is caused by such effects as time walk - dependence on a signal amplitude variation; time jitter - statistical fluctuations of a photoelectron yield; noise-induced jitter of a signal front; dependence of comparator response time on input signal slope. The last source can dominate for those signals with a risetime stretching to a few hundred nanoseconds, as in our case of 450 ns. To minimize this effect, the trigger point should be arranged at the high slope of the signal through the zero crossing. The optimum trigger level chosen at a certain fraction of the pulse height results in the best accuracy. The noise influence is relaxed by filtering in the TFA. Zero-crossing discriminators fix the triggering point at the top of the pulse [lo]. While the zero-crossing technique overcomes the walk problem, this method is inherently subject to problems of statistical fluctuations and noise, especially for signals with long rise times. The extrapolated leading edge timing (ELET) method records the times at which the input front crosses two different
MA704A
Fig. 5. Circuit diagram of the timing filter amplifier.
+12 v
Yu.G. Kudenkff et al. iNm2
Instr. and Seth.
threshold levels, and then extrapolates them to the start point [ll]. Although the ELET technique is suitable to handle slow-rising pulses with varying risetimes, it requires an empirical correction for the real waveform signals. An ELET discriminator prototype has been designed and tested. Although there have been satisfactory results, the discriminator had been rejected because of the complexity and insufficient dynamic range. Taking into account everything reported above, the constant-fraction method [12] was chosen to design the CFD. 4.2. CFD design The l&channel CFD module was designed in the TKO, a standard package accepted at KEK for large-scale experiments [13]. Fig. 6 shows the details of the basic circuit. The input stage is a differential receiver-inverter (C)(20197), after which the signal is faned out into 3 parts. The first part is fed to a threshold comparator. The signal polarity is reversed to positive to conform the input range of the dual comparator AD96687 ( - 2.5 to + 5 V). The other parts are furnished on the timing comparator with one part delayed for 300ns by a distributed passive delay line ZMIO-300 (Showa Electric Wire&Cable Co.). The delay line features
Fig. 6. Circuit
diagram
in Phm. Rex A 411 {I~~~~ 437-448
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a DIP-case, lOOn impedance, a risetime of 35ns, and a temperature coefficient of 1OOppm. The last part is differentiated with rRc = 300 ns before being sent to the inverting input of the timing comparator. Thereby, both of the compared voltages become equal in the crossing point at high relative slope (Fig. 7). In a more detailed photo (Fig. 7(a)) one can see the small spires generated by a hysteresis network. The hysteresis loop through C,,, and Rhys prevents possible oscillations of the comparator output state at the small slew rate signal presented on the inputs. Since the input range is over the maximum ratings for AD96687, protective Zener diodes are set on the non-inverting inputs. Such an arrangement extends the dynamic range of the CFD as long as the trigger point occurs below the damped region. The variable resistors in the differentiation network allow to tune the trigger point and output delay within the range of 60 ns. The common threshold for all 16 channels is adjusted at the local mode by a trimmer through the front panel. In the common mode the threshold vohage is supplied from the TKO backplane. The output signal width is variable from 20 to 3001-1s. Two separate ECL outputs per channel are available via 40-pin connectors.
of the constant-fraction
discriminator
Fig. 7. Attenuated and delayed voltage signals at the comparator inputs: (a) detailed view of a small pulse: (b) total view of a large pulse. The crossing point occurs at the constant fraction of a delayed Isignal and defines the timing of an output logic pulse.
5. Test results 5.1. Shaping ampli$er
The optimization of light collection in the CsI(T1) module by the crystal surface treatment, the choice of reflector, and the obtained uniformity of the light yield along the crystal length are described in detail in Refs. [3,14]. Here, we employ the typical CsI module (25 cm length. 18 x I8 mm2 PIN diode, CPpreamplifier) for measurements of the SA response. A noncollimated “Na y-ray source (1.275 MeV) was used to measure the photoyield, energy resolution and noise characteristics. Table 1 summarizes the results, as well as the results for the Ortec572 amplifier, which are shown for comparison. The linearity of both outputs was checked with an Ortec Research Pulser. The low-gain output was linear within the measurement accuracy of a 12-bit ADC. The high-gain output showed a light nonlinearity in an area of less than 5% of its full range, which corresponds to 1 MeV in the energy scale. The nonlinearity is caused by an inadequate response (bypass feedback through stray elements) of the BLR-circuit, and does not evoke a noticeable error in the energy measurement. A random pulse generator was applied to reproduce the real waveform input signals for the main amplifier. The spectrum distortion of the low-gain output pulses was monitored as a function of the
Table 1 Shaping amplifier performances Amplifier
Ortec572. ssh = 0.5 ps Ortec572, T,~= 1ps Ortec.571. r,,, = I! I*S High gain output Low gain output
ENC Photoyield FWHM (p.e.) (p.e./MeV) at 1.27 MeV
770 680 650 670 700
8600 10200 12300
11000 10~0
Pulse width Ar,, I
(X)
(W)
12.3 14.7 11.3 14.1
6 8 13 s 7
counting rate. It has been found that the effect of pile-up was negligible for the peak resolution ( - 0.8% fwhm) at a random rate up to 50 kHz. Pile-up leads to losses of useful events. Under losses we consider the signals, whose shift from the generator pulse peak due to piIe-up was more than 5%. This limit was chosen rather arbitrary to evaluate the BLR performance. The fraction of lost pulses versus the counting rate is presented in Fig. 8. It is worth noting that the losses exceeded 40% even at 1OkHz rate in the tested amplifier without a RLR. 5.2. Constant-fraction
discriminator
An artificial pulse with a rise time of 430ns was formed to measure the time walk over the dynamic
Yu.G. Kudenko et al./Nucl.
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Ins@. and Meth. in Phw. Res. A 4/I
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Random frequency,
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Table 2 Time resolution
obtained
Energy
window
(MeVt
Integral 559 9913 15-19 2428 32 36 4&44 49953
(5100)
in the cosmic ray test FWHM
(ns)
12.6 59 40 ‘5.3 16.7
13.x Il.0 9.6
kHz
Fig. 8. Relative pile-up losses of a random pulse generator vs. counting rate. The losses are counted if the pulse height is shifted more than 5% of the peak value.
‘-1
445
(1998) 437-448
VOLTAGE,
V
Fig. 9. Output pulse time walk vs. input voltage in the CFD. The input pulse reproduced the real waveform of the CsI signal after TFA.
range of 1: 70. The maximum voltage was limited by the input range of the CFD. Since amplitudes below 100mV drastically deteriorate the time resolution, 90mV was fixed to be the low limit of the tested region. The time walk versus the input voltage is shown in Fig. 9. The walk is confined within 6 ns, contributing a relatively small part in the total time spread, as will be seen from a beam test. The raise of the right part of the curve is caused by waveform nonlinearity. The comparator dependence on the input slew rate is considered to be responsible for the gradual slope of the remaining part. Cosmic rays were employed to measure the time resolution in the appropriate energy range. A plas-
tic scintillator (100 x 200mm) provided the start pulse, while the stop one was generated by the CsI(T1) crystal. A time resolution of 12.6 ns (fwhm) has been achieved in the integral range from 4 MeV up to an energy extended over 100MeV with the electronic set: CP-preamplifier + TFA + CFD. The differential time resolution was measured within the energy windows, and the results are listed in Table 2. Shifting of the peak in the time spectrum was observed at small energies below 1S MeV. It is supposed to be caused not by the time walk, but by the difference in the ground potentials between the CFD and the TFA. A few millivolts, usually not noticed, resulted in the relative large time shift ( - 10 ns) due to the long rise time of discriminated signals. That kind of walk is suppressed by the alignment of the ground potentials.
6. Performance
in the beam
The CsI electronics system went into operation in October, 1995, when E246 first took data. During the subsequent 2 year period of operation, none of the 768 photodiodes failed. One preamplifier had become dead soon after detector operation, presumably due to bad assembly, thus confirming the fact that it is the most unreliable part of the readout electronics. The electronic noise was determined from the widths of the pedestals, and can be separated into coherent (correlated) and incoherent contributions. The incoherent contributions sum in quadrature, whereas the coherent ones combine linearly. The
446
Yu.G. Kudenko et al. jNuc1. Instr. and Meth. in Phys. Res. A 411 (1998) 437-448
3000
3500
4000
ADC channel Fig. 10. Typical pulse height spectrum of the Xe-lamp for a single. CsI crystal under a high intensity beam. The trigger was started by the Xe-puker.
noise contribution to the energy resolution (rr) obtained for a cluster of nine CsI crystals is 250 keV. The contribution to the noise from each crystal (incoherent) does not exceed 75 keV (a), and the coherent noise was found to be about 11 keV (a) per crystal. The correlated deposit increases the noise of a cluster by only 10%. A Xe-lamp was employed to test the baseline stability at high counting rates. The Xe-lamp peak position was monitored at different beam intensities. A typical pulse height spectrum for one crystal at a counting rate of about 20 kHz is shown in Fig. 10. The energy spectrum is distorted from the true Gaussian shape by piling up of the Xe-pulse with randomly occurring scintillating pulses from the beam. The peak position and resolution were preserved at a maximum measured counting rate of 32 kHz, where the counts were registered if the energy deposit in a crystal exceeded 10 MeV. We
calculated the losses by counting the Xe-lamp events whose energy shift from the peak was more than 8 MeV. Comparing this number with the total number of triggers generated by the Xe-pulser, we obtained the fraction of real events lost due to pile-up (Fig. it). It should be noted that a part of pile-up events will be saved in off-line analysis by fitting the signal waveform digitized by the TD. Since the average counting rate per crystal in our experiment is under 10 kHz, the pile-up does not create any significant loss of data during the final analysis. We consider a good time resolution as an important factor to provide reliable identification of neutral particles and to suppress the background. The time distribution for all crystals and the corresponding photon energy distribution from the rr” decay for the KP3 events recorded during beam runs are shown in Fig. 12(a) and (b), respectively.
Yu.G. Kudenko et al. /Nucl. Instr. and Meth. in Phvs. Res. .4 4/l
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(1998) 437-448
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I
I I
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s
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21)
25
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Counting rate,
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Fig. 11. Relative piie-up losses of the Xe-lamp vs. the counting rate in a crystal. The Xe-lamp rate was fixed at 10 Hz. The losses are counted if the Xe-pulse height shift from the peak value is more than 8 MeV.
Shown in Fig. 12(a) is the time spectrum for the crystals which are the centers of the gamma clusters. Only events with energy deposited in a central crystal above a threshold of 10MeV are shown. The time peak is well fitted by a Gaussian with CI= 4.8 ns. Some asymmetry of the peak shape is explained by the time shift observed at small pulse amplitudes (low energies). As seen in Fig. 13, a time resolution of 3.5 ns at 1OOMeV deteriorates to 16.8ns in the low energy range of 10-20MeV. The cosmic ray test described in the previous chapter showed a better resolution at a similar energy deposited in the crystal. It stems from the high quality (large light yield) of the selected crystal under the cosmic test, while the data obtained in the beam combines all 768 CsI modules of the calorimeter. An incorrect peak alignment for different crystals aiso widens the total time distribution. Exploiting the obtained timing performance, the Csl signals from all CFD outputs were inserted in the trigger logic as the OR strobe. A 1501~swidth of a discriminator pulse guarantees the acceptance of all valuable events at a level of about loo%, while reducing the trigger rate by a factor of 2,
7. Conclusions The readout electronics for the photon detector has been developed specifically for the needs of the
1500
: 250
750
500 150
c (b)
,_h!TRGY, MeV
Fig. It. (a) Times~ctrum of all Csl crystals with aligned peak positions. One TDC channel = 0,711s. (b) DisIribu~~on of the energy deposited in a crystal (center of cluster) during beam runs.
experiment E246. The electronic system consists of the low-noise preamplifier, a main amplifier, a CFD, a time digitizer, a peak-sensitive ADC and a commercial TDC. The preamplifier in combination with a large-area PIN-photodiode provides excellent noise characteristics and also features a compact size. A typical noise level of 640 (840) electrons (rms) at a shaping time of 2l.t~ was
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Acknowledgements s 3
16.0 -
The authors would like to thank K. Mori of Clear Pulse Co. for valuable technical suggestions, M. Aoki for helpful discussions, 0. Sasaki for cooperation in the CFD production. We would like also to thank all members of the E246 collaboration who participated in the beam runs. This work was done during a stay of one of the authors (O.M.) at KEK as a foreign researcher of the Japan Ministry of Education, Culture, Science and Sport. This work was supported in part by the Russian Foundation for Basic Research Grant No. 96-02-16081.
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achieved with S3204-03 (S3584-05) photodiodes at room temperature. The 16ch. main amplifier combines the shaping amplifier with two outputs to cover the large dynamic range and the timing filter amplifier. The performances of the shaping amplifier are similar to that of an Ortec 572 amplifier with a shaping time of 1 us. Intrinsic jumpers give an option to halve the shaping time. A random rate test was performed in order to verify the new design of a baseline restorer. No baseline shift was observed under a counting rate of up to 50 kHz. A CFD unit adapted to input pulses with a long rise time has been designed. A time resolution of 11.4ns (fwhm) has been obtained in beam runs for a photon energy range from 10 to 200 MeV.
References [l] J. Imazato et al., KEK Report 91-S. November 1991. [2] D.V. Dementyev et al., Nucl. Instc. and Meth. A 379 (1996) 499. [3] Yu.G. Kudenko. J. Imazato, KEK Report 92-15. October 1992. [4] E. Gramsch et al., Nucl. In&r. and Meth. A 311 (1992) 529. [S] D.R. Marlow et al., Preprint, Princeton/HEP/95-10. [6] Clear Pulse Co., Ohta-ku, Tokyo, 143 Japan. [7] T. Taniguchi et al., IEEE Trans. Nucl. Sci. NS-36 (1)(1989) 657. [8] Clear Pulse Co., Technical manual to 4040 Filter Amplifier. [P] S. Ohkawa, K. Husimi, IEEE Trans. Nucl. Sei. NS-33 (1) (1986) 415. [lo] B.T. Turko, R.C. Smith, IEEE Trans. Nucl. Sci. NS-39 (5) (1992) 1311. [ll] B. Gottschalk, Nucl. Instr. and Meth. 190 (1981) 67. [12] EG%G ORTEC Detectors & Instruments for Nuclear Spectroscopy Catalog (1991/1992) 3-138. [I33 TKO specification. KEK Report 85-10, 1985. [14] M.P. Grigoryev et al., Instr. Exp. Tech. 39 (2) (1996) 164.