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Design of a compact silicon-based integrated passive band-pass filter with two tunable finite transmission zeros
Microelectronics Journal 49 (2016) 43–48
Contents lists available at ScienceDirect
Microelectronics Journal journal homepage: www.elsevier.com/locate/mejo
Design of a compact silicon-based integrated passive band-pass filter with two tunable finite transmission zeros Jie Pan a,b,n, Huijuan Wang a,b, Gengxin Tian a,b, Liqiang Cao a,b, Daquan Yu a a b
Institute of Microelectronics of Chinese Academy of Sciences, Beijing, China National Center for Advanced Packaging (NCAP China), Wuxi 214315, China
art ic l e i nf o
a b s t r a c t
Article history: Received 8 October 2015 Received in revised form 26 December 2015 Accepted 29 December 2015 Available online 19 January 2016
The thin film integrated passive device (IPD) has caused intensively attention due to its high integrated level and application in System in Package (SiP). Meanwhile, the IPD filter has shown great application value in modern wireless communication system. In this paper, the design method of a compact siliconbased lumped IPD band-pass filter with two tunable finite transmission zeros is shown. The passband of the designed filter is 2.4–2.5 GHz. Specific schematic is used to introduce two independently tunable transmission zeros. The specific schematic contains mutual inductor structure, which is realized by making two planar spiral inductors to partially overlap with each other and is helpful to decrease the size of the filter. The mutual inductance is adjustable by changing the overlap degree. To design and optimize the layout of the filter more efficiently, the schematic-layout mutual feedback method is presented. Finally, the measurement results verify the validity of the whole design process and method. The size of the designed second-order band-pass filter is 0.85 mm 0.85 mm 0.35 mm. To the best of our knowledge, the band-pass-filter developed in this paper has the smallest size while achieving equivalent electrical performance. & 2016 Elsevier Ltd. All rights reserved.
Keywords: IPD band-pass filter Tunable finite transmission zeros Mutual inductor structure Schematic-layout mutual feedback method
1. Introduction Miniaturization, cost and performance have been the driving factors of the development of modern communication system. In typical wireless communication system, there are plenty of passive components, which consume about 60–70% of the whole area of the system [1]. Silicon-based IPD technology offers a feasible solution for further decreasing the size and cost of the wireless communication system. Conventional discrete passive components used for wireless products are typically fabricated by ceramic technology, which has good electrical and thermal characteristics. Compared with ceramic technology, the silicon-based IPD technology shows the advantage of more excellent process parameter control, finer width and spacing, smaller form-factor and lower cost [2,3]. Also, the electrical performance of silicon-based IPD technology is good when using high resistivity silicon substrate. In addition, advanced process technology is promising in small form-factor filter design, such as TSV [4,5]. The silicon-based IPD process technology used in this paper is shown in Fig. 1 [6]. The inductor and capacitor are realized respectively. There are four metal layers (M1, M2, M3 and MIM) n
Corresponding author. E-mail addresses: [email protected] (J. Pan), [email protected] (D. Yu).
http://dx.doi.org/10.1016/j.mejo.2015.12.012 0026-2692/& 2016 Elsevier Ltd. All rights reserved.
and four dielectric layers. Layers M1 and MIM are used to form metal–insulator–metal capacitors. Inductors are implemented in layers M1, M2 and M3. Layers M1 and M2 are connected by via. Layer M2 is made of thick copper so as to increase the Q factor of inductors. In modern communication system, the filter is an important passive device which has the function of frequency-selecting. It allows desired signal to pass and suppresses the useless signal. The filter can be realized as lumped form by silicon-based IPD technology. Generally, most silicon-based IPD band-pass filter topologies are of the coupled-resonator type, as the values of LCs in the coupled-resonator topology are accessible with silicon-based IPD technology [7]. Based on the coupled-resonator topology, it’s easy to introduce transmission zeros into the filter. The transmission zeros can be introduced by changing the structure of resonator or using cross coupling [8,9]. By introducing proper transmission zeros, the rectangle coefficient of the filter is improved and the rejection of sensitive frequency band is increased. Some papers have presented the schematics of second-order filters with two finite transmission zeros [10,11]. However, in these papers locations of the two transmission zeros are dependent and cannot adjusted independently. In this paper, we have designed and fabricated a compact silicon-based IPD band-pass filter. Two independently tunable finite transmission zeros are introduced by using specific coupled-
44
J. Pan et al. / Microelectronics Journal 49 (2016) 43–48
copper aluminum
PI
Inductor
0 To obtain the expression of y21 , the nodal circuit and voltage 0 analysis is performed. The y21 can be expressed as: s U L3 C 4 e þ 1s U L3 C 4 f þ L1 Lþ1 L3 e þ L11C 3 h þ s13 U L1 Lþ1 L3 f þ L1gL3 y21 ð3Þ ' ¼ de ahÞ þ s12 U ðce þdf ag bhÞ þ s14 U ðcf bgÞ
Capacitor
M3
M3
M2
M2 Via
MIM
M1
Si 3 N4
Via M1
where
Oxide Substrate Fig. 1. The structure of silicon-based IPD process technology.
C5 C1
C2
L1
Port 2
Port 1 C3
L2
L3
a¼
1 1 1 þ þ C1 C3 C6
ð4Þ
b¼
1 1 1 þ þ C 1 C 3 L2 C 3 C 6 L2 C 1 C 3 L1
ð5Þ
c¼
1 L3 C 6 C 1 L1
ð6Þ
d¼
L3 C 4 C6
ð7Þ
e¼
1 C6
ð8Þ
f¼
1 1 C 3 C 6 L2 C 2 C 3 L1
ð9Þ
g¼
1 1 L3 þ þ C 2 C 6 C 2 L1
ð10Þ
h¼
L3 C 4 L3 C 4 þ L3 þ C2 C6
ð11Þ
C4
C6 Fig. 2. The proposed schematic of the two-pole second-order filter.
resonator topology. Mutual inductor structure is used to decrease the size of the filter.
C 1 ; C 2 ; C 3 ; C 4 ; C 5 ; C 6 ; L1 ; L2 and L3 represent the values of relevant capacitors and inductors. Finally, substituting (3) into (2), a fourth-order polynomial equation of s can be obtained as follows: l Us4 þ m U s2 þ n ¼ 0
2. Theory
l ¼ L3 C 4 e ðde ahÞC 5
The proposed schematic of the second-order filter is shown in Fig. 2. The topology of the filter belongs to the coupled-resonator type. The schematic consists of a second-order coupled resonator band-pass filter, an upper parallel capacitor C5 and a shunt ground capacitor C6. It will be demonstrated later that two finite transmission zeros are produced by introducing C5 and C6 into the schematic. One transmission zero locates in the lower stopband and the other locates in the upper stopband. The location of the transmission zeros can be changed independently by tuning the value of C5 and C6. Meanwhile, C5 and C6 have little impact on the passband characteristic of the filter. To prove the existence of the finite transmission zeros, the schematic is divided into two parts. One part is the upper parallel capacitor C5 and the other is the remanent part. The two parts are connected parallelly. Thus, the overall admittance matrix of the schematic can be express as: " Y¼
0 sC 5 þ y11
0 sC 5 þ y12
0 sC 5 þ y21
0 sC 5 þy22
# ð1Þ
0 0 0 ,y12 ,y21 where s ¼jω, C5 is the capacitance of capacitor C5 and y11 0 and y22 are the elements of the admittance matrix of the remanent 0 0 part. As the filter is passive, y12 equals to y21 . Based on the admittance matrix, the locations of the finite transmission zeros are determined by the following equation: 0 sC 5 þ y21 ¼0
ð12Þ
where
ð2Þ
m ¼ L3 C 4 f þ n¼
L1 þ L3 1 eþ h ðce þ df ag bhÞC 5 L1 C 3 L1
L1 þL3 g fþ ðcf bgÞC 5 L1 C 3 L1
ð13Þ ð14Þ ð15Þ
The two positive roots are the frequencies of the finite transmission zeros. As there are two tunable parameters (C5 and C6) in expression (12), the two positive roots are not dependent directly and can be adjusted freely. To further intuitively understand the introduction of transmission zeros, numerical analysis is performed. The values of the components in second-order coupled resonator band-pass filter part are chosen as: C1 ¼C2 ¼1.71 pF, C3 ¼C4 ¼1.45 pF, L1 ¼4.64 nH, L2 ¼L3 ¼2.76 nH, to ensure that the passband is 2.4–2.5 GHz. By introducing C5 and C6, two finite transmission zeros are produced, which is shown in Fig. 3(a). It also can be seen that the passband characteristics has changed little. By tuning the values of C5 and C6, the location of the transmission zeros can be adjusted freely, which is shown in Fig. 3(b). The intersection points in Fig. 3 (b) indicate the locations of the transmission zeros. Generally, the inductance of L1 is large and will greatly decrease the integration level of the filter. Here, the Y–Δ transformation can be applied to transform L1 into the mutual inductance between L2 and L3. Fig. 4 shows the schematics of the transformation. The transformation formulas are: LL2 ¼
L2 ðL3 þ L1 Þ L1 þ L2 þ L3
ð16Þ
J. Pan et al. / Microelectronics Journal 49 (2016) 43–48
45
Fig. 5. The top view of the mutual inductor structure.
V1 I_AC Iac=polar(1,0) A Freq=freq
Fig. 3. (a) The transmission and reflection responses of filters with and without C5 and C6. (b) The relationship of the locations of transmission zeros and the values of C5 and C6.
1
2
V2
Ref
S2P
Fig. 6. The inductance extraction circuit.
Fig. 4. The schematics of Y–Δ transformation.
LL3 ¼ M¼
L3 ðL1 þ L2 Þ L1 þ L2 þ L3
L2 L3 L1 þ L2 þ L3
ð17Þ ð18Þ
Obviously, the size of the mutual inductor structure is much smaller than two closely placed inductors. Thus the integrated level of the filter is greatly enhanced. Based on field and circuit simulation, the self and mutual inductance extraction method is proposed. Fig. 6 shows the inductance extraction circuit. The S2P module contains the information of scattering matrix, which is obtained by field simulation. I_AC is a sine AC (alternating current) current source. The amplitude of the current source is 1 A and its initial phase is 0. For the mutual inductor structure, V 1 ¼ L11
dI 1 dI 2 þL12 dt dt
ð19Þ
V 2 ¼ L21
dI 1 dI 2 þL22 dt dt
ð20Þ
3. Design of the mutual inductor structure The aforementioned schematic contains the mutual inductor structure. Generally, two inductors can constitute the mutual inductor structure by placing them close to each other. However, for the planar spiral inductor, which is most used in silicon-based IPD process, the coupling coefficient is too small by just placing two inductors close to each other. Inspired by the transformer, a novel mutual inductor structure is presented. The mutual inductor structure consists of two inductors, which are placed staggerly and partially overlapping with each other. The routing conflict can be avoided by using different non-adjacent metal layers in the conflict area. Fig. 5 shows the top view of the mutual inductor structure. The variable d indicated in Fig. 5 is the distance of two specific adjacent edges. It can be used to evaluate the overlap degree of the two inductors. By changing the variable d, the coupling coefficient of the mutual inductor structure can be adjusted.
where V1 and V2 are the voltages of inductor 1 and inductor 2, respectively, I1 and I2 are the currents of inductor 1 and inductor 2, L11 and L22 denote the self inductances, and L12 and L21 denote the mutual inductances. L12 equals to L21. For the specific inductance extraction circuit, I2 ¼ 0
ð21Þ
I 1 ¼ I 0 U ejωt
ð22Þ
By solving (19)–(22), L11 and L21 can be expressed as: L11 ¼
magðV1Þ 2π f req
ð23Þ
L21 ¼
magðV2Þ 2π f req
ð24Þ
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J. Pan et al. / Microelectronics Journal 49 (2016) 43–48
Self Inductance (nH)
3.0
d=10um d=30um d=50um d=70um d=100um
2.8 2.6 2.4 2.2 2.0
0.0 0.5 1.0 1.5 2.0 2.5 3.0 3.5 4.0 4.5 5.0 Frequency (GHz)
Mutual Inductance (nH)
1.6
d=10um d=30um d=50um d=70um d=100um
1.4 1.2
Fig. 8. The fitting result of the variable d and mutual inductance.
4. Layout design—the schematic-layout mutual feedback method
1.17nH 0.95nH
1.0
0.79nH 0.65nH
0.8 0.6
0.44nH
0.4 0.20.0 0.5 1.0 1.5 2.0 2.5 3.0 3.5 4.0 4.5 5.0 Frequency (GHz)
Fig. 7. (a) The simulated relationship of the self Inductance and variable d. (b) The simulated relationship of the mutual Inductance and variable d. Simulated by ADS (Agilent Advanced Design System).
Table 1 The concrete values of self and mutual inductances at 2.5 GHz for different values of d. d (um) L11 (nH) L21 (nH) k
10 2.2 1.17 0.53
30 2.2 0.95 0.43
50 2.2 0.79 0.36
70 2.3 0.65 0.28
100 2.3 0.44 0.19
L22 can be obtained by moving the current source from port1 to port2. If the two inductors have the same shape, then L11 equals to L22. Based on the inductance extraction method, several mutual inductor structures are analyzed and the coupling coefficients are calculated. For the mutual inductor structure, the coupling coefficient k is defined as: L21 k ¼ pffiffiffiffiffiffiffiffiffiffiffiffiffi L11 L22
ð25Þ
Fig. 7 shows the self and mutual inductances of the mutual inductor structures with different values of d. As the single inductor structures are almost the same, the values of self inductances are all about 2.2 nH at 2.5 GHz. With decrease of the variable d, the mutual inductance increases. The concrete values of self and mutual inductances at 2.5 GHz are listed in Table 1. For the specific mutual inductor structures, the relationship of d and L21 can be obtained by numerical fitting, so as to offer design reference. Fig. 8 shows the fitting result. It can be seen that d and L21 are linearly dependent to a first approximation. The linear fitting expression is: L21 ðnHÞ ¼ 0:008dðumÞ þ 1:22
ð26Þ
After design of the schematic and the physical structures of inductor, capacitor and mutual inductor, the final step of filter design is to implement the whole layout. However, the unavoidable parasitic effect usually makes the layout design a complex and formidable task. In this part, we present a convenient and accurate layout design method named the schematic-layout mutual feedback method. The key point of the method is to consider the component value shift and parasitic effect into schematic simulation. A concrete layout design example is given below. In this example, the layout is simulated with ADS (Advance Design System). The passband of the designed filter is 2.4–2.5 GHz. The locations of the two finite transmission zeros are chosen to be 1.9 and 4.8 GHz, which are major interference frequency bands. The first step of the schematic-layout mutual feedback method is to design the initial layout called layout V1. The layout V1 is designed according to the schematic V1. Fig. 9(a) shows the schematic V1 and the values of the components are listed. Variable R is the parasitic resistance of the inductor. Fig. 9(b) shows the simulation structure of layout V1. Fig. 9(c) compares the simulation results of the schematic V1 and layout V1. For the layout V1, the location of the transmission zero in upper stopband shifts toward low frequency compared to schematic V1. This is caused by the value shift of the inductor, capacitor and mutual inductor structure, and the parasitic effect. As the layout simulation considers the loss effect, the insertion loss of layout V1 increases compared to schematic V1. The second step is to adjust the schematic V1 so as to obtain the schematic V2. The adjusting criterion is to match the response of the schematic V2 with layout V1. If their responses are matched, it means that the schematic V2 considers layout parasitics and component value shift correctly. The adjusting process contains tuning the component values and adding extra parasitic components if necessary. From schematic V1 to schematic V2, the component values are changed as: C5: 0.08-0.12 pF, C6: 2.9-2.7 pF, LL2 and LL3: 2.22-2.44 nH, M: 0.95-0.91 nH, R: 0-1.83 Ohm. No extra parasitic components are added. Fig. 10 compares the filter responses of schematic V2 and layout V1. Their responses match quite well. The final step is to optimize the schematic V2 so as to obtain the schematic V3, which conforms to the filter design index. Then feed back the schematic variation to the layout V1 and get layout V2. As schematic V2 and layout V1 are equivalent, their component variations should be the same to obtain the desired filter response. From schematic V2 to schematic V3, the component values are changed as: C5: 0.12-0.04 pF. Thus, the layout optimization from layout V1 to layout V2 should be: C5: 0.08-0 pF. The variation of C5 is 0.08 pF for both the schematic and layout
J. Pan et al. / Microelectronics Journal 49 (2016) 43–48
Port 1
M
C1 C3
C2 Port 2
LL2 R
LL3 R
C4
C6
Variables C1=1.61pF C2=1.61pF C3=1.15pF C4=1.15pF C5=0.08pF C6=2.90pF LL2=2.22nH LL3=2.22nH M=0.95nH R=0Ω
0 -5 S Parameter (dB)
C5
47
S11
-10 -15
S21
-20 -25 -30
Schematic V3 Layout V2
-35 0.0 0.5 1.0 1.5 2.0 2.5 3.0 3.5 4.0 4.5 5.0 Frequency (GHz) Fig. 11. The comparisons of transmission and reflection responses of the schematic V3 and layout V2.
0.85um
0
S Parameter (dB)
S11 -10 -20
S21
-30 -40 -50 0.0
Schematic V1 Layout V1
0.5 1.0
1.5 2.0 2.5 3.0 3.5 Frequency (GHz)
0.85um 4.0 4.5 5.0
Fig. 9. (a) The schematic V1 and the values of the components. (b) The simulation structure of layout V1. (c) The comparisons of transmission and reflection responses of the schematic V1 and layout V1.
S Parameter (dB)
S11
-10 -15
S21
-20 -25
The concrete layout design example has presented the schematic-layout mutual feedback design method. Obviously, based on this method, the layout can be designed simply and accurately.
5. Experimental results
0 -5
Fig. 12. The microscope photo of the fabricated filter.
Schematic V2 Layout V1
-30 0.0 0.5 1.0 1.5 2.0 2.5 3.0 3.5 Frequency (GHz)
4.0 4.5 5.0
Fig. 10. The comparisons of transmission and reflection responses of the schematic V2 and layout V1.
adjustment. Fig. 11 compares the filter responses of the schematic V3 and layout V2. The response of layout V2 is consistent with schematic V3. Also, the locations of the transmission zeros of layout V2 are in the desired positions, which are about 1.9 GHz and 4.8 GHz, respectively.
A silicon-based IPD band-pass filter has been designed according to the above design method. It is fabricated with the 0.35um IPD process in SMIC (Semiconductor Manufacturing International Corporation). The passband of the fabricated filter is 2.4–2.5 GHz. The locations of the transmission zeros are chosen to be 1.75 GHz and 4.8 GHz for project need. Fig. 12 shows the microscope photo of the filter. Its size is about 0.85 mm 0.85 mm 0.35 mm. Fig. 13 presents the EM simulation and test results of the fabricated filter. The EM simulation is conducted by ADS and HFSS (High-Frequency Structure Simulator), respectively. Obviously, the simulation result of HFSS matches quite well with the test result. As the filter is fabricated by front-end process, the process deviation is little. For the simulation result of ADS, the location of the transmission zero in lower stopband shifts toward high frequency compared to test result. The frequency offset is about 70 MHz. This is probably caused by the deviation of ADS momentum simulation. Analyzing the test data, the passband insertion loss is better than 2.3 dB. The passband insertion loss can be optimized by using thicker conductors and replacing aluminum layer with copper
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J. Pan et al. / Microelectronics Journal 49 (2016) 43–48
in this paper has the smallest size while achieving equivalent electrical performance.
0
S Parameter (dB)
-5
S11
-10
Acknowledgment
-15 -20 -25 -30
S21 Test Simulated by ADS Simulated by HFSS
-35 0.0 0.5 1.0 1.5 2.0 2.5 3.0 3.5 4.0 4.5 5.0 Frequency (GHz) Fig. 13. The comparisons of EM simulation (ADS and HFSS) and test results of the fabricated filter.
layer. The fluctuation of passband is little. For the transmission zeros, their locations are accurately controlled. The suppressions of 1.75 GHz and 4.8 GHz are about 29 dB and 27 dB, respectively. The suppression degree is good. The size of the filter is about 0.85 mm 0.85 mm 0.35 mm. In Ref. [8], size of the bandpass filter with same center frequency is 2.50 mm 2.00 mm 0.80 mm. To the best of our knowledge, the fabricated filter has the smallest size while achieving equivalent electrical performance. The experimental results verify the advantages of introducing freely tunable transmission zeros and using mutual inductor structure. Also, it proves the correctness of the whole design flow and method.
6. Conclusion In this paper, a compact silicon-based IPD band-pass-filter with two independently tunable finite transmission zeros are designed and fabricated. Based on related circuit analysis, specific schematic is introduced to ensure two independently tunable transmission zeros. The mutual inductor structure is presented to realize the filter, which greatly decreases the filter size. The design and analysis method of the mutual inductor structure is shown. For the layout design, the schematic-layout mutual feedback method is presented so as to design the layout simply and accurately. Finally, the experiment results verify the correctness of the whole design flow and method. To the best of our knowledge, the designed filter
The authors appreciate the support of National Science and Technology Major Project under project No. 2014ZX03001015-003 and Daquan Yu would like to thank the support of the One Hundred Person Project of the Chinese Academy of Sciences. This paper’s proposed mutual inductor structure is patent pending.
References [1] L. Kai, R. Frye, R. Emigh, Band-pass-filter with balun function from IPD technology [C], in: Proceedings of the Electronic Components and Technology Conference, 2008 ECTC 2008 58th, 2008, pp. 718–723. [2] A.C. Kundu, M. Megahed, D. Schmidt, Comparison and analysis of integrated passive device technologies for wireless radio frequency module [C], in: Proceedings of the Electronic Components and Technology Conference, 2008 ECTC 2008 58th, 2008, pp. 683–687. [3] L. Yong-Taek, L. Kai, R. Frye, et al., Ultra-wide-band (UWB) band-pass-filter using integrated passive device (IPD) technology for wireless applications [C], in: Proceedings of the Electronic Components and Technology Conference, 2009 ECTC 2009 59th, 2009, pp. 1994–1999. [4] Q. Lu, Z. Zhu, Y. Yang, et al., Accurate formulas for the capacitance of taperedthrough silicon vias in 3-D ICs [J], Microw. Wirel. Compon. Lett. IEEE 24 (24) (2014) 294–296. [5] Q. Lu, Z. Zhu, Y. Yang, et al., Electrical modeling and characterization of shield differential through-silicon vias [J], IEEE Trans. Electron Devices 62 (5) (2015) 1544–1552. [6] L. Kai, R.C. Frye, Small form-factor integrated passive devices for SiP applications [C], in: Proceedings of the Microwave Symposium, 2007 IEEE/MTT-S International, 2007, pp. 2117–2120. [7] R.C. Frye, L. Kai, G. Badakere, et al., A hybrid coupled-resonator bandpass filter topology implemented on lossy semiconductor substrates [C], in: Proceedings of the Microwave Symposium, 2007 IEEE/MTT-S International, 2007, pp. 1757– 1760. [8] J.B. Thomas, Cross-coupling in coaxial cavity filters-a tutorial overview [J], Microw. Theory Tech., IEEE Trans. 51 (4) (2003) 1368–1376. [9] L. Wing-Yan, K.K.M. Cheng, W. Ke-Li, Multilayer LTCC bandpass filter design with enhanced stopband characteristics [J], Microw. Wirel. Compon. Lett., IEEE 12 (7) (2002) 240–242. [10] C. Chun-Fu, C. Shyh-Jong, Bandpass filter of serial configuration with two finite transmission zeros using LTCC technology [J], Microw. Theory Tech., IEEE Trans. 53 (7) (2005) 2383–2388. [11] Y. Lap Kun, W. Ke-Li, A compact second-order LTCC bandpass filter with two finite transmission zeros [J], Microw. Theory Tech., IEEE Trans. 51 (2) (2003) 337–341.
Contents lists available at ScienceDirect
Microelectronics Journal journal homepage: www.elsevier.com/locate/mejo
Design of a compact silicon-based integrated passive band-pass filter with two tunable finite transmission zeros Jie Pan a,b,n, Huijuan Wang a,b, Gengxin Tian a,b, Liqiang Cao a,b, Daquan Yu a a b
Institute of Microelectronics of Chinese Academy of Sciences, Beijing, China National Center for Advanced Packaging (NCAP China), Wuxi 214315, China
art ic l e i nf o
a b s t r a c t
Article history: Received 8 October 2015 Received in revised form 26 December 2015 Accepted 29 December 2015 Available online 19 January 2016
The thin film integrated passive device (IPD) has caused intensively attention due to its high integrated level and application in System in Package (SiP). Meanwhile, the IPD filter has shown great application value in modern wireless communication system. In this paper, the design method of a compact siliconbased lumped IPD band-pass filter with two tunable finite transmission zeros is shown. The passband of the designed filter is 2.4–2.5 GHz. Specific schematic is used to introduce two independently tunable transmission zeros. The specific schematic contains mutual inductor structure, which is realized by making two planar spiral inductors to partially overlap with each other and is helpful to decrease the size of the filter. The mutual inductance is adjustable by changing the overlap degree. To design and optimize the layout of the filter more efficiently, the schematic-layout mutual feedback method is presented. Finally, the measurement results verify the validity of the whole design process and method. The size of the designed second-order band-pass filter is 0.85 mm 0.85 mm 0.35 mm. To the best of our knowledge, the band-pass-filter developed in this paper has the smallest size while achieving equivalent electrical performance. & 2016 Elsevier Ltd. All rights reserved.
Keywords: IPD band-pass filter Tunable finite transmission zeros Mutual inductor structure Schematic-layout mutual feedback method
1. Introduction Miniaturization, cost and performance have been the driving factors of the development of modern communication system. In typical wireless communication system, there are plenty of passive components, which consume about 60–70% of the whole area of the system [1]. Silicon-based IPD technology offers a feasible solution for further decreasing the size and cost of the wireless communication system. Conventional discrete passive components used for wireless products are typically fabricated by ceramic technology, which has good electrical and thermal characteristics. Compared with ceramic technology, the silicon-based IPD technology shows the advantage of more excellent process parameter control, finer width and spacing, smaller form-factor and lower cost [2,3]. Also, the electrical performance of silicon-based IPD technology is good when using high resistivity silicon substrate. In addition, advanced process technology is promising in small form-factor filter design, such as TSV [4,5]. The silicon-based IPD process technology used in this paper is shown in Fig. 1 [6]. The inductor and capacitor are realized respectively. There are four metal layers (M1, M2, M3 and MIM) n
Corresponding author. E-mail addresses: [email protected] (J. Pan), [email protected] (D. Yu).
http://dx.doi.org/10.1016/j.mejo.2015.12.012 0026-2692/& 2016 Elsevier Ltd. All rights reserved.
and four dielectric layers. Layers M1 and MIM are used to form metal–insulator–metal capacitors. Inductors are implemented in layers M1, M2 and M3. Layers M1 and M2 are connected by via. Layer M2 is made of thick copper so as to increase the Q factor of inductors. In modern communication system, the filter is an important passive device which has the function of frequency-selecting. It allows desired signal to pass and suppresses the useless signal. The filter can be realized as lumped form by silicon-based IPD technology. Generally, most silicon-based IPD band-pass filter topologies are of the coupled-resonator type, as the values of LCs in the coupled-resonator topology are accessible with silicon-based IPD technology [7]. Based on the coupled-resonator topology, it’s easy to introduce transmission zeros into the filter. The transmission zeros can be introduced by changing the structure of resonator or using cross coupling [8,9]. By introducing proper transmission zeros, the rectangle coefficient of the filter is improved and the rejection of sensitive frequency band is increased. Some papers have presented the schematics of second-order filters with two finite transmission zeros [10,11]. However, in these papers locations of the two transmission zeros are dependent and cannot adjusted independently. In this paper, we have designed and fabricated a compact silicon-based IPD band-pass filter. Two independently tunable finite transmission zeros are introduced by using specific coupled-
44
J. Pan et al. / Microelectronics Journal 49 (2016) 43–48
copper aluminum
PI
Inductor
0 To obtain the expression of y21 , the nodal circuit and voltage 0 analysis is performed. The y21 can be expressed as: s U L3 C 4 e þ 1s U L3 C 4 f þ L1 Lþ1 L3 e þ L11C 3 h þ s13 U L1 Lþ1 L3 f þ L1gL3 y21 ð3Þ ' ¼ de ahÞ þ s12 U ðce þdf ag bhÞ þ s14 U ðcf bgÞ
Capacitor
M3
M3
M2
M2 Via
MIM
M1
Si 3 N4
Via M1
where
Oxide Substrate Fig. 1. The structure of silicon-based IPD process technology.
C5 C1
C2
L1
Port 2
Port 1 C3
L2
L3
a¼
1 1 1 þ þ C1 C3 C6
ð4Þ
b¼
1 1 1 þ þ C 1 C 3 L2 C 3 C 6 L2 C 1 C 3 L1
ð5Þ
c¼
1 L3 C 6 C 1 L1
ð6Þ
d¼
L3 C 4 C6
ð7Þ
e¼
1 C6
ð8Þ
f¼
1 1 C 3 C 6 L2 C 2 C 3 L1
ð9Þ
g¼
1 1 L3 þ þ C 2 C 6 C 2 L1
ð10Þ
h¼
L3 C 4 L3 C 4 þ L3 þ C2 C6
ð11Þ
C4
C6 Fig. 2. The proposed schematic of the two-pole second-order filter.
resonator topology. Mutual inductor structure is used to decrease the size of the filter.
C 1 ; C 2 ; C 3 ; C 4 ; C 5 ; C 6 ; L1 ; L2 and L3 represent the values of relevant capacitors and inductors. Finally, substituting (3) into (2), a fourth-order polynomial equation of s can be obtained as follows: l Us4 þ m U s2 þ n ¼ 0
2. Theory
l ¼ L3 C 4 e ðde ahÞC 5
The proposed schematic of the second-order filter is shown in Fig. 2. The topology of the filter belongs to the coupled-resonator type. The schematic consists of a second-order coupled resonator band-pass filter, an upper parallel capacitor C5 and a shunt ground capacitor C6. It will be demonstrated later that two finite transmission zeros are produced by introducing C5 and C6 into the schematic. One transmission zero locates in the lower stopband and the other locates in the upper stopband. The location of the transmission zeros can be changed independently by tuning the value of C5 and C6. Meanwhile, C5 and C6 have little impact on the passband characteristic of the filter. To prove the existence of the finite transmission zeros, the schematic is divided into two parts. One part is the upper parallel capacitor C5 and the other is the remanent part. The two parts are connected parallelly. Thus, the overall admittance matrix of the schematic can be express as: " Y¼
0 sC 5 þ y11
0 sC 5 þ y12
0 sC 5 þ y21
0 sC 5 þy22
# ð1Þ
0 0 0 ,y12 ,y21 where s ¼jω, C5 is the capacitance of capacitor C5 and y11 0 and y22 are the elements of the admittance matrix of the remanent 0 0 part. As the filter is passive, y12 equals to y21 . Based on the admittance matrix, the locations of the finite transmission zeros are determined by the following equation: 0 sC 5 þ y21 ¼0
ð12Þ
where
ð2Þ
m ¼ L3 C 4 f þ n¼
L1 þ L3 1 eþ h ðce þ df ag bhÞC 5 L1 C 3 L1
L1 þL3 g fþ ðcf bgÞC 5 L1 C 3 L1
ð13Þ ð14Þ ð15Þ
The two positive roots are the frequencies of the finite transmission zeros. As there are two tunable parameters (C5 and C6) in expression (12), the two positive roots are not dependent directly and can be adjusted freely. To further intuitively understand the introduction of transmission zeros, numerical analysis is performed. The values of the components in second-order coupled resonator band-pass filter part are chosen as: C1 ¼C2 ¼1.71 pF, C3 ¼C4 ¼1.45 pF, L1 ¼4.64 nH, L2 ¼L3 ¼2.76 nH, to ensure that the passband is 2.4–2.5 GHz. By introducing C5 and C6, two finite transmission zeros are produced, which is shown in Fig. 3(a). It also can be seen that the passband characteristics has changed little. By tuning the values of C5 and C6, the location of the transmission zeros can be adjusted freely, which is shown in Fig. 3(b). The intersection points in Fig. 3 (b) indicate the locations of the transmission zeros. Generally, the inductance of L1 is large and will greatly decrease the integration level of the filter. Here, the Y–Δ transformation can be applied to transform L1 into the mutual inductance between L2 and L3. Fig. 4 shows the schematics of the transformation. The transformation formulas are: LL2 ¼
L2 ðL3 þ L1 Þ L1 þ L2 þ L3
ð16Þ
J. Pan et al. / Microelectronics Journal 49 (2016) 43–48
45
Fig. 5. The top view of the mutual inductor structure.
V1 I_AC Iac=polar(1,0) A Freq=freq
Fig. 3. (a) The transmission and reflection responses of filters with and without C5 and C6. (b) The relationship of the locations of transmission zeros and the values of C5 and C6.
1
2
V2
Ref
S2P
Fig. 6. The inductance extraction circuit.
Fig. 4. The schematics of Y–Δ transformation.
LL3 ¼ M¼
L3 ðL1 þ L2 Þ L1 þ L2 þ L3
L2 L3 L1 þ L2 þ L3
ð17Þ ð18Þ
Obviously, the size of the mutual inductor structure is much smaller than two closely placed inductors. Thus the integrated level of the filter is greatly enhanced. Based on field and circuit simulation, the self and mutual inductance extraction method is proposed. Fig. 6 shows the inductance extraction circuit. The S2P module contains the information of scattering matrix, which is obtained by field simulation. I_AC is a sine AC (alternating current) current source. The amplitude of the current source is 1 A and its initial phase is 0. For the mutual inductor structure, V 1 ¼ L11
dI 1 dI 2 þL12 dt dt
ð19Þ
V 2 ¼ L21
dI 1 dI 2 þL22 dt dt
ð20Þ
3. Design of the mutual inductor structure The aforementioned schematic contains the mutual inductor structure. Generally, two inductors can constitute the mutual inductor structure by placing them close to each other. However, for the planar spiral inductor, which is most used in silicon-based IPD process, the coupling coefficient is too small by just placing two inductors close to each other. Inspired by the transformer, a novel mutual inductor structure is presented. The mutual inductor structure consists of two inductors, which are placed staggerly and partially overlapping with each other. The routing conflict can be avoided by using different non-adjacent metal layers in the conflict area. Fig. 5 shows the top view of the mutual inductor structure. The variable d indicated in Fig. 5 is the distance of two specific adjacent edges. It can be used to evaluate the overlap degree of the two inductors. By changing the variable d, the coupling coefficient of the mutual inductor structure can be adjusted.
where V1 and V2 are the voltages of inductor 1 and inductor 2, respectively, I1 and I2 are the currents of inductor 1 and inductor 2, L11 and L22 denote the self inductances, and L12 and L21 denote the mutual inductances. L12 equals to L21. For the specific inductance extraction circuit, I2 ¼ 0
ð21Þ
I 1 ¼ I 0 U ejωt
ð22Þ
By solving (19)–(22), L11 and L21 can be expressed as: L11 ¼
magðV1Þ 2π f req
ð23Þ
L21 ¼
magðV2Þ 2π f req
ð24Þ
46
J. Pan et al. / Microelectronics Journal 49 (2016) 43–48
Self Inductance (nH)
3.0
d=10um d=30um d=50um d=70um d=100um
2.8 2.6 2.4 2.2 2.0
0.0 0.5 1.0 1.5 2.0 2.5 3.0 3.5 4.0 4.5 5.0 Frequency (GHz)
Mutual Inductance (nH)
1.6
d=10um d=30um d=50um d=70um d=100um
1.4 1.2
Fig. 8. The fitting result of the variable d and mutual inductance.
4. Layout design—the schematic-layout mutual feedback method
1.17nH 0.95nH
1.0
0.79nH 0.65nH
0.8 0.6
0.44nH
0.4 0.20.0 0.5 1.0 1.5 2.0 2.5 3.0 3.5 4.0 4.5 5.0 Frequency (GHz)
Fig. 7. (a) The simulated relationship of the self Inductance and variable d. (b) The simulated relationship of the mutual Inductance and variable d. Simulated by ADS (Agilent Advanced Design System).
Table 1 The concrete values of self and mutual inductances at 2.5 GHz for different values of d. d (um) L11 (nH) L21 (nH) k
10 2.2 1.17 0.53
30 2.2 0.95 0.43
50 2.2 0.79 0.36
70 2.3 0.65 0.28
100 2.3 0.44 0.19
L22 can be obtained by moving the current source from port1 to port2. If the two inductors have the same shape, then L11 equals to L22. Based on the inductance extraction method, several mutual inductor structures are analyzed and the coupling coefficients are calculated. For the mutual inductor structure, the coupling coefficient k is defined as: L21 k ¼ pffiffiffiffiffiffiffiffiffiffiffiffiffi L11 L22
ð25Þ
Fig. 7 shows the self and mutual inductances of the mutual inductor structures with different values of d. As the single inductor structures are almost the same, the values of self inductances are all about 2.2 nH at 2.5 GHz. With decrease of the variable d, the mutual inductance increases. The concrete values of self and mutual inductances at 2.5 GHz are listed in Table 1. For the specific mutual inductor structures, the relationship of d and L21 can be obtained by numerical fitting, so as to offer design reference. Fig. 8 shows the fitting result. It can be seen that d and L21 are linearly dependent to a first approximation. The linear fitting expression is: L21 ðnHÞ ¼ 0:008dðumÞ þ 1:22
ð26Þ
After design of the schematic and the physical structures of inductor, capacitor and mutual inductor, the final step of filter design is to implement the whole layout. However, the unavoidable parasitic effect usually makes the layout design a complex and formidable task. In this part, we present a convenient and accurate layout design method named the schematic-layout mutual feedback method. The key point of the method is to consider the component value shift and parasitic effect into schematic simulation. A concrete layout design example is given below. In this example, the layout is simulated with ADS (Advance Design System). The passband of the designed filter is 2.4–2.5 GHz. The locations of the two finite transmission zeros are chosen to be 1.9 and 4.8 GHz, which are major interference frequency bands. The first step of the schematic-layout mutual feedback method is to design the initial layout called layout V1. The layout V1 is designed according to the schematic V1. Fig. 9(a) shows the schematic V1 and the values of the components are listed. Variable R is the parasitic resistance of the inductor. Fig. 9(b) shows the simulation structure of layout V1. Fig. 9(c) compares the simulation results of the schematic V1 and layout V1. For the layout V1, the location of the transmission zero in upper stopband shifts toward low frequency compared to schematic V1. This is caused by the value shift of the inductor, capacitor and mutual inductor structure, and the parasitic effect. As the layout simulation considers the loss effect, the insertion loss of layout V1 increases compared to schematic V1. The second step is to adjust the schematic V1 so as to obtain the schematic V2. The adjusting criterion is to match the response of the schematic V2 with layout V1. If their responses are matched, it means that the schematic V2 considers layout parasitics and component value shift correctly. The adjusting process contains tuning the component values and adding extra parasitic components if necessary. From schematic V1 to schematic V2, the component values are changed as: C5: 0.08-0.12 pF, C6: 2.9-2.7 pF, LL2 and LL3: 2.22-2.44 nH, M: 0.95-0.91 nH, R: 0-1.83 Ohm. No extra parasitic components are added. Fig. 10 compares the filter responses of schematic V2 and layout V1. Their responses match quite well. The final step is to optimize the schematic V2 so as to obtain the schematic V3, which conforms to the filter design index. Then feed back the schematic variation to the layout V1 and get layout V2. As schematic V2 and layout V1 are equivalent, their component variations should be the same to obtain the desired filter response. From schematic V2 to schematic V3, the component values are changed as: C5: 0.12-0.04 pF. Thus, the layout optimization from layout V1 to layout V2 should be: C5: 0.08-0 pF. The variation of C5 is 0.08 pF for both the schematic and layout
J. Pan et al. / Microelectronics Journal 49 (2016) 43–48
Port 1
M
C1 C3
C2 Port 2
LL2 R
LL3 R
C4
C6
Variables C1=1.61pF C2=1.61pF C3=1.15pF C4=1.15pF C5=0.08pF C6=2.90pF LL2=2.22nH LL3=2.22nH M=0.95nH R=0Ω
0 -5 S Parameter (dB)
C5
47
S11
-10 -15
S21
-20 -25 -30
Schematic V3 Layout V2
-35 0.0 0.5 1.0 1.5 2.0 2.5 3.0 3.5 4.0 4.5 5.0 Frequency (GHz) Fig. 11. The comparisons of transmission and reflection responses of the schematic V3 and layout V2.
0.85um
0
S Parameter (dB)
S11 -10 -20
S21
-30 -40 -50 0.0
Schematic V1 Layout V1
0.5 1.0
1.5 2.0 2.5 3.0 3.5 Frequency (GHz)
0.85um 4.0 4.5 5.0
Fig. 9. (a) The schematic V1 and the values of the components. (b) The simulation structure of layout V1. (c) The comparisons of transmission and reflection responses of the schematic V1 and layout V1.
S Parameter (dB)
S11
-10 -15
S21
-20 -25
The concrete layout design example has presented the schematic-layout mutual feedback design method. Obviously, based on this method, the layout can be designed simply and accurately.
5. Experimental results
0 -5
Fig. 12. The microscope photo of the fabricated filter.
Schematic V2 Layout V1
-30 0.0 0.5 1.0 1.5 2.0 2.5 3.0 3.5 Frequency (GHz)
4.0 4.5 5.0
Fig. 10. The comparisons of transmission and reflection responses of the schematic V2 and layout V1.
adjustment. Fig. 11 compares the filter responses of the schematic V3 and layout V2. The response of layout V2 is consistent with schematic V3. Also, the locations of the transmission zeros of layout V2 are in the desired positions, which are about 1.9 GHz and 4.8 GHz, respectively.
A silicon-based IPD band-pass filter has been designed according to the above design method. It is fabricated with the 0.35um IPD process in SMIC (Semiconductor Manufacturing International Corporation). The passband of the fabricated filter is 2.4–2.5 GHz. The locations of the transmission zeros are chosen to be 1.75 GHz and 4.8 GHz for project need. Fig. 12 shows the microscope photo of the filter. Its size is about 0.85 mm 0.85 mm 0.35 mm. Fig. 13 presents the EM simulation and test results of the fabricated filter. The EM simulation is conducted by ADS and HFSS (High-Frequency Structure Simulator), respectively. Obviously, the simulation result of HFSS matches quite well with the test result. As the filter is fabricated by front-end process, the process deviation is little. For the simulation result of ADS, the location of the transmission zero in lower stopband shifts toward high frequency compared to test result. The frequency offset is about 70 MHz. This is probably caused by the deviation of ADS momentum simulation. Analyzing the test data, the passband insertion loss is better than 2.3 dB. The passband insertion loss can be optimized by using thicker conductors and replacing aluminum layer with copper
48
J. Pan et al. / Microelectronics Journal 49 (2016) 43–48
in this paper has the smallest size while achieving equivalent electrical performance.
0
S Parameter (dB)
-5
S11
-10
Acknowledgment
-15 -20 -25 -30
S21 Test Simulated by ADS Simulated by HFSS
-35 0.0 0.5 1.0 1.5 2.0 2.5 3.0 3.5 4.0 4.5 5.0 Frequency (GHz) Fig. 13. The comparisons of EM simulation (ADS and HFSS) and test results of the fabricated filter.
layer. The fluctuation of passband is little. For the transmission zeros, their locations are accurately controlled. The suppressions of 1.75 GHz and 4.8 GHz are about 29 dB and 27 dB, respectively. The suppression degree is good. The size of the filter is about 0.85 mm 0.85 mm 0.35 mm. In Ref. [8], size of the bandpass filter with same center frequency is 2.50 mm 2.00 mm 0.80 mm. To the best of our knowledge, the fabricated filter has the smallest size while achieving equivalent electrical performance. The experimental results verify the advantages of introducing freely tunable transmission zeros and using mutual inductor structure. Also, it proves the correctness of the whole design flow and method.
6. Conclusion In this paper, a compact silicon-based IPD band-pass-filter with two independently tunable finite transmission zeros are designed and fabricated. Based on related circuit analysis, specific schematic is introduced to ensure two independently tunable transmission zeros. The mutual inductor structure is presented to realize the filter, which greatly decreases the filter size. The design and analysis method of the mutual inductor structure is shown. For the layout design, the schematic-layout mutual feedback method is presented so as to design the layout simply and accurately. Finally, the experiment results verify the correctness of the whole design flow and method. To the best of our knowledge, the designed filter
The authors appreciate the support of National Science and Technology Major Project under project No. 2014ZX03001015-003 and Daquan Yu would like to thank the support of the One Hundred Person Project of the Chinese Academy of Sciences. This paper’s proposed mutual inductor structure is patent pending.
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