Electronic circuits used in a semiconductor detector alpha spectrometer for coincidence measurements

NUCLEAR INSTRUMENTS AND METHODS 54 (t9fi7) igo--i98 ;

NORTH-HOLLAND PUBLISHING CO .

ELECTRONIC CIRCUITS USED IN A SEMICONDUCTOR DETECTOR ALPHA SPEC'CROMETER FOR COINCIDENCE MEASUREMENTS S. S. KLEIN*, L. HULSTMAN and J . BLOK Nataurku

Uboratorium drr Vr(je Universiteit, Aitateraant, The Netherlands Received 2 June 1967

An alpha spectrometer was developed for the purpose of coincidence measurements at high counting rates while retaining high resolution and stability. The alpha resolution, measured using a Th(B+ C) source, varies from 19 keV at counting rates below 2000 cps to Z.7 keV at 7400 cps; indications are given how this upperlimit may be improved . The stability is about 0.2 keV/h or 2 keV/day; a significant part of this is caused by temperature

elects in the detector . It will be very difficult to attain better stability with semiconductor detectors unless ambient temperature is stabilized. Consideration is given to the use of field effect transistors (FETs) in the input stage and transistors in the remainder of the amplifying circuits. It is concluded that the resolution may be unproved only by using cooled FETs .

1. introduction

to the third and fourth excited states of x°'Tl (fig . 1) which are only 19.5 keV apart. A good time resolution was important to attain a high ratio of true to chance coincidences at high counting rates. In view of the advantages with respect to space and power requirements we have carefully considered the possibilities for replacement of vacuum tubes by transistors (sect. 2). It appears that it is difficult or even impossible to obtain the stability and resolution we needed without using vacuum tubes in many places . To avoid the drawbacks of hybrid circuits we decided to use all-vacuum tube circuitry. Schematics and circuit description are given in sect . 3. The results obtained are given in sect. 4.

The properties of semiconductor detectors for charged particles have been adequately described 1). They have been used extensively wherever a small device for alpha spectrometry with reasonable resolution could be useful . For ultra-high resolution spectrometry of alpha emitters with reasonable lifetimes ofcourse magnetic spectrometers are superior by far, and no new possibilities were created in this field by the availability of semiconductor detectors. For coincidence measurements the situation is drastically different, however, because semiconductor detectors allow large solid angle detection without deterioration of the resolution . An apparatus for the measurement of o(-y angular correlations in Th(B + C) decay was described by Cobb') . Much attention was paid to the attainment of a short alpha pulse rise time. Resolution was 100 keV at 5700 cps, stability 15 keV/h. A resolving time of 9 nsec was attained . Goulding and Nakamura developed an a-y coincidence spectrometer for the investigation of low-intensity a transitions (down to 10- ' of the ground state transition) in which a counting rates of about 103 cps were used . This apparatus was used by Bjornholm and Lederer') for the investigation of nuclides in the actinide region ; therefore the y counting rate was relatively low, and a coincidence resolving time of 40 ns was sufficient . Resolution and stability were 40 keV and 12 keV/day, respectively . For our experiments on oc-y angular correlations in Th(B +C) decay') we needed an a spectrometer with a stability of better than 1 keV over several hours and a resolution better than 20 keV . These properties were necessary because it was planned to investigate separately the a-y angular correlations corresponding

* Present address : Technische Hogeschool, Eindhoven. 616

492 472

40 0

208

TI

Fig. 1. Level scheme of 20171 . 190

19 1

A SEMICONDUCTOR DETI?CTOR ALPHA SPECTROMETER

30 ns rise 1 ms decay

Fig. 2. Block scheme . The coincidence unit and 512 channel analyzer illustrate a possible application of the a spectrometer described in the text . Pulse forms are shown as observed on an oscilloscope when Th(B+C) is used as a source. To observe the low intensity lines a counting sates in excess of 2000 per second are necessary (212Bi a lines are indicated as follows: ao goes to ground state of 'ogT], al to the 40 keV state, a2 to the 328 keV state, etc.) .

2. Preliminary considerations 2.1 . GENERAL

The problem may be stated as follows : pulses from a semiconductor 2 detector are to be amplified with minimum noise and with stability better than 10 -4 over several hours. The a pulses have to be fed to the pulse shaper of a fast coincidence unit ; therefore a short risetime is desirable . (In investigations of a radiation from radioactive sources, this requirement is mitigated somewhat by the small energy region in which decay is possible . For a single isotope, energies seldom differ even by 10% .) If a fast pulse is taken off directly after the preamplifier, risetime considerations apply only to this part of the amplifier chain. The desired stability and resolution are attainable only if at some point the linear amplifier chain is interrupted and a biased amplifier inserted to cut off the low energy part of the spectrum and make a more detailed observation of the interesting spectrum part possible . The stability requirements are somewhat less after the thre, id. The block scheme that results from these considerations is shown in fig . 2. 2.2 . CONDITIONS FOR MINIMUM NOISt

It is assumed that only the input device contributes

significantly to the noise of the system . The contributions of shot noise, resistance noise and flicker noise in various parts of the input circuit are indicated in table l . The vacuum tube data were taken from Goulding') ; the FET contributions in lines l b and 4b were estimated froln recent publications 6 ° ? ) descrihinta TABLE 1 Noise contributions from carious wurt - es Noise source Ia . Anode current shot noise I b. Thermal noise in FET channel 2. Shot noise in detector and grid (gate) leakage current 3 . Thermal noise in input resistance 4a . Tube flicker noise 4b . FET flicker noise

! 0.019 C"-'(Sr) 2 x 10-5 TC'2'(Sr) 0. 16 ri 8 -r ;: R 0.C002 C unpredictable, but low for selected FET .

The table has been calculated for equal RC integrating and differentiating time constants. N = mean square fluctuation, etipressed in kev~ far Si dctoctors . C = total input capacitance (pH, S --= transconductance of the input &\ ices («1 A r - RC time of the shaping constants t,<
K t.lï 1 N

S4 . 8,

192

until the total input capacitance equals the detector capacitance, if detector current dominates [ref.`'), appendix A]. For the detector in the last column of table 3, this optimuni is rcached for 4 F88CC, 3 F 180F, E810F and about 6 1° ETs. with rcsp . 8 .8 keV, 8 .7 0.76 walls)* 0.35 Mi) keV ; 7.6 keV and 5.8 keV (cooled FETs) or 8.1 keV 0,57 (#0' 0.19 C(si)-1 (room temperature FETs). 0.45 (CsilS)* 0,12 C(Si)-* _ t __ Therefore the 7.8 keV obtainable with one E81OF is only by noise monoenergetic peak broadened significantly improved only for cooled FETs . It is R, the fwhm of a (pulser peak) is approximate! as 2.3 times the sum of the conquestionable if the small gain in ac resolution which will tributions in table 1 with exception of flicker noise. This approxibe obtained (taking account of the difficulties associmation is certainly valid for the case of dominant detector the dense a tracks) capacitance and current, but may be optimistic for low capaci- ated with charge collection in warrants the effort necessary to surmount the technical tance detectors. R is expressed in keV for Si detectors. as in table I and are the same meaning The other symbolshave difficulties that go with cryogenic techniques . Certainly expressed in the same units. the trouble to select FETs is not paid unless it is very preamplifiers using 2N3823 FETs . Only vacuum tubes important to avoid vacuum tube circuits . and FETs are considered. [Theoretically, tunnel diodes might have advantages ; an attempt by Jonassona) to 2.3. RismmE It follows from very general considerations applying realize these wasnot successful . The practical difficulties are, moreover, considerable because the favourable equally to systems with and without feedback, that the working region is very small and inherently instable .) risetime for a one stage amplifier configuration is Supposing the contributions of thermal noise in the approximately equal to 2.3 R.CQ ; in this Rp is an input resistance and flicker noise to be negligible, one equivalent anode resistance with a value AIS (A is the may calculate the optimal time constant and the voltage gain, S is the transconductance) and C., is the corresponding noise line width (table 2). These ex- sum of the capacitances seen from the anode (drain). This formula may be applied to a cascode amplifier pressions have been evaluated for typical cases in table 3. We may then conclude that for oc detectors, as well, provided that R.C. is lace compared with where detector capacitance and leakage current CkjSp (Sr is the transconductance of the "upper" dominate, an E81OF vacuum tube is superior to a FET tube, Ck the capacitance seen from its cathode) . Some at room temperature as input dt- vice, even if the FET is consideration is to be given to the much used bootstrap carefully selected to approach he theoretical minimum 4ircuits . It is certainly not true that the power gain of equivalent noise resistance . A selected FET may be provided by the bootstrapping cathode follower makes superior by a small amount if it is cooled to the shorter risetimes possible . The bootstrap part is to be observed optimal temperature of about 120'K. seen as an electronic simulation of a large anode resisParalleling FETs or tubes improve the characteristics tance; the inherent time delay may well cause oscillation

TAnLE 2 Time constant r for minimum noise and corty width R.

Wing noise peak

TABLG 3

Evaluation of r and R for some typical cases. Detector capacitance and leakage current

Input device characteristics Input device

E88CC EI80F E810F 2N3823 (21W K) 2N3823 (120° K)

S 10

12 .5

14,1

1

?8 8 8

is

17,5

I,

55 4 6

11

0.15 0.05

Negligible =0, id =0)

(Cd

I !

Intermediate (Cd =5,i,ß=8) 2.8 2.9 2.8 2.5 1 .6

Dominant ~ (Cd =50,în=100)

I

1

j

0.60 0.54 0.37 0.58 0.28

' 1

10 9.5 7.9 9 .7 6.9

capacitance of input device + 3 pF for parasitary capacitance . leakage current of input device + equivalent leakage current of grid (gate) and detector resistance (50 Mil for vacuum tubes, 1000 MQ for FETs). Cd, id : detector capacitance and leakage current. Other symbols as in xable 1 .

Cg : iR :

A SEMICONDUCTOR DETECTOR ALPHA S'ECTROMETER

tendencies when an attempt is made to attain tt short risetime by increasing feedback . Therefore preference is to be given to the input device for which h C., is a In inimttln . I'or :1 = 20, R (" is - h n%cc I'()r an 1:8101 : and -24 nsec for a 2N 3821 . It is clear that the E8IOF is superior in this respect . or many semiconductor detectors, however, the collectiontime is longer than the risetimes that follow from the above calculation'') . The advantage to be gained by using the E81OF is important only for detectors in which the base material has a resistivity below 4000 0-cm. 2 .4 . CHOICE BETWEEN SEMICONDUCTOR TUBE CIRCUITRY

AND VACUUM

In sect. 2.2. and 2.3. it has been shown that the E81OF is the best choice with respect to noise (excluding cooled FETs) and risetime. The choice between transistors and tubes in the remaining circuitry was made on the following grounds : 1 . Te minimize the effects of alinearity in the biased amplifier threshold and threshold fluctuations the input pulse should preferably be several volts above threshold . When a threshold of 90°,1o is desired, this sets the input voltage at several tens of volts . It will be difficult to achieve amplification of sufficient stability up to this level with transistors . Therefore the end stage of the main amplifier should be a vacuum tube . 2 . A biased amplifier based on semiconductor elements is necessarily dependent on ambient temperature ; compensation methods depend on matching the

19 3

characteristics of elements that are mn`tly in verv different situations with respect to working point, and anyhow not necessarily at the same temperature . In vacuum tube circuits, a dependence oil filament voltage v, observed . It is, how~-,ver, much easier to stabilize filament voltage than to stabilize ambient temperature . Therefore the biased amplifier should have a vacuum tube threshold circuit . 3. There now remain only (in our case) three amplifier tubes and some cathode followers to be replaced by transistors . It is probably not advantageous to provide separate low tension power for these parts . Anyhow, temperature gradients in the vicinity of the necessary vacuum tubes will be more severe, and this will impair stability of the semiconductor parts . For these reasons we decided to use no semiconductors . Perhaps a biased amplifier based on current switching techniques will avoid the difficulties described above . In this case the decision would have to be remade . 3. Circuit description 3 .1 . PREAMPLIFIER (fig. 3) In accordance with the results obtained in section 2 the E810F was chosen as the input tube. A bootstrap scheme was avoided to make it possible to achie\ e a fast output pulse . A high gain of the cascode input stage was attained instead by using one half of an E88CC as upper tube . An anode resistance of grout 10 kQ is thus made possible, provided that the sur_'lus

Fig . 3 . Preamplifier .

s. s. K tq IBIN et al.

194

were obtained, the risetime depending on detector capacitance and collecting time . Shorter risetimes have been obtained by increasing the feedback and using it low resistivity detector . The stability of the preamplifier was better than 0.01%. To treasure this, higher pulses than normal had to be used . We assume that the same properties will

apply for small pulses, however. .2.

Fig. 4a. Main amplifier: General setup including fanout to fast coincidemx- pulse shaper and shaping circuits for high resolution output (--A : inverter, +1 : white cathode follower. Asterisks indicate high stability resistors) .

anode current of the E810F is drained by a resistance connected to the anode supply voltage, or that the cathode of the E88CC is ac coupled to the anode of the E810F. The last possibility was chosen to effect a stable working point of the E88CC. This tube might have been driven easily into cutoff or saturation by relatively small changes in the E810F anode current in the first case . The anode voltage of the E810F was kept low to ensure a small grid current. This was made possible by adjusting the screen voltage independently, coupling the screen to the anode by a large capacitance to avoid partition noise and to profit from the grid-screen transconductance. The cascode was coupled to the output by a White cathode follower. Output pulses of about 10 mV per MeV with risetimes of 30 to 50 nsec

3x E 180F

MAIN AMPLIFIER AND PULSE FORMING CIRCUITS

The main amplifier (fig . 4a) consisted of two almost equal feedback loops with White cathode follower output stages, separated by a pulseforming circuit consisting of a short-circuited delay line differentiating stage and a RC integration stage. Before the first loop there is an RC differentiating network with RC = 10 Asec to avoid trouble from pile up of preamplifier pulses (decay time several msec). The undershoots caused by this after the delay line differentiating stage were removed by inserting a small resistance instead of the shor, circuit. At the input there was also a 5 lisec differentiating network connected to an inverter tube feeding the a input of the fast coincidence unit . The feedback loops contained a differential amplifier receiving the input pulse on one input and a constant fraction of the amplified pulse on the other input . This configuration is preferable to the conventional one in which two amplifier tubes and a cathode follower are used in the feedback loop because of the superior overload characteristics of the differential amplifier E 88CC

IN

Fig. 4b . Main amplifier : Feedback loop (Asterisks indicate high stability resistors) .

195

A SEMICONDUCTOR DETECTOR ALPI-IA SPECTROMETER

Fig. 5a. Simple bias circuit with series diode. input stage. As a differential amplifier has two outputs with equal and opposite amplification it seemed possible to use more than one tube between the differential amplifier output and the feedback input to achieve a high feedback gain and therefore a high degree of stability. A trial with two tubes outside the differential amplifier, however, proved it was difficult to avoid oscillation. As it was to be expected that this tendency would impair the reliability and stability of the amplifier, only one tube was used between negative output and feedback network . The resulting scheme is given in fig. 4b . The open loop gain was 13000. The closed loop gain %vas 66 ; this gives a stabilization factor of about 200. With increased feedback it was possible (with a careful layout) to attain a stabilization factor of 500. When the integrating stage was left out, the amplifier had a risetime of 40 ns for signals up to about 40 V. For large positive signals the risetime was worse because in the rising edge of the pulse the output tube was cut off. A risetime of 100 nsec was observed for positive pulses of 80 V. For negative pulses the risetime

deteriorated only at very high valttcs of dIL output. pulse height .* For both positive and negative pulses a rnammum output pulse height of more than 100 V was possible . In out vase the risetime of the pulses presented Rio difficulty, as the risetime of the input pulses to the second loop was always more than 0.4 psec due to the action of the integrating stage. The stability of the amplifier against variations of the tube parameters was excellent. To test this, the transconductance of one of the tubes was varied down to only 5% of the ideal value by increasing the negative grid potential. Still the amplifier was working correctly and hardly any variation of gain had been observed . For a satisfactory temperature stability it appeared necessary to screen the White cathode follower circuits well from the feedback loops. Although this effect is not understood, it was observed at several trials . In the screened configuration the gain was proved to be constant to within 0.004+-0-004- 1%1'C, whereas removal of the screening caused a temperature effect of 0.033%/'C. 3.3 . BIASED AMPLIFIER

This amplifier was built as indicated by Goulding') but the input circuit was modified to achieve a higher degree of temperature stability and to make vc more suitable for high counting rates. Fig. 5a give,, the basic As long as the

ee-back pulse is not approximately equ il to the input pulse th,. grid pulse at the output tube is large. This causes cutoff for positive output pulses . For negative output pulses the tube is driven in to the grid current region, but this has no serious ;.onseyuences because the grid current is dra~~ fi only for a very short time . Overload characteristics will be worse for negative pulses, however.

l

Fig. 5b . Modification of the circuit of fig. 5a to achieve better properties at high counting rates. The circuit is folïo~tie, 1 by amplifier to attain the optimal dispersion of the observed spectrum fraction .

.1

variable

S. S. KLEIN

input circuit. D, is reverse biased until Vjt4 exceeds V,g,As ; after this VOUT rises to VIN - P'lljAs, and remains there for some time after VIN disappears . At high courting rates, this stretching effect causes overload symptoms in the following circuitry. The effect was diminished by adding an inductance in series with a smaller resistance R' to speed up the descent of dour while retaining a reasonably flat top. The temperature stability was improved by replacing the semiconductor diodes in Gouldings scheme by one halfof an EAA 91 . The. other half was used in the compensation circuit described below. When a pulse is transmitted by the series diode D , a certain charge is carried away from the anode and must be restored in another way. At low counting rates a resistance connecting the anode to the bias supply suffices. The current necessary to attain equilibrium again causes a certain voltage drop across the resistor

et ai.

R. however. This voltage drop is equivalent to a change of the bias voltage as long as the recharge current is present. A pulse arriving in this period will therefore be transmitted by the diode with a diminished amplitude. At higher counting rates, thc average voltage drop causes a shift of the spectrum, and the fluctationz, of the instantaneous voltage drop with respect to the average result in fluctuations of the shift, that is worse resolution. A clamping diode across the resistance mum a faster recharge, but as the value of the voltage drop is only small, and the diode is not an ideal one, the worst part of the effect remains. One therefore has to assist the clamping diode somehow. Preferably this should be done by the pulse itself, or a pulse derived from it, as the statistical fluctuations in the charge are then compensated by those of the discharge. An obvious solution of this problem is to make the circuit symmetric around the input condenser

U40

l

104

a

U) z

n

r-N

U

lu

3

a492 et,

-10 1

200

Fig. 6. An a spectrum of Th(B+Q

300 ---- CHANNEL NUMBER

400

taken at about 100 cps. The peaks are indicated in the same way as the x transitions in fig. The spectrum was taken in about 3 h ; no degradation of the resolution due to shifts in amplification is observable.

197

A SliMICONDLICTC)R DETECTOR ALPHA SPECTROMETER

during which D, is forward biased . By inserting an adjustable series resistance It, the correct discharging current may be set. This can be done by observation of the baseline at the anode of l.)t on an oscilloscope . The correct setting depends on the bias of 1), and on the pulse height . The fluctuations are exactly compensated only when the input pulses represent a monoenergetic spectrum . Although in a spectra from one source this will be approximately true, it does not apply when different sources are present. This will occur when daughter or mother activities are present or when several decay modes are possible like in our investigation of ThC decay . As the 6 MeV a pulses from ThC and the 8 .78 MeV pulses from ThC' differ by 30°fß, the circuit was undercompensated for the latter but overcompensated for the former . This caused a remaining fluctuation that became apparent in the spectra at counting rates of 5 000 per second or more. The effect may be diminished by giving D1 some backward bias (e.g. by inserting an adjustable resistance between A and B in fig. 5b). The amplifier indicated by Goulding after the bias circuit was not changed except for the values of some coupling condensers which were augmented to diminish the effect of undershoots at higher counting rates. An extra amplifying stage was sometimes added to permit observation at a magnification of up to 10 mV/keV. The stability of the biased amplifier (inclusive bias circuit) was better than 0.01°,, between 15 and 30"C . , if filaineni voliage tivas stabilized to I°'

a0'

L

200

250

300

" CHANNEL NLIMB`ïR

Fig. 7 . Detlil of fig. 6 to show the possibility to do separate experiments on 2492 and 24 7 2.

and to feed it with bipolar pulses . This would give very exact compensation, but demands duplicating a large part of the circuit, and the simplest way to make a symmetric pulse, double delay line differentiation, causes a tripling of the noise power compared with single delay line pulseforming. [A method for attaining symmetric output pulses without adding noise is given by Blalock9). It should be very useful in extreme cases.] A simpler solution is the following one (fig . 5b). The pulse arriving at the input is fed directly to the anode of the series diod ,_- D I , and by a delay line of about 1 psec to the -  -J e of the discharge diode D2 . As this pulse arrives at the time the main pulse has started to decay there is an appreciable time interval

z

4. Results A resolution of about 19 kek' was acl:ieved

",0 4

0 U

CL IJLJ

z

10 3

233weV twhm

10

2

-15

0

15

-15

0

15

-15

0

15

-15

0

CHANNEL NUMBER RELATIVE TO CC, o Fig . 8 . Spectra of the high intensity pair, a  and 'X40, at different counting rates .

5

itigs .

198

3. 3. tt LEI N et 01.

and 7). The properties at high counting rates may be derived from fig. 8 . Stability of the amplifiers was checked with a precision pulse generator. Results for the separate units have been given above. A temperature, ca cient of0.004 ± 0.0040/x/° C was observed for all units together, connected as indicated in fig. 2. When a test measurement with detector pulses was performed, a temperature coefficient of 0.035 f 0.004°yo/°+C was measured. This temperature dependency is probably a detector effect; the value of the coefficient is in close agreement with the temperature dependence of the electron-hole pair creation energy which may be deduced from the measurements of Fabri, Gatti and Svelto '}. It was considered to compensate the effect by inserting a temperature dependant attenuator . As it is probable that overall stability would not be improved unless the detector and compensating network would be in close thermal contact, it was decided nit to use this compensation technique . This decision proved to be justified in the course of a long series of angular correlation measurements, in which changes of VB1s were seldom necessary with fresh detectors. After detection of about 10' particles/cm', however, stability became worse for each detector, to be excellent again for the following one. 5. Conclusion The apparatus described is a significant improvement

on hitherto described a spectrometers with respect to the combination of resolution, stability, high counting rate capacity and output pulse risctime . It is, however, not useful where a large number of detectors :tic to be

used in a small space. For these cases semiconductor circuitry will have to be used. The authors want to thank Mr. H. Ton for his assistance in the preliminary experiments. The assistance of Mr. A. Pomper in drawing the figures and of Miss C. Janssen in typing the manuscript is gratefully acknowledged . References 1) G. Dearnaley and D. C. Northrop,

Semiconductor counters for nuclear radiations (E . &. . F . N. Spon, London, 1963); J . M. Taylor, Semiconductor particle detectors (Butterworths Inc.,

1963). $) W. C. Cobb, Phys. Rev. 132 (1963) 1693 . 3) S. Bjarnholm, C. M. Lederer, F. Asaro and 1. Perlman, Phys. Rev. 130 (1963) 2000. d) S. S. Klein. Thesis Wrijje Universiteit, Amsterdam, 1966). 5) F. S. Goulding and W. L. Hansen, Nuci . Instr. and Meth. 12 (1%1)249 . s) T. V. Blalock, IEEE Trans. Nucl . Sci. NS-13, no . 3 (1966) 457. 7) K. F. Smith and J. E. Cline, ibid. 468. e) L. G. Jonasson, Nucl . lnstr. and Meth. 26 (1964) 1(4. 9) T. V. Blalock, Rev. Sci. Instr. 36 (1965) 1448. 1°) G. Fabri, E. Gatti and V. Svelto, Phys. Rev. 131 (1963) 134. 11) P. A. Tove and K. Falk, Nucl . Instr. and Meth. 12 (1961) 278, 29 (1964) 104.