Fractally slotted patch resonator based compact dual-mode microstrip bandpass filter for Wireless LAN applications

Int. J. Electron. Commun. (AEÜ) 107 (2019) 264–274

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Regular paper

Fractally slotted patch resonator based compact dual-mode microstrip bandpass filter for Wireless LAN applications S. Karthie ⇑, S. Salivahanan Department of Electronics and Communication Engineering, SSN College of Engineering, Kalavakkam 603 110, Tamil Nadu, India

a r t i c l e

i n f o

Article history: Received 25 January 2019 Accepted 25 May 2019

Keywords: Dual-mode filter Fractal filter Bandpass filter (BPF) Fractally slotted resonator (FSR) Transmission zero (TZ)

a b s t r a c t A miniaturized dual-mode microstrip bandpass filter (BPF) based on fractally slotted patch resonator is presented in this paper. The proposed filter is centered at 2.4 GHz desirable for Wireless LAN applications. The dual-mode filter is realized using the fractally slotted structure on an octagonal patch resonator with perturbation. The design offers transmission zeros (TZs) at 2.19 GHz and 2.66 GHz on either side of the filter passband ensuring high selectivity. The locations of TZs from the center frequency can be adjusted by tuning the perturbation size. The filter is fabricated on a substrate with high permittivity and found to have a 3-dB fractional bandwidth (FBW) of 5% in the passband. Moreover, the measured minimum insertion loss is less than 0.91 dB and the return loss is greater than 23 dB in the passband. The prototype filter is fabricated and tested. Good agreement is accomplished between the experimental and simulation results. Furthermore, the footprint of the fabricated filter is about 0.416kg
0.416kg where kg is the guided wavelength at 2.4 GHz. Ó 2019 Elsevier GmbH. All rights reserved.

1. Introduction Modern wireless communication systems require planar microstrip bandpass filters (BPFs) that can meet the demand of high selectivity, low insertion loss, high return loss, narrow bandwidth and compact size [1]. Increasing the order of the filter leads to improvement in selectivity and reduction in insertion loss at the expense of increase in the size of the filter structure due to the increase in the number of resonators. The dual-mode filter structure is a typical microwave BPF used to improve the selectivity by introducing transmission zeros (TZs) on either side of the passband and hence, these filters are of great interest to the microwave community. The attractive feature of the dual-mode filter is its compactness, as the number of resonators needed for a given degree of the filter is reduced by half. In addition, dual-mode filters often demonstrate elliptical or quasi-elliptical frequency responses resulting in high selectivity [1,2]. Several commonly used dualmode microstrip structures including crossed-slots [3,4], loaded crossed-slots [5], right crossed-slots [6] in a square patch and also coupled lines [7–9] are reported in the literature. Nevertheless, the filter structures proposed above occupy a fairly large circuit area. In order to reduce the circuit size, numerous resonator structures ⇑ Corresponding author. E-mail addresses: [email protected] (S. Karthie), [email protected] (S. Salivahanan). https://doi.org/10.1016/j.aeue.2019.05.037 1434-8411/Ó 2019 Elsevier GmbH. All rights reserved.

such as stepped impedance resonators (SIR) [10–12], loop resonators [13–15], ring resonators [16,17], and patch resonators [18,19] based on dual-mode configuration have also been reported in the literature. Recently, there has been considerable interest in using metamaterial transmission line (TL) structures for the design of microwave filters and antennas for further size reduction and high selectivity to realize the dual-mode operation. To achieve dual TZs with high selectivity in the passband, a novel composite right/left-handed (CRLH) metamaterial TL using Koch shaped extended complementary single split ring resonator (K-ECSSRR) pair as a dual-shunt circuit for harmonic suppression has been explored in [20]. Also, a physical implementation of this circuit as a diplexer has been analyzed. A novel dual-band double negative zeroth order resonator (DNG ZOR) antenna using a Wunderlich shaped extended CSSRR pair is proposed in [21] with the dual-mode operation. In [22], a dual-mode meta-atom with coplanar coupling is proposed for simultaneous large phase cover and high transmissions and its possible application in quad-beam transmit array is conceived and realized. Moreover, in [23], a novel memristor based multilayer dualmode resonators for BPF design are explored with a reconfiguration property. However, it is limited by the circuit complexity. In [24], an electronically reconfigurable (varactor-tuned) microstrip dual-mode filter with non-uniform Q lossy technique has been demonstrated but with a large size.

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Furthermore, in recent years, fractal structures are also gaining significance in the design of filters and antennas due to its selfsimilarity and space-filling property [25] that simultaneously provides multi-band response and miniaturization. Different fractal geometries such as Sierpinski [26,27], Minkowski [28,29], Koch [30,31], Greek-cross [32], Peano [33], and Moore [34] have been deployed as resonator structures in a square patch for the design of fractal BPFs. Inspired by the fractal concept, a wide variety of practical applications using metamaterial TL resonators have also been reported. A 3D multi-beam highly-directive lens antenna using gradientrefractive-index (GRIN) metamaterial based on closed Sierpinski fractal ring with flexible beam numbers and beam directions in super-extended C band was proposed and demonstrated in [35]. A novel octave-bandwidth highly-directive 3D half Maxwell fisheye (HMFE) lens antenna using GRIN metamaterial based on closed Sierpinski fractal ring, also in super-extended C band was proposed and experimented in [36]. To reduce cross-polarization radiation pattern of the microstrip antenna, a novel LH-TL metamaterial based on the fractal-perturbed mushroom structure using Koch Island was designed and investigated in [37]. In [38], a novel 3D LH-TL super lens using Sierpinski fractal geometry was demonstrated theoretically and experimentally to improve the imaging resolution at 5.35 GHz. A fully fractalperturbed resonant-type CRLH-TL structure using Koch-curveshaped CSRR was proposed in [39] and its possible application as Wilkinson power divider has been designed, fabricated and tested. A compact triple band absorber enhanced by a 2D artificial LH-TL metamaterial based on the fractal-perturbed element using quasi-Sierpinski fractal ring has been designed, fabricated and validated in [40] with high absorption peaks in S, C, and X bands, respectively. A more compact broadband balun operating at 1.5 GHz is proposed in [41] based on CRLH-TL structure using T-Koch fractal curves having FBW of 83.3% from 1 to 2.25 GHz. In [42], a novel fractal-perturbed resonant-type CRLH-TLs using Sierpinskishaped cascaded CSSRR and Minkowski-loop shaped CSRRs are proposed and its simple applications in the effective design of a dual-band (DB-BPF) and a mono-band (MB) branch-line coupler (BLC) are exploited. Although both the dual-mode design [20–22] and the fractal-shaped resonators [35–42] using metamaterial TL structures result in miniaturization, the high insertion loss in filter and low gain in antenna restrict the applications of metamaterials to some extent. The novelty of the proposed work is the design of a dual-mode octagonal patch resonator which is a very attractive structure for developing narrow, compact and high performance microwave BPF with fully planar fabrication techniques. In this paper, a miniaturized, highly selective dual-mode BPF is presented with fractal slots on an octagonal-shaped microstrip patch resonator. The novelty of the design is the etching of fractal slots up to the third iteration in the octagonal patch coupled with two V-shaped coupling arms. The space filling property of fractal shapes prompts fitting of very long electrical lengths into minimal physical spaces leading to miniaturization. First, by increasing the order of iteration of the fractally slotted resonator (FSR), the resonant frequency can be shifted towards the lower frequency, indicating a miniaturized property. Second, by adding a square perturbation element at the corner of the octagonal patch, the dual mode can be obtained in the frequency response. Consequently, a third order FSR is used to design a high-performance narrow band dual-mode BPF for applications in Wireless LAN (IEEE 802.11b/g/n) networks. Finally, a compact dual-mode fractal BPF of this type is designed and fabricated to demonstrate the application of the proposed resonator for designing miniaturized microwave filters. Moreover, the simulated results are well confirmed by the measured results.

265

Compared to the conventional octagonal patch resonator, the proposed FSR offers 22% size reduction. The simulations reported in this paper are carried out using CST Microwave Studio 2018 [43]. The organization of this paper is as follows: The design and analysis of the proposed FSR in dual-mode BPF are illustrated in Section 2. Section 3 investigates the performance of the filter by parameter optimization, mode-splitting characteristics and surface current distribution. The validation of the experimental results and comparison with other reported works in literature are discussed in Section 4. Finally, a conclusion is made in Section 5.

2. Proposed dual-mode fractal BPF design The proposed structure of the single-mode octagonal shaped resonator is depicted in Fig. 1(a), showing symmetry along the horizontal ðA  A0Þ and vertical ðB  B0Þ planes. To realize a singlemode response, the resonator is fed by a pair of 50 X feed lines coupled parallel along the horizontal or vertical plane. Also, each feed line is connected to a V-shaped coupling arm. The simulated frequency response of single-mode resonator resonating at 3 GHz is shown in Fig. 2, which indicates Chebyshev characteristics. To realize a dual-mode response for the resonator, mode splitting and coupling characteristics of two degenerate modes are very much significant in the design. In order to achieve the dual-mode design, a square perturbation element of size p is introduced at 135° (or 45°) from the input and output feed ports (#1 and #2) of the resonator along the diagonal symmetry plane ðD  D0Þas indicated in Fig. 1(b). Using a square patch perturbation, the two degenerate modes are obtained and they are coupled using orthogonal feed lines connected to V-shaped coupling arms for a dual-mode response. The filter structures are designed on an RT/Duroid 6010 substrate having a permittivity (er) of 10.2, thickness (h) of 1.27 mm and loss tangent (tan d) of 0.0023. The simulated response of dual-mode resonator at 2.95 GHz is shown in Fig. 2, which indicates quasi-elliptic characteristics with a TZ at 2.57 GHz and 3.34 GHz on either side of the passband. The generation of these TZs in dual-mode response depends on the size and location of the perturbation element as well as on the position of feed lines [13]. In Fig. 3(a–d), the iterative configurations of the proposed fractal slotted geometry from 0 to 3 are presented. Fig. 3(a) depicts the octagonal patch resonator of zeroth iteration at 2.95 GHz without any fractal slots. As the order of iteration increases, the size of the fractally slotted islands decreases, and there are variations in center frequency, bandwidth, and TZs. The center frequency is influenced by the space created by the fractal slots and it decreases down the frequency spectrum, as the space increases [29]. In Fig. 3(b), a Greek-cross FSR is introduced by etching horizontal and vertical slots located at the center of the octagonal patch to obtain first iteration. The fractal slots of the 1st iteration create 28% of space in the patch and the center frequency is lowered to 2.456 GHz as opposed to 2.95 GHz of the original patch in the 0th iteration for the fixed side length. It implies that the side length of the patch is reduced by 20% if the operating frequency remains unaltered. The second iteration depicted in Fig. 3(c) is obtained by introducing four Greek-cross fractal slotted islands along the four diagonal corners of the center fractal slot of Fig. 3(b), and a total of 35.8% space is produced. The center frequency shifts from 2.456 GHz to 2.424 GHz related to 2nd iteration patch. The process is repeated again for the third iteration shown in Fig. 3(d), thereby incurring sixteen Greek-cross fractal slotted islands along the diagonal corners of the 2nd iteration fractal slotted islands. The space required for the 3rd iteration patch is about

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Fig. 1. Proposed octagonal shaped resonator: (a) Single-mode and (b) Dual-mode configuration.

The geometry of the dual-mode BPF based on the proposed third order FSR is shown in Fig. 5. The circumference of the octagonal shaped resonator is considered as one guided wavelength approximately and is determined using Eq. (1).

300 mm pffiffiffiffiffiffiffi f 0 ðGHzÞ eeff

kg ¼

ð1Þ

where eeff is the effective permittivity, and kg is the guided wavelength [1]. Here, kg = 52.8 mm at the design frequency, f0 = 2.4 GHz. The side-length of the resonator (L) excluding feed lines is about 18 mm, which is approximately 0.34kg. The sidelength of the octagonal patch (a) is about 6.88 mm, which is equivalent to 0.13kg. The space created by area An of the Greek-cross fractal slotted islands in the octagonal patch can be derived and obtained using Eq. (2). Fig. 2. Simulated frequency response of single-mode and dual-mode resonator.

38% area of the patch, only 2.2% more than the 2nd iteration patch. It is noticed that applying the 3rd iteration FSR, the center frequency is shifted towards the lower frequency (2.408 GHz), indicating a miniaturized property with patch size reduction of about 22% as compared to the octagonal patch resonator of 0th iteration. Above the 3rd iteration, the space produced and change in center frequency by the fractal slotted islands will be little, leading to complexity in fabrication of the resonator design. As can be seen in Fig. 2, the center frequency f0 of the original patch in the 0th iteration is 2.95 GHz. Fig. 4 depicts the simulated frequency response of a dual-mode BPF for iterative fractal configurations from 1 to 3. The simulated results clearly show that when the fractal iteration order increases, the center frequency of the resonator decreases. Due to the space-filling property of fractal structures, the operating frequency in terms of subwavelength resonance is reduced drastically, indicating miniaturization [38]. Table 1 summarizes the simulated results of increasing iteration orders of the fractally slotted patch resonator. As seen in Table 1, the fractal iterations exhibit narrow-band frequency responses with dual TZs as rejection band levels on either side of the passband. Further, it is also clearly observed that the insertion loss decreases from 1.5 dB to 0.396 dB and the reduction in patch size increases from 0 to 22% for the dual-mode BPF using the proposed FSR from zeroth to third iteration, respectively. This reduction in patch size indicates miniaturization and the implementation of the FSR retains all the desired characteristics of the original resonator.

An ¼

 n pffiffiffi 0:56
2a2 ð1 þ 2Þ; 2

nP0

ð2Þ

where the integer n is the order of iteration and a is the side-length of the octagonal patch. The length ln and width wn of the fractal slots in each iteration can be found using Eqs. (3) and (4).

ln ¼

0:56 ð4Þn1

wn ¼


L;

0:22 ð4Þn1


L;

nP1

nP1

ð3Þ

ð4Þ

The physical dimensions of the proposed fractal filter structure depicted in Fig. 5 after optimization are listed in Table 2. Here, the gap between the octagonal patch and V-shaped coupling arm, g is fixed as 0.2 mm. Moreover, the optimum size of square perturbation, p is set as 2.1
2.1 mm2 in this design. The proposed FSR and the resulting dual-mode BPF are simple to design and fabricate as they are formed of conventional PCB materials on a single conductive layer without any vias or changes in the ground plane. 2.1. Analysis of fractal resonator To understand the design of the proposed FSR based dual-mode BPF, the fractal resonator has to be analyzed. The quality factor Q is a key parameter in the design of any resonator as it gives an idea about the insertion loss and selectivity of the filter. In other words, higher the Q, smaller is the loss. Moreover, larger Q value provides narrow bandwidth and a more steep frequency response. The unloaded quality factor QU can be obtained from the transmission characteristics of the proposed resonator and is given by Eq. (5).

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267

Fig. 3. Iterative configurations of dual-mode filter with fractal slotted islands: (a) Iteration 0, (b) Iteration 1, (c) Iteration 2, and (d) Iteration 3.

Fig. 4. Simulated response of the dual-mode fractal filter for different iterations.

QU ¼

QL 1  jS21 j

ð5Þ

where jS21 j corresponds to the magnitude in linear scale of the transmission characteristic (insertion loss) at the resonant frequency [2]. The loaded quality factor of the resonator QL can be calculated using Eq. (6).

QL ¼

f0 Df

ð6Þ

where f0 is the resonant frequency and Df is the 3dB bandwidth of the resonator. Fig. 6 illustrates the unloaded quality factor QU and normalized resonance frequency ratio R (defined by fn/f1) as a function of iteration order of the resonator [27]. Here, fn denotes the resonant frequency of the nth order of iteration and f1 denotes the resonant frequency of 1st iteration order of resonator. From Fig. 6, it is inferred that there is a gradual decrease in the resonance frequency ratio R as the order of iteration increases from 1 to 4 and attains a minimum value at n = 4. Furthermore, it is observed that the unloaded quality factor QU increases as the order of iteration increases and reaches the maximum value at n = 3. The results indicate that high quality factor and miniaturization of the resonator are obtained up to the third order of iteration. When the iteration increases above third order, the decrease in resonant frequency of the resonator is little and QU falls to a minimum value at n = 4. In addition, the increase in QU is also due to the degradation of conductor and radiation losses as the order of iteration increases in the resonator. This degradation is mainly because of the etched fractal slots on the resonators and also due to the fractal structure itself, as reported in [27,28,32]. The parallel resonant circuit inset in Fig. 6 forms a typical equivalent circuit model of a patch resonator. It indicates the resonant mode of the resonator and also has a parallel coupling with the source and load impedance. The variable inductance represents the various current paths for different orders of iteration. The reason for a decrease in resonance frequency is due to the increase in inductance as the etched fractal slots on resonator take longer current paths. Nevertheless, as the iteration order is greater than 3,

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Table 1 Iterative results of fractally slotted patch resonator. Simulation

Original patch (0th iteration)

1st iteration fractally slotted patch

2nd iteration fractally slotted patch

3rd iteration fractally slotted patch

f0 (GHz) 3dB BW (MHz) FBW (%) Insertion loss (dB) Return loss (dB) Lower and Upper TZs (dB) Patch size reduction

2.95 324 10.98 < 1.5 >6 54.80 @ 2.57 GHz, 49.28 @ 3.34 GHz 0%

2.456 168 6.84 < 0.408 > 23.3 52.05 @ 2.19 GHz, 49.47 @ 2.74 GHz 20%

2.424 190 7.84 < 0.395 > 21.2 54.31 @ 2.15 GHz, 52.55 @ 2.70 GHz 21.3%

2.408 162 6.73 < 0.396 > 35.5 53.96 @ 2.17 GHz, 51.46 @ 2.69 GHz 22%

Fig. 5. Layout of the proposed dual-mode filter with fractal slots.

the etched fractal slots on resonator are too small to form a longer inductance path internally. As a result, the decrease in resonant frequency is very little for higher order.

3. Performance evaluation The performance of the proposed BPF in Fig. 5 is investigated in this section based on the effect of the length of V-shaped coupling arms, lc and the gap size g. Fig. 7 shows the simulated frequency response of the filter for four different lengths of the V-shaped coupling arms with a fixed gap size of 0.2 mm. It is evident from Fig. 7 that when the length of the coupling arm is considered equal to the side length of the octagonal patch resonator, i.e., lc = 6.88 mm, the optimal response is obtained for both transmission and reflection characteristics of the dual-mode fractal filter. As the length of the coupling arm decreases, the lower TZs shift towards the lower frequency and the upper TZs move towards the

higher frequency with out-of-band rejection better than 30 dB. This clearly shows that the TZs can be controlled by the coupling arm length. However, it is found that as the coupling arm length reduces, the two degenerate modes are excited and gradually move away from each other exhibiting poor return loss characteristics for the dual-mode filter. Fig. 8 shows the simulated frequency response of the filter for four different gap sizes g between the coupling arms and resonator with a fixed coupling arm length, lc of 6.88 mm. In the reflection characteristics, it is apparent from Fig. 8 that mode splitting occurs as the size of the gap is greater than 0.2 mm and for gap size equal to 0.2 mm, only single mode is excited. As the gap size increases above 0.2 mm, the two degenerate modes are excited and their resonant mode frequencies gradually move away from each other. Here, the simulation results of |S11| in the passband imply that the gap spacing affects the upper mode frequency slightly more than the lower mode frequency, resulting in extended bandwidth. From the simulation results of |S21| of the filter, it is evident that as the gap size increases from 0.2 mm to 0.5 mm, the lower TZs move towards the higher frequency, while the higher TZs move towards the lower frequency. Hence, the location of the TZs depends on the gap size between the I/O coupling arms and resonator. Further, as the gap size increases, the coupling between the resonator and the feed port gradually weakens exhibiting poor frequency response for the simulated filter i.e., increase in gap size above 0.2 mm results in high insertion loss and poor return loss characteristics. 3.1. Mode-splitting characteristics With respect to the size of perturbation p, the splitting of mode frequencies and the coupling coefficient between degenerate modes are shown in Fig. 9. To excite and couple the degenerate modes, two V-shaped orthogonal feed ports are utilized as shown in Fig. 5. The coupling coefficient (k) can be computed using the relationship between the two split modes, defined from [1] and is given by Eq. (7).

k¼

2

2

2

2

f 01  f 02

ð7Þ

f 01 þ f 02

where f01 and f02 indicate the resonance frequencies of the mode-I and mode-II, respectively. The coupling between the modes can be varied by varying the size of the perturbation element.

Table 2 Physical dimensions of the proposed fractal filter. Order of iteration (n)

Fractal slots Length, ln (mm)

Width, wn (mm)

1 2 3

10 2.6 0.4

4 1 0.2

Gap between 1st and 2nd iteration slots, g1 (mm)

Gap between 2nd and 3rd iteration slots, g2 (mm)

0.2

0.2

Coupling arms

Feed line

Length, lc (mm)

Width, wc mm)

Gap, gc (mm)

Length, lf (mm)

Width, wf (mm)

6.88

0.707

0.282

3.3

1

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269

Fig. 6. Normalized resonance frequency ratio R and unloaded Quality factor QU as a function of iteration order, n of the resonator. Fig. 9. Simulated dual-mode resonant frequencies and coupling coefficient against perturbation size, p.

Fig. 7. Simulated response of the filter for varying lengths lc of the coupling arms with fixed gap size g.

Fig. 8. Simulated response of the filter for varying gap size g between the coupling arms and resonator with fixed length lc.

From Fig. 9, it is evident that f01 = f02 = 2.417 GHz and k = 0 when p < 2.1 mm. Since the coupling coefficient is zero for perturbation size less than 2.1 mm, there is excitation of single mode

only. As the size of perturbation increases, it is noticed that the mode is split into two degenerate modes. The mode splitting is attributed to the symmetrical distribution of square perturbation element along the diagonal plane. Moreover, the coupling coefficient also increases with increase in perturbation size. When no perturbation is added i.e., p ¼ 0, only single mode is excited and no mode splitting is obtained. For port 1 excitation in patch resonator, the electric field pattern in Fig. 10(a) shows that the excited resonant mode corresponds to TMz100 mode (where z is normal to the ground plane). When the excitation is changed to port 2, the field pattern is rotated by 90° for the associated mode corresponding to TMz010 . With p–0, these degenerate modes are no longer orthogonal as shown in Fig. 10(b). Irrespective of excitation in port 1 or 2, both these modes are excited and coupled to each other, which causes the split in the resonance frequency. Here, the degree of coupling modes is decided by the perturbation size, which in turn controls the mode splitting in the resonator. Moreover, the two coupled modes act like two coupled resonators and thus, a dual-mode BPF is obtained. Hence, to form a dual-mode filter, a small square perturbation of size p = 2.1 mm is added to the diagonal corner of the octagonal patch resonator for exciting and coupling both modes-I and II, respectively. The simulated electric field pattern of mode-I is shown in Fig. 10(b) with higher resonance frequency f01 after mode splitting for p = 2.1 mm. As can be seen, the location of poles and zeros are observed along the opposite diagonals, respectively. The field pattern of mode-II is rotated by 90° from that of mode-I and has lower resonance frequency f02 after mode splitting. As there is no coupling for p < 2.1 mm, the effect of perturbation size on insertion loss and return loss characteristics of the dualmode filter for values of p  2.1 mm are investigated in Fig. 11. From the partial enlarged view inset in Fig. 11(a), it is inferred that when the perturbation size increases above 2.1 mm, two degenerate modes (mode-I and mode-II) are obtained, resulting in two peaks on the insertion loss characteristics. Here, the increase in perturbation size has little effect on modeI frequency (higher resonance) whereas the mode-II frequency (lower resonance) decreases towards the lower side of the passband. The same effect is also experienced in Fig. 11(b) showing two poles on the return loss characteristics for p  2.1 mm. Moreover, increase in perturbation size contributes to increase in bandwidth. Table 3 shows the influence of perturbation size on the return loss, insertion loss, center frequency, bandwidth, and mode resonances. In addition, the two TZs changes its location on either side of the passband as the perturbation size is tuned from 2.1 mm to

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Fig. 10. Electric field pattern of patch resonator (a) Single mode (p = 0) and (b) Dual-mode (p – 0).

Fig. 11. Influence of perturbation size, p on (a) insertion loss and (b) return loss characteristics.

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2.9 mm. From Table 3, a more satisfactory frequency response is observed for a perturbation size of 2.1 mm on the insertion loss and return loss characteristics. 3.2. Surface current distributions Fig. 12 shows the simulated surface current distributions of the proposed dual-mode fractal BPF. With the electrical path shown in Fig. 12(a) and (b), the two TZs at 2.17 GHz and 2.69 GHz are represented with high attenuation in the output port #2. At these frequencies of TZs on either side of the passband, there is no current flow in the output port #2 and all the signals are reflected back to the input port #1. Thus, there is no signal coupling between the I/O ports leading to stop-band conditions. In addition, from Fig. 12(c) and (d), it is evident that there is current flow along the edges of the octagonal patch, central fractal slot and its four diagonal fractal slotted islands. Here, for the resonant mode at 2.38 GHz, the current distribution is along the edges of the

fractal slotted islands i.e., 1 and 3 and for the other resonant mode at 2.43 GHz, it is along 2 and 4. Both the modes exhibit identical surface current distributions along the edges owing to the dualmode resonances with 90° phase shift. In the same way, the sixteen small fractal islands in the diagonal corners will also have current distribution along their edges. At these resonant frequencies in the passband, the input signals couple almost completely between the input port #1 and the output port #2.

4. Experimental results and discussion The miniaturized dual-mode filter so designed is fabricated using the optimal design parameters shown in Fig. 5. The prototype photograph of the fabricated filter with third order iteration is depicted in Fig. 13. The circuit size of the resonator including the feed lines is 22
22 mm2 which is equivalent to 0.416kg
0.416kg, where kg is the guided wavelength at 2.4 GHz. The measurement

Table 3 Simulation parameters of fractally slotted patch resonator against perturbation size, p. Simulation parameters

p = 2.1 mm

p = 2.3 mm

p = 2.5 mm

p = 2.7 mm

p = 2.9 mm

Insertion loss (dB) Return loss (dB) Center frequency (GHz) Bandwidth (MHz) Mode-I and II resonances (GHz)

< 0.396 > 35.5 2.408 162 2.408, 2.408 53.96 @ 2.17 GHz 51.46 @ 2.69 GHz

< 0.481 > 14.57 2.392 192 2.432, 2.360 50.01 @2.12 GHz 50.36 @ 2.71 GHz

< 1.108 > 7.64 2.384 248 2.448, 2.320 55.62 @ 2.08 GHz 52.37 @ 2.72 GHz

< 1.176 > 7.30 2.368 270 2.440, 2.296 52.80 @ 2.05 GHz 52.69 @ 2.74 GHz

< 2.172 > 4.47 2.360 316 2.456, 2.264 51.33 @ 2.01 GHz 53.56 @ 2.74 GHz

Lower TZ (dB) Upper TZ (dB)

Fig. 12. Surface current distributions: TZs at (a) 2.17 GHz and (b) 2.69 GHz and dual-mode resonances at (c) 2.38 GHz and (d) 2.43 GHz (‘The arrow indicates current flow’).

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Fig. 13. Photograph of the fabricated filter.

Fig. 14. Simulated and measured response of dual-mode filter.

and simulation results of the proposed filter are illustrated in Fig. 14. In the passband range of 2.348–2.468 GHz, the measured minimum insertion loss including the connector loss is less than 0.91 dB at the center frequency of 2.4 GHz with a 3-dB FBW of 5%. The experiment is carried out using Keysight N9917A microwave analyzer. Moreover, the measured return loss is greater than 23 dB in the passband and the out-of-band rejection is better than 24 dB in the stopband, respectively. From Fig. 14, it can also be seen that the passband has a pair of TZs obtained by the square perturbation element at the diagonal corner of the octagonal patch. The measurement demonstrates TZs with attenuation levels of 50.04 dB and 46.12 dB at 2.19 GHz and 2.66 GHz, respectively. These two TZs validate high frequency selectivity on either side of the passband leading to quasi-elliptic response for the proposed dual-mode fractal filter. Fig. 15 illustrates the group delay results of the dual-mode filter. The measured group delay is less than 4.8 ns in the passband. Meanwhile, the group delay variation between the simulated and experimental results is between 1.3 ns and 0.9 ns over the frequency range of 2.35–2.51 GHz. The differences between experimental and simulation results of Figs. 14 and 15 may be due to the soldering error at the feed ports and the cable losses during measurement. Table 4 illustrates a performance comparison of the proposed dual-mode BPF with the previous works. The filter presented using the fractal slots on an octagonal patch resonator has smaller electrical dimensions than those reported in [7,8,16]. Though the proposed FSR based filter at 2.4 GHz exhibits slightly larger size, it provides low insertion loss and better return loss characteristics as compared with the SIR based filter realized in [10] at 2.52 GHz, while both are being implemented on the same substrate. Also, the microstrip resonator-based filter implemented on a different substrate in [24] presents high-loss characteristics at 1.6 GHz despite being smaller in size than the proposed filter. Compared with the structures in [7–9,12,14–16,19,23,34], the proposed dual-mode BPF performs favorably well in terms of insertion loss and return loss. In addition, the presented fractal filter also offers narrow bandwidth with better out-of-band rejection

Fig. 15. Simulated and measured group delay of dual-mode filter.

Table 4 Performance comparison with those reported in literature. Ref.

[7] [8] Filter-B [10] Single-band Design-II [16] [23] L-BPF design [24] This work NA – Not Applicable.

Substrate: er, thickness (mm)

Center frequency, f0 (GHz)

3dB BW (GHz)

2.65, 1.0 4.3, 1.58 10.2, 1.27 2.65, 1.0 2.2, 0.508 3, 1.02 10.2, 1.27

2.6 2.4 2.52 3.2 1.6 1.6 2.4

0.364 0.914 0.121 0.659 0.112 0.160 0.120

FBW (%)

14.0 38.1 4.8 20.6 7.0 10.0 5.0

Insertion loss (dB)

< < < < < < <

1.4 1.2 1.87 2.2 2.35 2.6 0.91

Return loss (dB)

> > > > > > >

12 13 20 12.5 18 10 23

Size of filter (mm2)

(kg 2)

12.5
41.6 39.2
22.2 15.8
14.5 70
40 NA 40
45 22
22

0.146
0.486 0.57
0.32 0.343
0.315 1.06
0.61 NA 0.3
0.34 0.416
0.416

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than those filter designs in [7,8,16,24,29]. Due to the significant features of low-profile nature, miniaturization, low-loss, high selectivity, easy assembly, and fabrication without vias in PCB, the resonator proposed in this paper actually exhibits a good potential to function as a high-performance dual-mode filter. Among the popular wireless bands such as Wi-Fi, Bluetooth, and 4G-LTE deploying 2.4 GHz ISM band, the Wi-Fi spectrum (2400– 2483 MHz) is an extremely congested one. To reject interference from adjacent bands, sharp out-of-band suppression is highly required for the narrow 2.4 GHz Wi-Fi spectrum. Hence, the proposed narrow band fractal filter is suitably applicable to 2.4 GHz Wireless LAN systems due to its compactness, low-loss characteristics, and good out-of-band rejection.

5. Conclusion A novel miniaturized microstrip BPF using fractal slots on an octagonal shaped patch centered at 2.4 GHz is designed and realized. The footprint of the proposed filter is 22
22
1.27 mm3. The dual-mode response is obtained by varying the size of square perturbation at the diagonal corner of the patch. In this design, the measured minimum insertion loss is less than 0.91 dB, while the measured return loss is better than 23 dB with a 3 dB-FBW of 5%. Good frequency selectivity is attained owing to two inherent TZs of the dual-mode octagonal patch resonator. As compared with the octagonal patch resonator, the proposed FSR based BPF design provides 22% size reduction indicating miniaturization by the use of fractal slots. Further, the mode splitting characteristics also have been discussed. The simulation and measurement results demonstrate that the dual-mode fractal filter offers good performance in terms of low insertion loss, better return loss, high skirt selectivity, deep out-of-band suppression, and ease of design. The compactness in circuit size, narrow band, and favorable filter characteristics makes the proposed design suitable for applications in Wireless LAN systems covering the 2.4 GHz range for the channels of IEEE 802.11b/g/n. Declaration of Competing Interest The authors declared that there is no conflict of interest. References [1] Hong J-S. Microstrip filters for RF/Microwave applications. 2nd ed. New Jersy, NJ: John Wiley & Sons, Inc.; 2011. [2] Jarry P, Beneat J. Design and realizations of miniaturized fractal RF and Microwave filters. New Jersy, NJ: John Wiley & Sons, Inc.; 2009. [3] Zhu L, Wecowski PM, Wu K. New planar dual-mode filter using cross-slotted patch resonator for simultaneous size and loss reduction. IEEE Trans Microw Theory Tech 1990;47(5):650–4. [4] Wu S, Weng MH, Jhong SB, Lee MS. A novel crossed slotted patch dual-mode bandpass filter with two transmission zeros. Microwave Opt Tech Lett 2008;50 (3):741–4. [5] Zhu L, Tan BC, Quek SJ. Miniaturized dual-mode bandpass filter using inductively loaded cross-slotted patch resonator. IEEE Microw Wireless Compon Lett 2005;15(1):22–4. [6] Sung Y. Dual-mode dual-band filter with band notch structures. IEEE Microw Wireless Compon Lett 2010;20(2):73–5. [7] Wu Y, Hu B, Nan L, Liu Y. Compact high-selectivity bandpass filter using a novel uniform coupled-line dual-mode resonator. Microwave Opt Tech Lett 2015;57 (10):2355–8. [8] Velidi VK, Sanyal S. Sharp-rejection microstrip bandpass filters with multiple transmission zeros. AEU-Int J Electron Commun 2010;64(12):1173–7. [9] Chen CJ. Design of parallel-coupled dual-mode resonator bandpass filters. IEEE Trans Compon Packag Manufact Tech 2016;6(10):1542–8. [10] Xiao J-K, Li Y, Zhu M, Zhao W, Tian L, Zhu WJ. High selective bandpass filter with mixed electromagnetic coupling and source–load coupling. AEU-Int J Electron Commun 2015;69(4):753–8. [11] Hung CY, Weng MH, Jhong SB, Wu S, Lee MS. Design of the wideband dualmode bandpass filter using stepped impedance resonators. Microwave Opt Tech Lett 2008;50(4):1104–7.

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S. Karthie is working as Assistant Professor in the Department of Electronics and Communication Engineering, SSN College of Engineering, Chennai and has over 15 years of teaching experience. Currently, he is pursuing research towards his Ph.D. in the area of microstrip filter design from Anna University, Chennai. He has published research papers in National and International journals and conferences. He has authored a book on Electromagnetic Field Theory. His research interests include electromagnetics, microstrip filter design, and RF/microwave integrated circuits. He is an active member of the IEEE MTT Society. He is also a Life member of both IETE and ISTE (India).

S. Salivahanan is the Principal of SSN College of Engineering, Chennai. He has four decades of teaching, research, administration and industrial experience, both in India and abroad. He has industrial experience as scientist/engineer at Space Applications Centre, ISRO, Ahmedabad, India, Telecommunication Engineer at State Organization of Electricity, Iraq, and Electronics Engineer at Electric Dar Establishment, Kingdom of Saudi Arabia. He is the author of 48 popular books and he has published several research papers at national and international levels. His research interests include electromagnetics, RF/microwave integrated circuits, and VLSI design. He is a Senior Member of IEEE, Fellow of IETE, Fellow of Institution of Engineers (India), Life Member of ISTE and Life Member of Society for EMC Engineers. He is also a member of IEEE societies in Microwave Theory and Techniques, Communications, Signal Processing, and Aerospace and Electronics.