Fuel cell power conditioning system design for residential application

international journal of hydrogen energy 34 (2009) 2340–2349

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Fuel cell power conditioning system design for residential application Yong Wanga,*, Seeyoung Choib, Eunchul Leec a

Energy Lab, Samsung Advanced Institute of Technology, Mt. 14-1, Nongseo, Giheung, Yongin, Republic of Korea Digital Appliance Division, Samsung Electronics CO., LTD, Maetan-3Dong, Suwon, Gyeonggi, Republic of Korea c Willings CO., LTD, SK Ventium, 522 Dangjung-Dong, Gunpo-Si, Gyeonggi-Do, Republic of Korea b

article info

abstract

Article history:

This paper presents a design of a high performance proton exchange membrane fuel cell

Received 30 May 2008

(PEMFC) power conditioning system (PCS) for residential application. Firstly, a high effi-

Received in revised form

ciency PCS topology is described which can improve the PCS maximum efficiency up to

21 December 2008

92.9%. Furthermore, a novel PCS controller is presented, which succeeds in suppressing the

Accepted 23 December 2008

low frequency current ripple, controlling the dc link voltage and inverter output current.

Available online 6 February 2009

The controller also achieves reliable power grid integration. The experimental results show that a residential fuel cell PCS with high performance can be achieved.

Keywords:

ª 2009 International Association for Hydrogen Energy. Published by Elsevier Ltd. All rights

Efficiency

reserved.

Fuel cell Grid integration Power conditioning system Ripple current

1.

Introduction

High efficiency, low ripple current, inverter current control and power grid integration are the major requirements for the residential fuel cell power conditioning system (PCS) design [1– 3]. The PCS usually includes two-stage converters of a dc–dc converter and a power grid connection dc–ac inverter as Fig. 1 shows [4], where Ifc, Vfc are the fuel cell input current and voltage, Idc, Vdc are the dc link current and voltage, IR, VR are the inverter output current and voltage, respectively. In this paper, firstly an efficient PCS topology is presented. Furthermore, this paper presents a novel PCS controller applied to the new topology. The controller includes a single loop PI controller of input current to track the ripple current directly and a d–q rotating coordination controller to achieve the dc link voltage, the inverter current control and the power grid integration.

2.

High efficiency PCS topology design

Efficiency of the two-stage PCS mainly lies in high transmission ratio dc–dc converter. Many dc–dc converters have been proposed and investigated during the last decade [5,6]. For fuel cell applications, a voltage-fed full bridge (FB), operating with a high transmission ratio transformer and a diode rectifier has been proposed [7,8]. It is believed that the FB topology is suitable for medium power used in green power applications. The power losses in a FB converter are mainly from switching devices, transformer, rectifier diode, and snubber circuit. The losses from switching devices can be divided into conduction losses and switching losses. The phase shifting modulation full bridge (PSFB) converter is proposed to reduce the switching loss in FB converter by soft switching [9].

* Corresponding author. Danfoss Solar Inverters, Jyllandsgade 28, 6400 Soenderborg, Denmark. Tel.: þ45 5035 6666; fax: þ45 7488 1301. E-mail address: [email protected] (Y. Wang). 0360-3199/$ – see front matter ª 2009 International Association for Hydrogen Energy. Published by Elsevier Ltd. All rights reserved. doi:10.1016/j.ijhydene.2008.12.090

international journal of hydrogen energy 34 (2009) 2340–2349

DC-DC converter Ifc + Fuel Cell Vfc

DC-AC Inverter

DC link Idc +

IR VR

Vdc

Grid

-

-

Fig. 1 – Block diagram of PEMFC PCS in residential application.

A leakage inductor must be inserted into the main circuit to maintain the soft switching in PSFB. As a result of this, we will observe duty loss of switching devices combined with relatively large circulating current flowing through switching devices. Further, we will see that the conduction losses of the switching devices and transformer are still high compared with FB converter [10], especially in the low input voltage and high input current fuel cell PCS. Based on the above realization, this paper proposes a novel, efficient, parallel–series full bridge (P–SFB) converter as shown in Fig. 2. The main power stage consists of two traditional FB converter connected in parallel at the dc input side and in series at the rectifier output. As shown in Fig. 2(b), phase shifting modulation is used to regulate the output voltage. Each bridge operates at the same switching frequency and leg-to-leg phase shift to process half of the output power apiece. Furthermore, it adds two capacitors paralleled with the leading

leg’s two MOSFETs, respectively to reduce the switching loss. The uniqueness of the proposed topology lies in that the phase shifting modulation is used without the insertion of leakage inductor. Where in Fig. 2, Ifc1 and Ifc2 are sub-converter 1 and 2 input current, Irms is sub-converter primary side current, Vdc1 is sub-converter 1 dc link voltage, Vdc2 is sub-converter 2 dc link voltage, n is transformer turns ratio, Vrms is transformer primary side voltage,Vrms2 is transformer secondary side voltage, Lf is output filter inductance, Rsn is snubber resistance, and Io is sub-converter 1 dc link current. As the MOSFET switching loss is concerned, the leading leg’s turn on loss is zero because Lf makes it easy to achieve ZVS on, and the turn off loss is reduced by the shunt capacitor, which will be experimentally analyzed later. As for the lagging leg, the ZVS off is easy to be reached according to the PSFB principal. However, the lagging leg still operates with hard switching conditions when MOSFET turns on. No duty loss can be observed since there is no leakage inductor inserted in P–SFB. Hence, the transformer’s turn ratio n in each sub-converter is smaller than half of that in PSFB. Therefore the peak current and voltage, in the secondary side as well as the circulating current in primary side, are reduced to less than half of that in PSFB. The rectifier diode’s rated voltage could be selected as 600 V, not 1200 V as in PSFB converter. So the losses of the transformer, rectifier diode and snubber are reduced. Based on the above, the novel two-stage efficient PCS for fuel cell residential application is designed as Fig. 3.

3.

a

Ifc

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A novel PCS controller design

Ifc1 Vrms2

S2

S1

Cdso

Vfc

Irms S3

Lf

Io

Vrms 1:n Vdc1

S4 Vdc

Ifc2 S5

S6

S7

S8

Vdc2

b S1, S5 S2, S6 S3, S7

This paper presents a novel PCS controller applying on the above new topology. The controller diagram is shown in Fig. 4. The single input current control loop can significantly suppress the ripple current. Meanwhile, a d–q rotating coordination controller succeeds in controlling the dc link voltage and the inverter output current. In the d–q controller, IR is transformed into dc component. Meanwhile, the dc link voltage PI compensator’s output IRd* is taken as the d axis current reference. In this way, the dc link voltage is kept constant. The model analysis later will show that the input current instruction Ifc* controls the inverter output current through the dc link voltage control also. On the other hand, the reactive current (reactive power) disturbance variable IRq* (DQ) on q axis supplies an easy way for PCS to be reliable integrated with the power grid, which will be detailed in Section 3.3.

3.1.

Ripple suppressing

S4, S8

V rms’’

Irms’’

Fig. 2 – P–SFB topology and phase shifting modulation. (a) topology (b) phase shifting modulation.

The low frequency current ripple is believed to be generated and propagated from the inverter output as shown in Fig. 5. With a linear load, the inverter output inductor current is sinusoidal as the output voltage, while the input voltage and current are dc component. However, the input current Idc contains 120 Hz current ripple because of the rectification effect of the IGBT’s shunt diode bridge. Also the current ripple continually propagates through the dc–dc converter and back to the fuel cell.

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international journal of hydrogen energy 34 (2009) 2340–2349

P-SFB

H-Bridge Inverter

Fuel cell

Grid

Fig. 3 – Novel two-stage PCS block diagram.

With a typical 1.2 kW PCS design, measurements reveal a peak-to-peak input current ripple of 30%. Unlike the other PCS, the current ripple causes not only the reduction of the PCS’s output capability, but also the consumption of fuel. If the ripple rate is 30%, for example, it implies that 15% more fuel is consumed. Further, it is believed that the current ripple is harmful to the fuel cell’s lifespan [11]. A fuel cell dc–dc converter’s controller is presented in [12] to suppress the low frequency ripple. This is a double loop controller, which consists of the outside control loop of dc link voltage and the inside control loop of the output inductor current. The double loop controller suppresses the ripple current compared with the conventional single control loop of dc link voltage. The PCS controller in Fig. 4 consists of the input current control, the dc link voltage and the inverter output current control, together with the power grid integration control. Unlike [12], the PCS controller in this paper configures a single control loop of the dc–dc converter input current to track the low dc side ripple current directly. Meanwhile, the dc bus voltage is no longer controlled by the dc–dc converter, but

P-SFB

I fc

Idc

+

3.2. The dc link voltage and the inverter current control based on d–q coordination The dc link voltage and the inverter current are controlled based on the d–q rotating frame coordination. This new control method transforms inverter current from the stationary frame ac variable to d–q rotating frame dc variable. In the rotating frame, the dc link voltage is controlled constant by adding the control variable IRd* on d axis. Also, the single phase inverter current control becomes easy and equivalent to that of three phase inverters, so that an infinite loop gain at the fundamental frequency of the system and fast dynamic response could be achieved. To explain the controller better,

H-bridge inverter

DC link +

grid integration

IR

Vdc

V fc

Fuel Cell

through the inverter d–q controller. It is expected to have better effect than [12], as it is not the dc link current, but the converter input current which is tracked directly. Compared with the double loop controller, it is more advantageous to design a steady control loop with high gain, high bandwidth to suppress the 120 Hz ripple current for the single input current control loop.

{

VR

Gr id

-

-

DR

-

+

PWM scale

IRout

+

ripple mitigation

* IRd

IRq*( Q) IRq-PIout

IRd-PIout +

Ifc*

IR

dq RI

350V

Ifc

frequency detection

Vg

current compensation IRd

dc link control

reactive power disturbance

IR q

RI dq

1

II

inverter control

control circuit Fig. 4 – Block diagram of PCS controller.

90°delay

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international journal of hydrogen energy 34 (2009) 2340–2349

120Hz

60Hz

Idc


IRdPIout ¼ IRd  IRd C2 ðSÞ

(5)

  IRqPIout ¼ IRq  IRq C3 ðSÞ

(6)

where: IR Vdc

C

C2 ðSÞ ¼ kp2 þ

ki2 S

(7)

C3 ðSÞ ¼ kp3 þ

ki3 S

(8)

VR

kp2, kp3 are proportional coefficients, and ki2, ki3 are integration coefficients of PI controller in d–q coordinate. If applying (3), (4) to (5), (6), we get:

Fig. 5 – Low frequency current ripple generated by H-bridge inverter.


IRdPIout ¼ IR cos ut þ II sin ut  IRd C2 ðSÞ the PCS controller block diagram in Fig. 4 is modeled as shown in Fig. 6. In Fig. 6, Gvi1, Gvi2 are the voltage to current transfer function of dc–dc sub-converter, Gid1, Gid2 are the current to control transfer function of each sub-converter. Given Ifc1 ¼ Ifc2 ¼ Ifc =2, Vdc1 ¼ Vdc2 ¼ Vdc =2, then: Vdc Vdc1 þ Vdc2 Gid1 $Gvi1 þ Gid2 $Gvi2 ¼ ¼ Ifc Ifc Gid1 þ Gid2

IRd ¼

(2)

where C1 ðSÞ ¼ kp1 þ ki1 =S, kp1, ki1 are proportional and integration coefficients respectively. Vdc reaches around 350 V with the stimulation of Ifc1 and Ifc2, which enables the inverter connection to the power grid. The inverter output current IR is sampled back and delayed 90 by an all pass filter to generate an imaginary current II. IR and II are transferred into d–q coordinate variables IRd, IRq by d–q transformation as: IRd ¼ IR cos ut þ II sin ut

(3)

IRq ¼ IR sin ut þ II cos ut

(4)

With IRd, IRq, PI compensator output variables in d–q rotating coordinate IRdPIout ; IRqPIout can be expressed as:

D

Ifc +-

  IR sin ut þ II cos ut  IRq C3 ðSÞ


2 IRout ¼ IR cos2 utþII sinutcosutIRd cosut C2 ðSÞþ IR sin ut  II cosutsinut C3 ðSÞ

(11)

IR ¼ I cosðut þ fÞ

(12)

II ¼ I sinðut þ fÞ

(13)

where I is the amplitude of inverter output current, u is the fundamental frequency of inverter output current in rad/s, 4 is the initial phase of inverter output current. If Eq. (2) is applied to Eq. (11), it can be concluded that the input current instruction Ifc controls the inverter output current through dc link voltage control.

3.3.

Power grid integration means that PCS connects to the grid at normal grid-conditions, and disconnects again in faulty conditions. The controller in Fig. 4 demonstrates an easy way for grid integration. When inverter is connected to the grid, the active and reactive power of the inverter, the load and the grid reach

ifc1

Gid1(s)

ifc2

Gid2(s) *

I Rd

RI

Power grid integration

Gvi1(s)

Vdc1

350V

Vdc + D

-

Gvi2(s)

PI

Vdc2 I Rd-PIout

I Rd +

dq

PI

RI

I Rq 0

90 delay

+-

I Rout

dq

II

1

(10)

And applying (9), (10) to d–q reverse transformation in Fig. 6, the inverter output modulation current becomes as:

D

PI

IR



(1)

  Gid1 $Gvi1 þ Gid2 $Gvi2 $Ifc  350 C1 ðSÞ Gid1 þ Gid2

Ifc*

IRqPIout ¼

(9)

PI I Rq-PIout

* ( Q) I Rq

Fig. 6 – Model of the novel PCS controller.

++

Vg

PWM scale

Inverter PWM

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international journal of hydrogen energy 34 (2009) 2340–2349

PFC+jQFC PG+jQG FC

P-SFB

Inverter

Grid

A PL+jQL

Passive load

Fig. 7 – Power balance at the common point.

balance at common point A, as Fig. 7 and Eqs. (14) and (15) show. PL ¼ PFC þ PG QL ¼ Q FC þ QG ¼ VL2

(14) 

1  uC uL

 (15)

where PFC, Q FC: the active and reactive power of PCS, PL, Q L: the active and reactive power of the load, PG, QG: the active and reactive power of the grid,

Fig. 8 – 1.2 kW PEMFC PCS prototype.

Fig. 9 – Leading leg MOSFET switching signal (without shunt capacitor). Vds: MOSFET drain–source voltage, Vg: MOSFET gate voltage.

international journal of hydrogen energy 34 (2009) 2340–2349

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Fig. 10 – Leading leg MOSFET switching loss (without shunt capacitor). Vds: MOSFET drain–source voltage, Id: MOSFET drain– source current.

VL, u: the common point voltage and frequency, respectively, R, L, C: the load resistance, inductance and capacitance, respectively. Considering the worst situation, before grid fault happens, the reactive power of PCS and the load are equal, which means,

QG ¼ 0 and QL ¼ QFC. While in the d–q controller, a periodical dc component IRq is added on q axis, as Fig. 4 shows. It means the reactive power disturbance DQ added at A. After grid fault, the reactive power balance at A will be broken because of DQ. Therefore, the output voltage frequency is forced to change, so reactive power of A can reach the new balance according to Eq.

Fig. 11 – Lagging leg MOSFET switching signal. Vds: MOSFET drain–source voltage, Vg: MOSFET gate voltage.

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international journal of hydrogen energy 34 (2009) 2340–2349

Fig. 12 – Lagging leg MOSFET switching loss. Vds: MOSFET drain–source voltage, Id: MOSFET drain–source current.

(15). This frequency change can be detected to confirm grid fault. Therefore, PCS is disconnected from the grid afterwards.

4.

Experimental results

A 1.2 kW PEMFC PCS prototype, as shown in Fig. 8, has been developed to testify the proposed PCS topology and the

controller. The switching frequency of P–SFB is 30 kHz, while the inverter’s switching frequency is 15 kHz. The rated fuel cell voltage Vfc is 40 V, while the operational voltage range is 30–70 V. The switching waveforms and switching losses in P–SFB are shown in Figs. 9–12. Figs. 9 and 10 show that, the turn on loss is zero for the leading leg MOSFET since Lf makes ZVS turn on achieved. The

Fig. 13 – Leading leg MOSFET turn off loss reduced by Cdso [ 40 nF. Vds: MOSFET drain–source voltage, Id: MOSFET drain– source current.

international journal of hydrogen energy 34 (2009) 2340–2349

Fig. 14 – P–SFB efficiency.

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turn off loss is as large as 40 mJ per period when there is no shunt capacitor implemented. Figs. 11 and 12 show that, the lagging leg MOSFET reaches ZVS turn off, and one MOSFET has only 2 mJ turn on power loss per period. Fig. 13 shows the leading leg’s turn off power loss is reduced due to Cdso paralleled. 50% of reduction has been accomplished by optimizing the Cdso as 40 nF. Fig. 14 shows 95.5% maximum efficiency of the P–SFB converter around 1200 W. Hence, the PCS reaches 92.9% efficiency.

Fig. 15 – Ripple current propagation to fuel cell with conventional controller.

Fig. 16 – Fuel cell ripple current with proposed PCS controller.

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international journal of hydrogen energy 34 (2009) 2340–2349

Fig. 17 – Grid connected inverter input dc link voltage & current, output voltage & current.

Fig. 18 – The inverter disconnection from the grid at the grid faulty conditions.

Figs. 15 and 16 are the waveforms of the input current ripple before and after the PCS controller is used. Fig. 15 shows Ifc has about 30% ripple rate with normal dc link voltage controller. While in Fig. 16, the input ripple current rate is only 3% with the PCS controller proposed in this paper. Fig. 17 shows the waveforms of the grid connected inverter dc link voltage & current, and output voltage & current. Fig. 18 shows the waveform which shows the inverter disconnects from the grid when the grid faulty conditions happen.

5.

Conclusions

Experimental results verified that the maximum total efficiency of the novel PCS is 92.9%. The novel controller reduces the PCS low frequency input current ripple to about 3%. Also,

the controller controls dc link voltage as well as the inverter output current with improved performance. Finally, the controller has been proven to enable a reliable PCS grid integration. Therefore, these theoretical derivations and experimental results verify that these approaches are highly effective to reach an efficient and reliable residential fuel cell PCS.

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