Gated charge sensitive preamplifier

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(I969) 328-332;

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NORTH-HOLLAND

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GATED CHARGE SENSITIVE PREAMPLIFIER *

L. V. EAST University of California, Los Alamos Scientific Laboratory, Los Alamos, New Mexico 87544, U.S.A.

Received 20 February 1969 A charge sensitive preamplifier incorporating an internal linear gate is described. The preamplifier was developed for use with proportional counters subjected to accelerator beam pulses. Many features are incorporated that make it generally useful for

other applications requiring a stable, moderately low-noise gatable preamplifier. Detector arrays with total input capacitance of greater than 10a pF may be used with minimal loss in preamplifier charge sensitivity.

1. Introduction

in the input stage. The noise sources present in the input stage are: 1. F E T channel resistance noise (Johnson noise); 2. Channel 1/f noise (equivalent to "flicker noise" in vacuum tubes); 3. Input current noise due to FET gate (and detector) leakage current; 4. Parallel resistance noise due to the effective input resistance (parallel combination of feedback resistor and detector bias resistor). If the preamplifier is followed by a simple pulseshaping network consisting of a single R C integrator and R C differentiator having equal time constants, z, then it has been shown that the effective mean square charge at the preamplifier input due to the above noise sources are given (to within a small constant factor) by the following expressions2):

A charge sensitive, or integrating, preamplifier incorporating a linear gate is described in this paper. The internal linear gate was designed to allow the preamplifier to be turned off during an accelerator beam burst in order to reduce recovery problems in successive amplifier stages. Previous methods 1) of off-gating preamplifiers tend to introduce large gating transients that produce recovery problems in the preamplifier or successive amplifiers. In the present design, transients are held to a minimum, and complete recovery is obtained within one amplifier pulse width following the gate pulse. The preamplifier was developed for use with large proportional counter arrays; however, the circuit incorporates many features that make it useful with ionization chambers, solid state detectors (particularly large-area surface barrier detectors), and scintillation detectors. High open-loop gain in the input stage and the use of a high transconductance field-effect transistor (FET) as the input element reduce the effects of input capacitance on the preamplifier charge sensitivity and noise contribution. The charge sensitivity and output pulse decay time may be varied independently over wide ranges by simple component changes to meet the varying requirements of different detectors. A review of preamplifier noise considerations, a discussion of the preamplifier circuit, and test results are presented in the following sections. 2. Preamplifier noise

In discussions on preamplifier noise, it is usually assumed that the noise contributions from the first stage dominate any succeeding noise sources. With moderate care in design, this assumption is generally valid, even when using low-noise field-effect transistors

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* Work performed under the auspices of the U.S. Atomic Energy Commission.

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= absolute temperature, = Boltzmann's constant, = total input capacitance, the FET transconduetance, = flicker noise constant, ~-- electron charge, = total input leakage current (detector plus F E T gate junction), total input shunt resistance.

In present day high-quality FET preamplifiers the dominant noise terms when using solid state detectors

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FET's in parallel to increase the effective 9m. The contribution of shunt resistance noise can also become non-negligible if too small a value of detector load resistor (less than ~ 105 ohm) is used.

and pulsed ionization chambers are the channel noise, eq. (1), and leakage current noise, eq. (3), with flicker noise usually being negligible. The resistance noise can be made small by using large-value resistors ( ~ 108 to 1 0 9 ohm) for feedback and detector loads. Since these noise sources have opposite dependences upon the amplifier time constant, ~, there will be an optimum value of z (usually in the range of 0.5 to 5 #s) that results in minimum total noise.

3. Preamplifier circuit A schematic diagram of the preamplifier is shown in fig. I. The charge sensitive input stage consists of Q~, Qa and Q 3 . A 2N4860 was chosen for Qt because of its very high g+, (typically 20-30 mA/V) in order to reduce the effect of input capacity on the preamplifier noise output. Other high gm FET's, such as a TIS42, 2N4861, or TIS75, could also be used for Qt. Q2 is a grounded base amplifier forming the upper half of a cascade amplifier with Q~. Charge feedback is accomplished by means of the 2.2 pF capacitor, Cr, from the emitter of Q3 to the gate of Qt. The 20M resistor shunting Cf results in a pulse decay time of ~ 50 #s. The output from the emitter follower Q3 is bootstrapped back to the collector load of Q2 in order to increase the open-loop gain. A further increase in the open-loop gain is achieved by applying positive feedback 3'4) to the emitter of Q2 with the resistor Rp.

When using proportional counters with FET or vacuum tube preamplifiers, the spectrum broadening due to electronic noise is usually insignificant compared to the broadening due to the statistical variations in multiplication and charge collection in the counter tube. In this case, z is chosen from considerations of charge collection time in the detector and counting rates rather than noise minimization. However, for large arrays of proportional counters operated in parallel, the capacitance associated with the counters and connecting cables may result in the contribution of channel noise becoming quite large. The contribution from channel resistance noise can be reduced by selecting a FET with a large gin, or by using several

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It is desirable to make the open-loop gain of the charge sensitive amplifier large in order to reduce the dependence of charge sensitivity on input capacity. As may be easily shownS), the output voltage Vo resulting from a charge Q at the input of a charge sensitive amplifier is given by Vo = - ( Q / C f )

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and hence the charge sensitivity is essentially independent of input capacity. With the proper choice of Rv, the open-loop gain, A, can be made to approach infinity. A further reduction in the value of Rp results in the amplifier appearing

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to operate into a negative impedance, and the charge sensitivity will increase rather than decrease with increasing input capacity. It was found that under certain combinations of input capacity and shunt resistance, the amplifier would become unstable when Rp was chosen for near infinite open-loop gain. Therefore, Rv was chosen to give an open-loop gain of approximately 104, resulting in stable operation under all operating conditions and a reduction in charge sensitivity of only about 4% for an input capacitance of 103 pF. The rise time of the charge loop can be improved under conditions of high input capacity by shunting Rp with a small capacitor. However, the use of too much capacitance will result in severe pulse ringing or oscillation. The linear gating action is accomplished by Q12; a positive signal applied to the base of Q~3 causes Q12 to be turned off, preventing signals from being fed to the output amplifier. A potentiometer in the emitter of Q3 is adjusted to eliminate any pedestal when Q12 is switched. In order to reduce the effects of tem-

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perature and counting-rate variations on this adjustment, the dc voltage at the emitter of Q3 is stabilized in a separate dc loop consisting of Q2, Q3, Q9 and Qt0. Qt~ acts as a temperature-compensated zener diode and supplies the dc reference for this loop and the bias on the base of QI4. The temperature coefficient of the on-resistance of Q~2 (approxim a t e l y + l . 7 o h m / ° C ) is compensated by the series 100 ohm thermistor. Recovery problems in the input loop that might occur from large overloading pulses occurring while the preamplifier is turned off (as could result, for example, from pile-up pulses produced during an accelerator beam burst) are reduced by using Q¢ to shunt the feedback capacitor during off-gating with a 20 K resistor. This results in a decay time of ~ 50 ns for any input pulses occurring during this time. It would be desirable to short-out the feedback capacitor during off-gating, thereby making the input loop inoperable; however, the input loop will not remain stable under most input conditions with the feedback shorted. Switching transients resulting from coupling through the gate capacity of Q4 are neutralized by applying an opposite polarity pulse to the input through capacitor Co. The amount of neutralizing charge applied through C c is adjustable by means of a potentiometer in the emitter of Q6. This adjustment will depend upon the preamplifier input capacitance, and to some extent upon the gate pulse width, and therefore should be made under actual operating conditions to minimize the base-line disturbance at the end of the gate pulse. Transistors Q4 and Q5 also serve to protect Q~ from damage by large positive voltage transients appearing at the preamplifier input. This arrangement I

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(a)

(b) Fig. 4. Preamplifier output pulses obtained with the gain set at ×5 and the output terminated in 100ohm. Vertical scale is 200 mV/div., horizontal scale is 5 ps/div. (a) Offgate time of 20 IZS. (b) Gate length increased to 35 ps, with same input pulses as in (a).

has been found to be quite effective in preventing Q~ from being damaged by detector power supply transients. Transistors Q14-Q 17 comprise the output amplifier, which is a feedback stabilized non-inverting amplifier of standard design. The output is linear to > 3.8 V into an open load, or ~ 2 V into a 10092 load, for positive output pulses (negative preamplifier input). The linear range of operation is reduced slightly for negative output pulses. As constructed by the author, output pulse decay times of approximately 2/as, 4 ps, or 50/as can be selected by a jumper on the circuit board. Signal coupling to the output stage is through a pole-zero compensated network for short output pulses. Any output pulse decay time in the range of 0.5 ps to 50/as can be obtained by changing one capacitor and one resistor in the coupling network. A small amount of integration is introduced between the input and output stages by a 470 pF capacitor to ground to reduce the amplitude of gating transients ( ~ 0.1 V or less at maximum gain) from Qtz. This results in an output pulse rise time of approximately 0.2 to 0.3/as. Faster rise times can be obtained by reducing or eliminating this capacitor, with the result that larger gating transients will occur at the preamplifier output.

332

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4. Test results

The preamplifier noise output as a function of input capacity, corrected for charge sensitivity variations with input capacity, is shown in fig. 2. Also shown in the figure for comparison are the relative noise outputs of a bipolar transistor input preamplifier (Model 223QN) and a low-noise F E T preamplifier (Model 580) commonly used at this laboratory. The low-noise FET preamplifier used for these tests also used a hightransductance FET (TIS42) in its input stage. The noise performance of the gated preamplifier is not as good as that of the low-noise preamplifier (due to the gated preamplifier's larger input capacity and smaller input resistance), but markedly better than the performance of the bipolar transistor preamplifier. For z = 2 #s, the equivalent noise line width of the gated preamplifier for a silicon detector is typically 4.2 keV for zero input capacity, 8.0 keV for an input capacity of 150 pF, and ~ 50 keV for an input capacity of 1500 pF. The normalized charge sensitivities of the three types of preamplifiers are shown in fig. 3 as a function of input capacity. The action of the linear gate is illustrated in fig. 4. In fig. 4a a series of output pulses are shown following

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the off-gated time. With the output amplifier gain set at x 5 and operating into al00- ohm load, a 100 mV negative-going transient is produced at the preamplifier output at the beginning of the gate pulse, and a positive transient of less than 50 mV is produced at the end of the gate pulse. As may be seen from the figure, the output pulses reach full amplitude in a time that is less than one amplifier pulse width following the gate pulse. In fig. 4b the same input pulses were applied as for fig. 4a, but the gate length has been increased to inhibit the output. It may be seen that no feedthrough pulses occur during the time that the gate pulse is present. Fig. 5 illustrates the effectiveness of the gated preamplifier when used with a bank of 3He proportional counters located near a pulsed (D, T) neutron generator. A PuLl neutron source was located between the (D, T) target and the detector. The (D, T) source was repetitively pulsed on for 40 ms and off for 80 ms. During the beam pulse, the neutron flux at the detector was increased by a factor of g l0 s. For the lower spectrum in the figure, the preamplifier was gated off during the 40 ms beam pulse. The shape of this spectrum is the same as with the PuLl source alone, except for a small peak in the vicinity of channel 20 produced by the gating transients. The upper spectrum shows the effects of amplifier recovery time from the pile-up pulses that occurred when the preamplifier was not gated off during the beam pulse. The maximum variation in charge sensitivity with temperature was found to be less than 0.02%/°C at 25 °C for input capacities less than 100 pF. For an input capacity of 600 pF, the temperature coefficient is typically - 0 . 0 4 % / ° C with the output amplifier gain set at × 5 and using 4 ps output pulse shaping. The author gratefully acknowledges the many valuable discussions held with R. D. Hiebert, and the very able and patient assistance of M. M. Stephens.

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t) K. Fukuda, S. Okabe and Y. Sato, Nucl. Instr. and Meth. 50 (1967) 150. 2) See, for example, F. S. Goulding, Nucl. Instr. and Meth. 43 (1966) 1. 3) W. W. Goldsworthy, Nucl. Instr. and Meth. 52 (1967) 343. 4) j. H a h n and R. Mayer, I R E Trans. Nucl. Sci. NS-9, no. 4 (1962) 20. 5) See, for example, R. L. Chase, Nuclear pulse spectrometry (McGraw-Hill, New York, 1961) ch. 2.