Research and design of a power management chip for wireless powering capsule endoscopy

Microelectronics Reliability 53 (2013) 129–135

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Research and design of a power management chip for wireless powering capsule endoscopy Jianguang Chen a,c,⇑, Liang Feng b, Yuhua Cheng a,c a

System-On-Chip Laboratory, Institute of Micro-Electronics, Peking University, Beijing, China Parade Technologies Inc., (Shanghai), Xin Yuan Technology Square, 16th Floor, Num 418, Gui Ping Road, Shanghai, China c Shanghai Research Institute of Micro-Electronics, Peking University, Pudong New District, Shanghai, China b

a r t i c l e

i n f o

Article history: Received 21 February 2012 Received in revised form 18 June 2012 Accepted 30 July 2012 Available online 29 September 2012

a b s t r a c t This paper describes a power management chip for wireless powering capsule endoscopy, including overvoltage protection circuit RF limiter, CMOS rectifier and low dropout regulator (LDO). Bulk commutation technique is applied to suppress the substrate leakage current problem of CMOS rectifier. A bandgap circuit used as reference for LDO is optimized carefully and has a more than 50 dB PSR at 2 MHz, which can effectively suppress the ripple of rectifier output used as input voltage of LDO. It incorporates a voltage controlled current source (VCCS) compensation network to make sure the loop’s stability is not dependent on the output capacitor’s Equivalent Series Resistance (ESR). The proposed chip has implemented in 0.35 lm CMOS technology and the measurement results show that it can provide as high as 63.8 mA output current at a 3 V output voltage. The PMC has been used in a wireless powering system designed for capsule endoscope application. Experiment results show that the Power Management Chip (PMC) can deliver as high as 191 mW DC energy to the load, imitating the physical condition in a biomedical environment in a human body. Ó 2012 Published by Elsevier Ltd.

1. Instruction Capsule endoscopes powered by a lithium-ion battery have been used for diagnosis of gastrointestinal diseases for several years. Such treatments require energy as high as several hundreds of mW. A lithium-ion battery has a limited power capacity and cannot provide enough energy to support the whole diagnosis process, which requires a heavy photo-image taking and lasts usually 8 h or longer. Furthermore, the lithium-ion battery has its own potential security risks, which limits further improvement of this power solution. As a real-time energy supply, wireless powering system has been widely studied to replace the lithium-ion battery in a capsule endoscope. Fig. 1 shows the block diagram of a wireless powering system used for endoscope applications. The DC energy is converted to AC energy through a class E amplifier and transmitted into space in the form of electromagnetic wave. A LC series resonator harvests the AC energy, which is rectified by a rectifier and becomes a DC voltage with small ripple. Linear regulator regulates the rectifier output voltage and provides a clean and stable voltage for the loads, which are other IC chips in the capsule endoscopy. An RF limiter block is connected to the receiver anten-

⇑ Corresponding authors. Address: Shanghai Research Institute Of Micro-Electronics, Peking University, Pudong New District, Shanghai, China. Tel.: +86 021 61091006x813; fax: +86 021 61091002. E-mail address: [email protected] (J. Chen). 0026-2714/$ - see front matter Ó 2012 Published by Elsevier Ltd. http://dx.doi.org/10.1016/j.microrel.2012.07.035

na to protect circuits from over-drive damage by limiting the amplitude of the induced voltage. Many PMCs with the system structure shown in Fig. 1 have been reported for wireless powering system [1,2]. To rectify the received AC signal for its higher efficiency, a CMOS full-wave rectifier is usually used, where a low-dropout regulator with Equivalent Series Resistance (ESR) compensation is preferred. It has been found that the value of ESR in such a LDO is often uncertain and fluctuates with temperature, which may cause LDO unstable and hence the output voltage oscillates. VGS-reference circuits with a simpler design are used to generate the required reference voltage for the regulator [1], but its poor PSRR often cannot meet the system requirement. Also, a popularly-used band-gap reference circuit with Op-Amp has low temperature variation output but its PSRR is not good at high frequency such as 2 MHz. In this paper, a PMC with high power energy and stability is presented which uses bulk commutation technique to suppress rectifier’s leakage current and adopts pre-regulation method to improve the PSR of bandgap voltage reference. A VCCS compensation scheme which is independent of the ESR value is used to ensure the LDO loop is stable within 0–60 mA of load current. Experiment results for the PMC in a wireless powering system are given to show its excellent capability to provide energy supply for a capsule endoscope. The outline of the paper is as follows. In Section 2 the circuit implementations are described, including RF limiter, CMOS rectifier,

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DC enegy in vitro

Vin+(Vin-)

DC output

Class E Power Driver

RF Limiter

Rectifier CS

Voltage Regulator

M3 opamp

MPn

Receiver LC Tank

M4

R2

MP1

Load

On-chip power regulator(except Cs)

R1

M2 M5

Fig. 1. Block diagram of a wireless powering system.

high PSR band-gap and LDO. Section 3 analyzes the experimental results and makes some comparisons. The paper is ended with conclusion in Section 4.

Vin-(Vin+) Fig. 3. Schematic of optimized RF limiter.

buffer is applied to accelerate the open speed of M3 for its large gate capacity. For the protection threshold voltage VDD_TH, the approximate function can be expressed as:

2. Circuit implementation 2.1. RF limiter For the reason that the coupling angle and distance between transmitter and secondary coils often fluctuates and may causes induction voltage too high to destroy the inner circuits, so each input terminal of the chip should connect to a RF limiter to provide over-voltage protection. When the coupling voltage is greater than the security voltage, the RF limiter block starts work and provides a discharge path. Usually Zener diode is suit for the function, but it is not compatible in the CMOS process. A RF limiter compatible with CMOS process to provide the similar over-voltage function in this work is shown in Fig. 2 [3]. As shown in Fig. 2, the left part represent a Zener diode in CMOS process and the right half illustrates how it is realized. M1, R1, M2, R2 form the voltage detection circuits and M3 provides the current discharge path when over-voltage occurs. Note that M1 is made by a series of diode-connect PMOS transistors. When induction voltage Vin is above the protection threshold voltage VDD_TH, the branch of M1 and R1 turns on and the gate voltage of M2 is bigger than threshold voltage of M2. The gate of M3 is pulled down by M2 and M3 acts as a switch to discharge the current from Vin to GND. As M3 is on, the equivalent resistance to the coils is low, so the coupling voltage also decreases. To provide a higher protection voltage, all the transistors in Fig. 2 should use high voltage transistor. Since the two terminal of induction are both AC large signal and we connect one RF limiter between the two input terminals directly, instead of between input terminal and common ground by using two RF limiters, as shown in Fig. 3. However, by directly applying the origin circuit in Fig. 2, we meet severe bulk leakage current problem, so by adding transistors M4 and M5 to set a proper bias of bulk voltage to suppress bulk current leakage effect. The Vin

MP1 R2

nx M1

M3

MPn M2 R1

GND Fig. 2. Schematic of origin RF limiter.

V DD

TH

¼

  1 W R1  lp C ox ðV SG1  V thp Þ2 þ n  V SG1 2 L

ð1Þ

In our design, n is chosen to be 5 and the ratios of width and length of M3 is set to be 200 l/1 l. And VDD_TH has a value about 8 V, which is 20% smaller than the oxide breakdown voltage. The quiescent current is 7.5 lA. Experimental results show that when the input voltage exceeds 8 V, the RF limiter circuit can provide a peak discharge current 70 mA and the slew rate of the discharge current is about 180 mA/V. As the RF limiter circuit starts to work, the input impedance of the PMC will decrease and the electromagnetic induction potential drops subsequently, which bring the input voltage of PMC back to a safe range. Since the voltage generated from the receive coil is limited, this performance can meet the requirement of the high voltage protection. 2.2. CMOS rectifier In most bio-implantable applications, the rectifier block is either a hybrid diode bridge, which increases the size of the implant, or an inefficient half-wave rectifier using the substrate or off-chip diode. An on-chip full-wave bridge rectifier implemented in the CMOS technology can significantly reduce the chip size, help the power regulator to achieve better AC–DC conversion efficiency (compared to half-wave rectifiers), and work fast to operate in the mega hertz range due to eliminating off-chip interconnect and parasitic components. As reported in [4], however, full wave bridge rectifier using diode-connected MOSFET has a serious bulk current leakage problem which greatly reduces power conversion efficiency and may cause latch-up effect because of there is no constant highest voltage available in a rectifier. Because there is not a constant highest voltage node in rectifier, the voltage of the N_well in P-substrate CMOS process cannot be set to an appropriate value and the effect of substrate leakage can be avoided effectively. Using bulk commutation technique can effectively solve this problem [5–7]. As shown in Fig. 4, Vin is the input-AC-coupled voltage, MN1, MN2, MP1 and MP2 are used to build up the basic full-wave bridge rectifier. MP1–MP2 are diode-connected and MN1–MN2 act as switches to return current from grounded substrate to the coil. For each diode-connect PMOS, there are two auxiliary PMOS transistors act as switch to dynamic bias the N-well potential to the higher voltage either Vin+/Vin or Vout to eliminate body effect and reduce the possibility for latch-up. With the help of auxiliary pMOS transistors, leakage current is significantly reduced and power dissipation in the rectifier block is mainly caused by the dropout voltage of the rectifier. The dropout voltage VDS can express as:

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J. Chen et al. / Microelectronics Reliability 53 (2013) 129–135

Vout MP3

MP1

PAC ¼

MP5

MP2

1 T

Z

T

iDS V in dt

ð4Þ

0

where ids is the input current and has a expression

(

Vin +

M P4

-

MN1

CL

RL

MP6

iDS ¼

g¼

Fig. 4. CMOS rectifier with bulk commutation.

M24

M22

M3

M16

Vrec M8

M9

M15 M10

M11

M1

PDC P AC

ð6Þ

R1 8 : 1 Q1

2.3. Wide range high PSR bandgap reference M19

M2

Q2

Vref

A high quality voltage reference with wide input range and high power rejection ratio is of great importance in telemetry powered system. Since the carrier frequency of wireless powering system is 1 MHz, the output of full-wave rectifier will have small ripples with the frequency of around 2 MHz. For the reason that the bandgap and LDO are powered by the output of the rectifier, the small ripples will influent their performance directly. Usually a bandgap with an OpAmp used as voltage reference has a poor PSR at high frequency, such as 2 MHz. A VGS-reference with negative feedback reported in [8] obtains the benefit of improved PSR but without good temperature characteristic. Using the pre-regulate method reported in [10], we proposed a bandgap reference with a wide range high PSRR, as shown in Fig. 5, Q1–Q3, M1–M5, M7–M9 and R1–R3 are used to be the core of the bandgap reference, M6, M10– M19 and C1 form the pre-regulate feedback loop and the start-up circuits is made by M20–M24. For low-cost consideration, the type of transistors Q1–Q3 is lateral BJTs [11]. The idea of pre-regulation is using a feedback to form a local stable voltage and use this voltage to power the core circuits of bandgap. As shown in Fig. 5, when input voltage VDD varies, the local voltage Vrec also changes. Assume the fluctuation of Vrec is vrec, the vgs of transistor M11 is about vrec because the small signal voltage of node four is almost zero due to the large resistor ratio between rds8gm4rds4 and (1/gm12 + 1/gmQ2). The voltage variation of

M17

: 1 Q3

M13

M12

M14

GND Fig. 5. Proposed bandgap reference circuits.

V DS ¼ jV GS j ¼ jV THP j þ

sffiffiffiffiffiffiffiffiffiffiffiffiffiffiffiffiffiffiffiffiffiffiffiffiffiffiffiffiffi 2IDS lp C OX ðW=LÞp

ð2Þ

where VTHP is the MOS threshold voltage, lpCOX is the intrinsic transconductance and (W/L)p is the size of PMOS transistor. VTHP can be minimized in the circuit design by eliminating the body effect and big W/L ratio should be used to minimize the second term in (2) as the rectifier area consumption and its parasitic capacitance permit. The useful DC power PDC delivered to the load RL is:

PDC ¼

ð5Þ

C1

R3 Vref

M18

M6 Vref

R2

M21

V in  V out 6 jV THP j

M4 M5

M20

0;

)

Considering about the conductor impedance, the sizes of MP1 and MN1 are chosen to be 5000 l/0.6 l and 5000 l/0.5 l respectively. Simulation results show that the rectifier can deliver more the 30 mA current to the load. When the amplitude of input sin voltage is between 4.75 and 8 V, output voltage of rectifier is more than 3.2 V to make sure the 3 V LDO can work properly within 200–1000 X load range, and the power conversion is greater than 65%.

VDD

M7

lC OX WL ðV in  V out  jV THP jÞ2 ; V in  V out > jV THP j

The most import coefficient of rectifier is power conversion efficiency and can calculate as in the following expression:

MN2 GND

M23

1 2

V 2out RL

ð3Þ

Input AC signal power can calculate as follows:

Class AB Error Amplifier

VCCS compensation network

VDD

Vb1 M7

1:5

M12 M8

VBP

Vb2

M11

Pass Transistor

M21 Vfb

Vref M1

M2 R f1 Vfb

M3

M4

MP Vout Vout

M5

M6

Rf2

RL

M22

5sC1Vout C1

V1

Vb3

Vfb

V2 VBN

Cout

GND Fig. 6. Schematic of LDO with VCCS compensation and class AB OTA.

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 1 þ sc6 r6 g m11 r6 g m19 r ds16   R3 þ 1=g mQ3 þ 20 log rds5 þ R3 þ 1=g mQ3

PSRV ref VDD jdB ¼ 20 log



ð7Þ

where c6 and r6 are the small signal capacitor and resistor of node 6, gm11, gm19, gmQ3 are transconductance of M11, M19 and Q3, rds16, rds15 are small resistor of M15 and M16, respectively. The first term of function (7) is PSRV rec VDD and the second one is PSRV ref V rec . Simulation results show that the PSR of whole bandgap is 116 dB at low frequency, and at 2 MHz still as low as 47 dB. The measured typical temperature coefficient is about 25.5 ppm/°C with a range from 36 °C up to 40 °C without calibration. 2.4. Low dropout regulator (LDO) Although DC–DC switching voltage regulator has a better power efficiency refer to linear regulator, the need for external inductors and clock signals makes them not available in inductively powered system. Besides linear regulators also take advantage of low output noise to be used in various wireless applications. Usually the ESR of output capacitor which generates a left panel zero at the output of LDO is used to compensate for loop stability. Unfortunately, the value of ESR is usually uncertain and varies with temperature. Here we present a LDO with new compensation scheme to overcome the disadvantage of ESR compensation to keep the loop’s stability within 0–30 mA load current [9]. The whole schematic of LDO with voltage control current source (VCCS) compensation and class AB OTA is shown as Fig. 6. As shown in Fig. 6, the left half is a class AB OTA used as error amplifier and the right half is the VCCS compensation network, MP is a pMOS pass transistor and Rf1 and Rf2 are the feedback sample resistors network. Class OTA is load systematic which reduces systematic offset has a push–pull output stage with a 10 lA slew current to reduce the charging and discharging time of the gate

Fig. 7. Micro-photograph of the chip.

vrec is amplified by cascade amplifier formed by M11, M18, M17, M13

RL=200

2

RL=600 RL=1K

1

No Load

0 0

2

4

6

8

LDO Output Voltage (V)

3

0

20

40

60

LDO Load Current (mA)

(a)

(b)

Measured LDO Line regulation

Measure LDO Load regulation

3 RL=200

2

RL=600 RL=1K

1

No Load

0 0

2

4

6

8

LDO Output Voltage (V)

LDO Input Voltage (V)

2 3.00 1 2.96 0 0.64 mV/mA

2.92

-1 0

20

40

LDO Load Regulation (mV/mA)

LDO Output Voltage (V)

LDO Output Voltage (V)

and the output is converted to current by M19, through voltage– current feedback Vrec keeps stable while VDD changes. The PSR of Vrec refers to VDD and PSR of Vref refers to Vrec are noted as PSRV rec VDD and PSRV ref V rec , respectively. The total PSR of Vref refers to VDD is the sum of the two terms. Using the analysis method reported in [10], we can get the PSR expression of the whole circuit as:

60

LDO Input Voltage (V)

LDO Load Current (mA)

(c)

(d)

Fig. 8. (a) Simulated LDO line regulation for a 0–8 V input voltage. (b) Simulated LDO load regulation. (c) Measured LDO line regulation with different load conditions for a 0– 8 V input voltage. (d) The 3 V output as a function of the load current and load regulation factor for the 3 V output vs. load current.

J. Chen et al. / Microelectronics Reliability 53 (2013) 129–135

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(a) 63.8mA load current switching

(b) 15 mA load current switching Fig. 9. Measured transient response of LDO with load slew rate @20 mA/us.

Table 1 Performance summary. Process Area (including PADs) Quiescent ground current Output regulated voltage Minimum dropout voltage Line regulation Maximum output current Load regulation DVout (full load transient)

0.35 lm 2p2 M 5 V CMOS 1.69 mm2 96 lA 3V 200 mV 5.21 mV/V for 15 mA load current 63.8 mA 0.64 mV/mA <100 mV

Fig. 11. Experiments of the wireless powering system with our PMC to power ten LEDs and to mimic the biomedical condition. (a) Entire system view. (b) Class E amplifier. (c) Experiment system with LED load. (d) Power conversion efficiency with different load current.

connect with Vfb with its value equals to sC1Vout. The parasitic poles of VCCS can be pushed beyond the GBW of LDO and have little effect on the loop stability. For the path from Vout to Vfb, we have the function:

ðV out  V fb Þ

V fb 1 þ 5sC 1 V out ¼ Rf 1 Rf 2

ð8Þ

The transfer function is:

5sC 1 þ R1 V fb f1 ¼ 1 1 V out þ R R f1

Fig. 10. Capsule endoscopy embedded with PMC chip and receive inductance. (a) System block overview. (b) Separated component. (c) Integrated capsule system.

of MP which has a big size thus a big gate capacitor. The ratio of current mirror of M3–M4, M5–M6, and M7–M8 are 1:4, 1:4 and 1:1, respectively. The input of VCCS is the output of LDO Vout, through source follower M21, Vout has a variation equal to V2 though a feedback connected OTA. The ac small signal current sC1Vout generated by V2 through C1 is multiplied by 5 through a 1:5 current mirror and sent to the output of VCCS which connect to Vfb. The whole VCCS network can be seen as a Vout control current source

ð9Þ

f2

The loop contains a left panel zero 1/5Rf1C1 and can be used to compensate the non-dominate pole at the gate of MP. This zero has no relation with the ESR and do not varies with temperature. Simulation results show that the parasitic pole is around 450 kHz and a phase margin larger than 70° can be achieved with the use of VCCS compensation scheme, with a load current range from 0 to 30 mA. The value of Rf1 is chosen according to requirement of the quiescent current, poly-resistive area and system stability at zero loads current mostly. In addition to the location of the zero point, the value of C1 is chosen according to the trade of between current consumption and capacitor area additionally. The DC loop-gain is bigger than 50 dB at 0–15 mA load current and the UGB is about 200 kHz with a 10 lF Tantalum capacitor. 3. Experimental results The PMC chip with on-chip rectifiers is fabricated in a 0.35-lm CMOS 5 V process with special high-voltage LDMOS. And the oxide breakdown voltage of such transistors of this technology is designed to 10 V. The micro-photograph is shown in Fig. 7. Measured results show that the PMC can deliver a load current up to

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J. Chen et al. / Microelectronics Reliability 53 (2013) 129–135

Table 2 Performance comparison with other power regulators for capsule endoscope.

CMOS process Distance between coupling coils (mm) Output voltage (V) Maximum output current Maximum power get through coupling (mW)

This work

[12]

[13]

[14]

0.35 lm 200 3 63.8 191

1.5 lm – 5/3/1.5 17.3 86.5

0.5 lm 25 3.3 1.7 5.61

0.18 lm >50 1.8 40 72

63.8 mA with a quiescent ground current of 96 lA. The simulation and experimental results of input–output transfer characteristic of the low dropout regulator is shown in Fig. 8a and c, with a 0–8 V input voltage in different load conditions LDO has a stable 3 V output voltage when the input exceeds 3.2 V, which means the minimum dropout voltage of LDO is 200 mV. When the load current is 15 mA, the variation of the output is about 25 mV with an variation of the input voltage from 3.2 V to 8 V, that translates to a line regulation of 5.21 mV/V. The simulation and experiment results of output of the low dropout regulator with different loads are shown in Fig. 8b and d. The output current varies from 1 mA to 63.8 mA and the output voltage has a deviation of 40 mV which means a load regulation of 0.64 mV/mA. The load regulation is shown in Fig. 8d and the regulator has average 0.64 mV/V load regulation. Compared with the power regulator reported in [12] which is also used in wireless powering application, and has a 150 mV output voltage variation when load current changes from 0 to 17.3 mA, which means a load regulation of 8.6 mV/mA, the regulator we design here has a much better load regulation performance. In order to demonstrate LDO’s stability to the ESR value of output capacitance, a 2.2 lF ceramic capacitance connected serially with a resistance is placed at the output of the LDO. The value of the resistor changes from 0 to 1 X, the output of the LDO is still stable even if there is a load variation at the output. When the resistor is shorted, the transient response of the LDO with a 0–63.8 mA load transition with the load slew rate about 20 mA/us is shown in Fig. 9a. The overshoot and undershoot voltage are less than 80 mV with a recovery time about 20 ls. The overshoot voltage when load current switching from 15 mA to 0 and undershoot voltage when load current switching from 0 to 15 mA are both less than 25 mV, the setting time is about 10 ls, which is our designable power consumption, as shown in Fig. 9b. Both of the outputs have no oscillation, which proves that the phase margin is large enough to keep the loop’s stability. The measured results of the PMC are summarized in Table 1. Experiment results for a wireless powering system with the PMC design here also show that our design has a good efficiency and stability. As shown in Fig. 10a, the entire system is divided into two parts: Transmitter part and Receive part. The Resonant antenna is excited through a class E power amplifier and an electromagnetic cavity is generated in the center space of circular bucket, as shown in Fig. 11a. The capsule endoscope with PMC embedded is placed in the electromagnetic cavity, imitating practically medical condition. Both coils used in our wireless powering system are cylindershaped and the dimensions are 40 cm for the transmitter coil and 10 mm for the receive inductance coil, respectively, as shown in Fig. 10b. The length of the receive inductance coil is designed to be 13 mm for the limitation of the size of capsule. The inductance of the transmitter and secondary coils are 292 lH and 73 lH, respectively. Both coils were wound out of full copper wire with a diameter of 46 AWG (0.04 mm). Fig. 10b includes all separate components embedded in the capsule and Fig. 10c shows the integrated capsule system, where the endoscope is not included in our current experiment system, which should be the load of the PMC.

When the class E transmit circuits was excited with a voltage of 25 V, the regulator connected to the secondary coil can provide a stable 3 V to a 47 X resistor, which means a more than 63.8 mA load current can be delivered to the load through wireless powering. In order to illustrate our system more intuitively, six parallelly connected LEDs with a 20 mW power for each are used as the load of out system, the PMC can light the LEDs, as shown in Fig. 11c. The secondary coil was covered with a thick meat in our experiment to mimic the effect of human biological tissue on the system. The experimental results showed that the typical frequency shift due to the induction is about 40 kHz but the remote DC power of PMC was still above the required minimum target of 135 mW. The measured results show that the chip and system we design here can provide a more than 191 mW DC energy to the load through wireless powering with a 20 cm coupling distance, and the power conversion efficiency of the whole system is as high as 1.8% at the maximum power energy [15], as shown in Fig. 11d. Table 2 gives a comparison between our results and published papers reported in [12–14], the PCM and system we design here have the highest output energy through electronic–magnetic coupling with the longest coupling distance between the two coils. 4. Conclusion In this paper, a PMC with high efficiency and stability is presented successfully verified by experimental results. Using CMOS process realized a RF limiter to provide simple over-voltage protection and bulk commutation technique to over the substrate leakage current of CMOS rectifier. The proposed wide range and high PSRR bandgap reference suppresses rectifier out ripple and adopt a VCCS compensation to keep LDO stability with 0–63.8 mA load current. Experiment results shows that the PMC we design has a good line and load regulation performance, and the regulator is not dependent on the ESR value of the output capacitance. Experimental results of the wireless powering system with our PMC show that the PMC can provide more than 191 mW DC energy with a 20 cm coupling distance. Our design is not only suitable for the application in wireless powering capsule endoscopes but also used in implanted device or other wireless powering system. Acknowledgments This research was supported by the 863 National High Technology Research and Development Program of China (2008AA010705) and Research Program of Science and Technology Commission of Shanghai (08500700400). References [1] Sodagar Sodagar M, Najafi Khalil. A Wide-range supply-independent CMOS voltage reference for telemetry-powering applications. In: Proc IEEE ICECS, 2002. p. 401–4. [2] Wang G, Liu W, Bashirullah R et al. A closed loop transcutaneous power transfer system for implantable devices with enhanced stability. In: Proc ISCA, vol. 4; 2004. p. 17–20. [3] Kaiser Ulrich, Steinhagen Wolfgang. A low-power transponder IC for highperformance identification systems. IEEE J Solid-State Circ 1995;30(3):306–10.

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