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Sensitivity estimation of CMOS optical BDJ detector
Materials Science and Engineering C 21 (2002) 203 – 210 www.elsevier.com/locate/msec
Sensitivity estimation of CMOS optical BDJ detector G.N. Lu a,*, J.-M. Galvan b, C. Jeloyan a, G. Goumy a, V. Marcoux a a
Laboratoire d’Electronique, Nanotechnologies, Capteurs (LENAC)-Baˆt. 203-Universite´ Lyon 1, 43 bd du 11 Nov 1918, 69622 Villeurbanne, France b Laboratoire d’Imagerie et Signal Acoustique (LISA)-CPE, 43 bd du 11 Novembre 1918, 69616 Villeurbanne, France
Abstract We report the characterisation of the CMOS optical Buried Double p – n Junction (BDJ) detector in terms of sensitivity, with the aim of developing micro-systems for biochemical analysis. We have first used a statistical approach with pA-meters for signal and noise measurements. The obtained results show a minimum noise level limiting the sensitivity performance, due to noise contributions of the coupled measuring instruments. Then we have adopted another approach with implementation of noise-reduction techniques, including lownoise preamplification and synchronous detection. This has allowed the sensitivity estimation much closer to the achievable performance of the detector. The CMOS BDJ detector has two outputs, and its detectivity depends on the used output signals. When the detector is employed as a photodetector with its output current (I1 + I2) as response, the evaluated detectivity is 2.4 1012 cm Hz1/2 W 1. When choosing the other output signal I1, or using both outputs for wavelength detection, the corresponding detectivity is 1.9 1012 cm Hz1/2 W 1. By using the synchronous detection technique, an optical signal of less than 10 fW (10 14 W) is still detectable. D 2002 Published by Elsevier Science B.V. Keywords: Sensitivity; CMOS BDJ detector; Low-noise preamplifiers; Synchronous detection
1. Introduction Optical detection is involved in a variety of analysis techniques, such as fluorescence, chemiluminescence, and absorptiometry. Developments of micro-systems for such purposes usually require photodetectors to be incorporated. With trends toward system miniaturisation, there has been increasing interest in CMOS optical sensing devices such as CMOS photodiodes because they offer many technical and economic advantages related to their monolithic integration with CMOS electronic circuitry [1]. For example, their onchip integration using merely a CMOS process allows improvements in reliability and security for the system because of the reduction in number of components like wire links. Also, the costs of fabrication, assembling, packaging and testing operations can be reduced [2]. The CMOS optical Buried Double p– n Junction (BDJ) detector, in comparison with conventional photodiodes, offers an additional function possibility because it can also be operated as a wavelength-sensitive device [3]. The
* Corresponding author. Laboratoire d’Imagerie et Signal Acoustique (LISA)-CPE, 43 bd du 11 Novembre 1918, 69616 Villeurbanne, France. Tel.: +33-4-7243-2739; fax: +33-4-7243-2740. E-mail address: [email protected] (G.N. Lu).
0928-4931/02/$ - see front matter D 2002 Published by Elsevier Science B.V. PII: S 0 9 2 8 - 4 9 3 1 ( 0 2 ) 0 0 0 8 7 - 5
wavelength-sensing operation is based on the strong wavelength-dependence of silicon optical absorption depth (defined as the reverse of the silicon absorption coefficient). Short-wavelength incident light is absorbed near the silicon surface, while long-wavelength light has a deeper penetration. Since photon absorption results in the generation of electron-hole pairs, the distribution in depth of photogenerated carriers is directly related to wavelengths of incident light. The CMOS BDJ detector basically consists of two buried p –n junctions in stacked form at different silicon depths (Fig. 1). Under reverse-bias conditions, the two buried junctions collect separately photo-carriers in the shadow and deep areas. Two photocurrents are thus produced. One current I1 flows through the shallow junction, while the other I2 flows across the deep junction. Both currents can be determined by measurements. When we calculate a photocurrent ratio, e.g., I2/I1, and use it as the detector response, it exhibits a monotonic increase as a function of wavelength. In the case of nonmonochromatic radiation, the corresponding photocurrent ratio represents a sensitive response to spectral variations of the incident optical signal. Each buried junction has a spectral response, respectively defined as: S1 ðkÞ ¼ I1 =Popt
ð1aÞ
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Fig. 2. CMOS BDJ optical detector designed in an array of 3 3 pixels connected in parallel (the chip was fabricated in a 1 Am CMOS process, by ATMEL-ES2 foundry). Fig. 1. Physical structure of CMOS Buried Double p – n Junction (BDJ) detector.
and S2 ðkÞ ¼ I2 =Popt
ð1bÞ
where Popt is the incident monochromatic light power. It is noted that the photocurrent ratio I2/I1, which is equal to the ratio S2(k)/S1(k), is independent of incident light intensity. Therefore, this term of photocurrent ratio is insensitive to intensity fluctuations. The CMOS BDJ detector under reverse-bias conditions provides two output currents representing, respectively, I1 and (I1 + I2). When using the device for photodetection, one of the two output signals can be chosen as response. The responsivity of the detector depends on the chosen output. Exploiting the output current I1 means that only the detector’s shallow junction is effectively used. The sensitive range of S1(k) is located at short wavelengths around 480 nm [4]. For most applications, the other output signal (I1 + I2) is preferred, because it results from the collection of photo-generated carriers by both shallow and deep junctions, and the corresponding spectral response [S1(k) + S2(k)] covers both visible and near-IR regions. The maximum sensitive wavelengths are around 660 nm. The two-fold operation of the CMOS BDJ detector makes the device attractive especially for opto-chemical detection. It has recently been applied to detecting absorption transmission and fluorescence emission [5 –7]. With the aim of developing miniaturised fluorescence detection systems for microanalysis, we have investigated the CMOS BDJ detector to estimate its sensitivity performance. It is a key performance aspect for determining the instrumental limitation of detection. This performance is closely related to noise effects, and the device characterisation involves signal and noise measurements at low intensity levels. We have carried out this work with two
approaches. Firstly, we have used pA-meters for direct measurements of output signals of the device under test. A statistical method is applied for signal and noise determination. Secondly, we have combined the device under test with low-noise preamplifiers, and used synchronous detection for sensitivity estimation.
2. Statistical measurements The BDJ optical detector to be characterised has been designed and fabricated in a 1 Am CMOS process (Fig. 2). It consists of an array of 3 3 detector elements connected in parallel to form a single large-surface detector. Each element has an active window of 180 180 Am. The detector has a total active surface of about 0.29 mm2, occupying an area of 0.36 mm2. Under reverse-bias condition and at room temperatures, each detector’s junction has a dark current typically below 1 pA. The sensitivity characterisation of the CMOS BDJ detector using the statistical approach is shown in Fig. 3. Via an optic fibre, the BDJ detector under test receives a beam of light coming from a light source, which basically consists of
Fig. 3. Sensitivity estimation via statistical measurements.
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five LEDs of different colours: blue (470 nm), green (525 nm), yellow (590 nm), orange (623 nm) and red (660 nm). It is a programmable light source allowing LED selection and intensity setting. For further intensity attenuation, an attenuator consisting of neutral filters is inserted between the light source output and the detector active surface. At the two detector outputs, two piles serve to ensure reverse-biasing, and two identical pA-meters (KEITHLEY 485) are
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employed to measure output currents. Each pA-meters has a resolution of 0.1 pA for a full range of 2 nA. It includes an input amplifier stage (an inverting transimpedance amplifier), which gives an equivalent input noise voltage of about 5 AV p –p over a noise bandwidth between 0.1 and 10 Hz. The experimental method for statistical measurements consists of successive sampling to acquire a large number of points (e.g., 500 points). The average value of these points
Fig. 4. Photocurrent fluctuations DI1 (a) and DI2 (b) (which represent rms noise over a noise band width of 10 Hz) vs. incident flux.
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represents the signal, while fluctuations around the mean level represents superimposed noise. With the assumption that the noise has a Gaussian distribution, its root mean square (rms) value is equal to the standard deviation of the measured points. We can thus determine signals such as I1, I2 and I2/I1, and their fluctuations in rms values. When input intensity is very low, and unity signal-tonoise ratio is reached (i.e., SNR = 1), we consider the measured signal as minimum measurable level. We have thus obtained a value of 0.2 pA, which is close to the resolution limit of the employed pA-meters (0.1 pA). Measured results also show a minimum noise level between 0.04 and 0.1 pA (Fig. 4), which is intensityindependent. This minimum noise level is mainly due to noise contributions of the coupled input amplifier stage of each pA-meter. By assuming this minimum noise level to be 0.1 pA, which corresponds to a rms noise current of about 0.03 pA in a 1-Hz bandwidth, we can deduce the noise equivalent power (NEP) from the spectral responses of the BDJ detector, and calculate the detectivity defined by D = A1/2/NEP, where A is the active area of the detector (0.29 mm2). For input optical signals at wavelengths in the visible range, the obtained detectivity corresponding to both detector outputs is typically around 0.7 1012 cm Hz1/2 W 1.
3. Measurements with low-noise preamplifiers and synchronous detection As we have seen in the Statistical measurements section, the additional noise of measuring instruments coupled at the detector outputs may have major contributions and set a limit to detection sensitivity. One approach to minimise this additional noise effect is to combine the BDJ detector under test with low-noise preamplifiers. This has led us to design a testing circuit including the detector and preamplifiers. 3.1. Low-noise preamplifiers The testing circuit has two identical channels (A and B) connected to both outputs of the detector under test (Fig. 5). Each channel includes a low-noise transimpedance preamplifier and an ac-gain-variable amplifier. Each amplifier stage output is accessible for signal and noise measurement. The input stage of each channel is a transimpedance amplifier fulfilling several requirements. It converts the corresponding detector output current into a more readily observable voltage. Also, it has a low input impedance as a load for the detector, which is needed for the detector to be operated in photoconductive mode. In addition to that, lowimpedance loading reduces the effects of parasitic capacitances at detector outputs. The transimpedance amplifier is formed using a highgain, low-noise op-amp and a feedback resistor RfA,B (the
Fig. 5. Two-channel testing circuit associated with the detector under test.
subscript ‘A,B’ means ‘A’ or ‘B’). The noise characteristics of the op-amp represented by two equivalent input noise generators (1 noise voltage and 1 noise current) will be presented in the following section. At low frequencies, the transfer function of the stage is given by: AzA;B ¼ RfA;B =ð1 þ jxx1 c Þ
ð2aÞ
with xc ¼ RfA;B CfA;B
ð2bÞ
where CfA,B ( c 2 pF) is an equivalent capacitor in parallel with RfA,B. The value of RfA,B ( = 5 1010 V) is chosen with the consideration of its noise effect. Even at the chosen value, RfA,B generates a thermal noise comparable to a shot noise from a 1-pA dark junction current in the detector ( < 1 pA at room temperatures). Apparently, increasing RfA,B increases noise voltage at the amplifier output, but the increase in signal amplitude is much faster. This leads to a higher signal-to-noise ratio (SNR). However, the consequence of raising Rf is the decrease of cut-off frequency ( fc c 1.5 Hz for the chosen value of RfA,B). It limits the operating frequency and measuring rate in application cases. Another problem is that when 1/f noise is dominant, operation at a lower frequency means the presence of a higher noise level. One solution to this latter problem is to use selective filters for noise rejection, in which case the increase in noise density with frequency lowering will be compensated by the decrease in noise bandwidth. The second stage in each channel is a band-pass gainvariable amplifier consisting of a low-noise op-amp and a RC network. This stage performs further signal amplification to a sufficient level so as to minimise SNR deterioration when the output is connected to measuring instruments. The bandwidth of the stage is limited to a frequency range of interest. This bandwidth narrowing serves to reduce noise
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level for avoiding amplifier saturation induced by overwhelming noise. The transfer function of the stage is: 2 1 1 AvA;B ¼ Avo jxx1 cl =½1 þ jxðxcl þ xch Þ þ ðjxÞ
ðxcl xch Þ1
ð3Þ
where Avo is the maximum voltage gain in the bandwidth, xcl and xch are cutoff angular frequencies determining the bandwidth. This stage has a signal bandwidth ranging from 0.3 to 6 Hz, with a gain selection between 22 and 42 dB through a switch control. This two-channel testing circuit is realised in a shielded board incorporating the detector for the test. 3.2. Measurements with synchronous detection The scheme of sensitivity characterisation using preamplifiers and synchronous detection technique is shown in Fig. 6. The input optical signal is provided by a programmable light source, which can be controlled for signal chopping (i.e., on/off modulation). For obtaining very low intensity levels required for the testing, an attenuator is added. The optical signal is fed to the detector surface via an optical fibre cable. The main measuring instruments are an oscilloscope and a lock-in amplifier. The oscilloscope serves to monitor signal and noise. The lock-in amplifier is employed for sensitive signal and noise measurements. This latter instrument enables us to use the synchronous detection technique for weak signal recovery. To do so, the reference oscillator output of the instrument is linked to the light source for the control of signal chopping. Such a lock-in system has a SNR improvement factor [8]: SNRo =SNRi ¼ Bi =Bo
ð4Þ
where SNRo and SNRi represent, respectively, its output and input signal-to-noise ratios, and Bi and Bo are, respectively, its input and output noise bandwidths. An adequate output noise bandwidth can be chosen via time constant setting. There is a trade-off between detection sensitivity and measurement time.
Fig. 6. Sensitivity evaluation using low-noise preamplifiers and synchronous detection technique.
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When used for signal measurement, the lock-in amplifier performs amplitude demodulation, and provides a quasistatic output signal proportional to the amplitude of input modulated signal. Also, it can be operated to measure noise spectral density. A noise spectrum can be obtained by measuring rms noise over a transmission window ( = 2Bo) centred at a reference frequency, and by scanning the reference frequency over the frequency range of interest. Automatic measurement can be made via PC-control with a data-acquisition software. 3.3. Detectivity evaluation The detectivity corresponding to each detector output is estimated as follows. First, the noise spectral density at each preamplifier output is measured. Then the measured noise voltage is reflected to the preamplifier input. This gives a total noise current (ItA for the detector output A, or ItB for its output B) contributed by the detector and the preamplifier. We can derive the noise equivalent input (optical) power (NEP) referring, respectively, to the detector outputs A and B: ðNEPÞA ¼ 4:7ItA =S1
ð5aÞ
and ðNEPÞB ¼ 4:7ItB =ðS1 þ S2 Þ
ð5bÞ
where the coefficient value 4.7 represents a ratio between on/off signal magnitude and rms noise in the case of unity signal-to-noise ratio, i.e., SNR = 1. Finally, the corresponding detectivity is determined according to D = A1/2/(NEP). Let us first consider the BDJ detector as a photodetector, with the chosen output current (I1 + I2) as its response. In this case the corresponding detectivity of the detector can be derived from noise measurement at the preamplifier outputs of channel B. Fig. 7 shows the measured noise spectral density of the second amplifier stage output VoB2. The results are obtained with no optical signal input. Two curves correspond, respectively, to two chosen gain of the second amplifier stage: Av0 = 11 and 110. They have a fairly flat portion at frequencies around 1 or 2 Hz, and a faster attenuation starting at a few hertz. By reflecting these noise voltages to the input of channel B (by dividing them by corresponding transfer functions), we can determine the level of the total noise current ItB ( c 1.9 fA Hz 1/2) which is practically a white noise over the observed frequency range (0.5 – 100 Hz). The level of the total noise current ItB determines the detectivity of the detector referring to channel B (see Eq. (5b)). For an optical signal centred at 525 nm, for example, the corresponding responsivity of the detector is about 0.39 A/W. Accordingly, the obtained detectivity is 2.4 1012 cm Hz1/2 W 1.
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Fig. 7. Output noise voltages of channel B (in mV Hz 1/2).
Similarly, the detectivity referring to channel A is evaluated by measuring the output noise of the second amplifier stage of the channel VoA2. Measured results of VoA2 (in mVHz 1/2) for Av0 = 11 and 110 are shown in Fig. 8. The estimated detectivity referring to this channel is about 1.9 1012 cm Hz1/2 W 1. For the detector to be used as a wavelength-sensing device, its corresponding detectivity will also be 1.9 1012 cm Hz1/2 W 1. These results show higher detectivity performance compared with those previously obtained from statistical measurements. This improvement is due to the use of low-noise preamplifiers, which have lower additional noise contributions. 3.4. Dominant noise identification The total noise currents at both detector outputs ItA and ItB are contributed either by noise generators of the detector and by those of the coupled transimpedance amplifiers. Let us first consider noise in the detector. A noise model can be built with two noise current generators Ij1 and Ij2 related, respectively, to the two buried junctions (see Fig. 9).
Both noise generators of the detector Ij1 and Ij2 are a superimposition of several principal types of noise: shot noise, generation – recombination (G – R) noise and 1/f noise. Shot noise in the detector is produced when a current flows through a junction. Such a junction current may include a photocurrent component and a dark current. This shot noise can be estimated by Ish1,2=[2q(Iph + Idark)Df]1/2. It becomes intensity-dependent when Iph > Idark. G – R noise IG – R1,2 is due to random mechanism of generation, recombination, and trapping of carriers. It is often the dominant noise mechanism in photodetectors [9]. The spectral distribution of G –R noise is white up to a frequency determined by the lifetime of the carriers in the photodetector [10]. Obviously, 1/f noise also has a characteristic frequency distribution. Both G –R noise and 1/f noise are proportional to junction current flow, and thus intensity-dependent. Compared with the shot noise which is proportional to the square root of junction current, G – R noise and 1/f noise have a faster increase with input signal intensity. On the other hand, each coupled transimpedance amplifier has three noise sources: two equivalent input noise generators EnA,B and InA,B for the op-amp and a thermal noise generator IfA,B for the feedback resistor RfA,B. The
Fig. 8. Output noise voltages of channel A (in mV Hz 1/2).
G.N. Lu et al. / Materials Science and Engineering C 21 (2002) 203–210
input noise voltage generator EnA,B is a superimposition of a 1/f noise and a white noise. According to the characteristics of the op-amp, this noise voltage has a typical value of 60 nV/Hz1/2 at 1 Hz, with a corner frequency at 100 Hz. The other equivalent input noise generator of the op-amp is a noise current InA,B, which is a white-noise at low frequencies (before starting a steady increase with frequency due to Miller effect). This noise current is a shot noise due to leakage current of input transistor’s gate of the op-amp, given by InA,B=(2qIdc)1/2. This noise current has a typical value of 0.6 fA/Hz1/2. The third noise generator from the feedback resistor can be calculated by IfA,B=(4kT/RfA,B)1/2. It gives a comparable noise current of 0.6 fA/Hz1/2. It is noted that all the noise generators in each transimpedance amplifier are intensity-independent. From the obtained noise spectral densities ItA and ItB (deduced from measured results of VoA2 and VoB2) which show no frequency dependence over the observed frequency range, we can first rule out the possibility of 1/f noise preponderance. The remaining noise generators contributing to ItA and ItB can be written as: 2 2 2 2 2 2 2 ¼ Ij12 þ InA þ IfA ¼ IGR1 þ Ish1 þ InA þ IfA ItA
ð6aÞ
2 2 2 ¼ Ij12 þ Ij22 þ InB þ IfB ItB 2 2 2 2 2 2 ¼ IGR1 þ Ish1 þ IGR2 þ Ish2 þ InB þ IfB
ð6bÞ
Apart from the G –R noise sources IG – R1 and IG – R2, the other terms in the above expression can be calculated by taking the typical values of dark junction currents and the input bias current of the op-amp. Summing the calculated terms (not including IG – R1 and IG – R2) gives noise levels below 1.1 fAHz 1/2. They are distinctly lower than the experimentally obtained total noise currents (e.g., ItB = 1.9 fA Hz 1/2). Such differences may be explained by assum-
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Fig. 10. Noise current ItB as a function of optical signal intensity.
ing that G – R noise may have a major contribution, because it is not included in the calculations. In order to confirm this assumption, we have observed the dependence of noise on signal intensity. Taking channel B as an example, we can see from Fig. 10 that the measured noise current ItB (in fA Hz 1/2 at 1 Hz) has a steady increase with signal intensity, which indicates the preponderance of noise generators in the detector. Furthermore, the noise level seems to have a linear intensity-dependence, which accounts for the presence of dominant G – R noise. 3.5. Minimum detectable signal By using the synchronous detection technique, we can recover a signal buried in noise. According to Eq. (4), provided that the output bandwidth of the lock-in amplifier is much narrower than its input bandwidth, i.e., BobBi, a signal in the case where SNRib1 is still detectable. This condition can be met by increasing the time constant of the instrument sc, which is inversely proportional to Bo. This will lead to sensitivity improvement at the expense of measurement rate. Experimentally, we have obtained a minimum detectable optical signal of 9.6 fW with sc = 1 s, and 3.8 fW with sc = 5 s.
4. Conclusion
Fig. 9. Schematic diagram of the CMOS BDJ detector for noise analysis [4].
We have estimated the sensitivity of the CMOS optical BDJ detector, which is a key performance aspect for its applications to biochemical analysis. The detector sensitivity is closely related to noise effects, and this performance will deteriorate if the coupled measuring instruments or associated electronics have major noise contributions. To minimise additional noise, we have combined low-noise preamplifiers with the detector. This has led to the sensitivity estimation much closer to the achievable performance of the device. We have thus evaluated the detectivity of the BDJ detector, which depends on its operation and the use of its two outputs (which provide, respectively, two currents, I1
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and (I1 + I2)). For photodetection with the use of (I1 + I2) as the detector response, the corresponding detectivity is 2.4 1012 cm Hz1/2 W 1. When choosing the other output signal I1, or using both outputs for wavelength detection, the obtained detectivity is 1.9 1012 cm Hz1/2 W 1. The measured noise seems to be mainly contributed by the G – R noise sources of the detector. This assumption is confirmed by observation of its intensity-dependence. We have also used synchronous detection for weak signal recovery. At a reasonable measurement rate, the minimum detectable optical signal is lower than 10 fW (10 14 W).
[3]
[4]
[5] [6]
[7]
References [8] [1] M.L. Simpson, G.S. Sayler, G. Patterson, D.E. Nivens, E.K. Bolton, J.M. Rochelle, J.C. Arnott, B.M. Applegate, S. Ripp, M.A. Guillorn, An integrated CMOS microluminometer for low-level luminescence sensing in the bioluminescent bioreporter integrated circuit, Sens. Actuators, B, Chem. 72 (2001) 134 – 140. [2] A. Pauchard, B. Furrer, Z. Randjelovic, A. Rochas, D. Manic, R.S.
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Popovic, Integrated microsystem for blue/UV detection, Sens. Actuators, A, Phys. 85 (2000) 99 – 105. G.N. Lu, M. Ben Chouikha, G. Sou, M. Sedjil, Colour detection using a buried double p – n junction structure implemented in the CMOS process, Electron. Lett. 32 (6) (1996) 594 – 596. G.N. Lu, G. Sou, F. Devigny, G. Guillaud, Design and testing of a CMOS BDJ detector for integrated microanalysis systems, Microelectron. J. 32 (2001) 227 – 234. M. Sedjil, G.N. Lu, A sea water pH determination method using BDJ detector, Meas. Sci. Technol. 9 (1998) 1587 – 1592. G.N. Lu, A dual-wavelength method using the BDJ detector and its application to iron concentration measurement, Meas. Sci. Technol. 10 (1999) 312 – 315. G.N. Lu, G. Guillaud, G. Sou, F. Devigny, M. Pitaval, P. Morin, Investigation of CMOS BDJ detector for fluorescence detection in microarray analysis, IEEE-EMBS’2000, Oct. 2000, Lyon, France, Proc., 2000, pp. 381 – 386. M.L. Meade, Lock-in Amplifiers: Principles and Applications, P. Peregrinus on behalf of IEE, London, Appendix 3, 1983. C.D. Motchenbacher, J.A. Connelly, Low-Noise Electronic System Design, Wiley-Interscience, New York, 1993, Chap. 8. K.M. Van Vliet, J.R. Fassett, in: R.E. Burgess (Ed.), Fluctuations Phenomena in Solids, Academic Press, New York, 1965, Chap. 7.
Sensitivity estimation of CMOS optical BDJ detector G.N. Lu a,*, J.-M. Galvan b, C. Jeloyan a, G. Goumy a, V. Marcoux a a
Laboratoire d’Electronique, Nanotechnologies, Capteurs (LENAC)-Baˆt. 203-Universite´ Lyon 1, 43 bd du 11 Nov 1918, 69622 Villeurbanne, France b Laboratoire d’Imagerie et Signal Acoustique (LISA)-CPE, 43 bd du 11 Novembre 1918, 69616 Villeurbanne, France
Abstract We report the characterisation of the CMOS optical Buried Double p – n Junction (BDJ) detector in terms of sensitivity, with the aim of developing micro-systems for biochemical analysis. We have first used a statistical approach with pA-meters for signal and noise measurements. The obtained results show a minimum noise level limiting the sensitivity performance, due to noise contributions of the coupled measuring instruments. Then we have adopted another approach with implementation of noise-reduction techniques, including lownoise preamplification and synchronous detection. This has allowed the sensitivity estimation much closer to the achievable performance of the detector. The CMOS BDJ detector has two outputs, and its detectivity depends on the used output signals. When the detector is employed as a photodetector with its output current (I1 + I2) as response, the evaluated detectivity is 2.4 1012 cm Hz1/2 W 1. When choosing the other output signal I1, or using both outputs for wavelength detection, the corresponding detectivity is 1.9 1012 cm Hz1/2 W 1. By using the synchronous detection technique, an optical signal of less than 10 fW (10 14 W) is still detectable. D 2002 Published by Elsevier Science B.V. Keywords: Sensitivity; CMOS BDJ detector; Low-noise preamplifiers; Synchronous detection
1. Introduction Optical detection is involved in a variety of analysis techniques, such as fluorescence, chemiluminescence, and absorptiometry. Developments of micro-systems for such purposes usually require photodetectors to be incorporated. With trends toward system miniaturisation, there has been increasing interest in CMOS optical sensing devices such as CMOS photodiodes because they offer many technical and economic advantages related to their monolithic integration with CMOS electronic circuitry [1]. For example, their onchip integration using merely a CMOS process allows improvements in reliability and security for the system because of the reduction in number of components like wire links. Also, the costs of fabrication, assembling, packaging and testing operations can be reduced [2]. The CMOS optical Buried Double p– n Junction (BDJ) detector, in comparison with conventional photodiodes, offers an additional function possibility because it can also be operated as a wavelength-sensitive device [3]. The
* Corresponding author. Laboratoire d’Imagerie et Signal Acoustique (LISA)-CPE, 43 bd du 11 Novembre 1918, 69616 Villeurbanne, France. Tel.: +33-4-7243-2739; fax: +33-4-7243-2740. E-mail address: [email protected] (G.N. Lu).
0928-4931/02/$ - see front matter D 2002 Published by Elsevier Science B.V. PII: S 0 9 2 8 - 4 9 3 1 ( 0 2 ) 0 0 0 8 7 - 5
wavelength-sensing operation is based on the strong wavelength-dependence of silicon optical absorption depth (defined as the reverse of the silicon absorption coefficient). Short-wavelength incident light is absorbed near the silicon surface, while long-wavelength light has a deeper penetration. Since photon absorption results in the generation of electron-hole pairs, the distribution in depth of photogenerated carriers is directly related to wavelengths of incident light. The CMOS BDJ detector basically consists of two buried p –n junctions in stacked form at different silicon depths (Fig. 1). Under reverse-bias conditions, the two buried junctions collect separately photo-carriers in the shadow and deep areas. Two photocurrents are thus produced. One current I1 flows through the shallow junction, while the other I2 flows across the deep junction. Both currents can be determined by measurements. When we calculate a photocurrent ratio, e.g., I2/I1, and use it as the detector response, it exhibits a monotonic increase as a function of wavelength. In the case of nonmonochromatic radiation, the corresponding photocurrent ratio represents a sensitive response to spectral variations of the incident optical signal. Each buried junction has a spectral response, respectively defined as: S1 ðkÞ ¼ I1 =Popt
ð1aÞ
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Fig. 2. CMOS BDJ optical detector designed in an array of 3 3 pixels connected in parallel (the chip was fabricated in a 1 Am CMOS process, by ATMEL-ES2 foundry). Fig. 1. Physical structure of CMOS Buried Double p – n Junction (BDJ) detector.
and S2 ðkÞ ¼ I2 =Popt
ð1bÞ
where Popt is the incident monochromatic light power. It is noted that the photocurrent ratio I2/I1, which is equal to the ratio S2(k)/S1(k), is independent of incident light intensity. Therefore, this term of photocurrent ratio is insensitive to intensity fluctuations. The CMOS BDJ detector under reverse-bias conditions provides two output currents representing, respectively, I1 and (I1 + I2). When using the device for photodetection, one of the two output signals can be chosen as response. The responsivity of the detector depends on the chosen output. Exploiting the output current I1 means that only the detector’s shallow junction is effectively used. The sensitive range of S1(k) is located at short wavelengths around 480 nm [4]. For most applications, the other output signal (I1 + I2) is preferred, because it results from the collection of photo-generated carriers by both shallow and deep junctions, and the corresponding spectral response [S1(k) + S2(k)] covers both visible and near-IR regions. The maximum sensitive wavelengths are around 660 nm. The two-fold operation of the CMOS BDJ detector makes the device attractive especially for opto-chemical detection. It has recently been applied to detecting absorption transmission and fluorescence emission [5 –7]. With the aim of developing miniaturised fluorescence detection systems for microanalysis, we have investigated the CMOS BDJ detector to estimate its sensitivity performance. It is a key performance aspect for determining the instrumental limitation of detection. This performance is closely related to noise effects, and the device characterisation involves signal and noise measurements at low intensity levels. We have carried out this work with two
approaches. Firstly, we have used pA-meters for direct measurements of output signals of the device under test. A statistical method is applied for signal and noise determination. Secondly, we have combined the device under test with low-noise preamplifiers, and used synchronous detection for sensitivity estimation.
2. Statistical measurements The BDJ optical detector to be characterised has been designed and fabricated in a 1 Am CMOS process (Fig. 2). It consists of an array of 3 3 detector elements connected in parallel to form a single large-surface detector. Each element has an active window of 180 180 Am. The detector has a total active surface of about 0.29 mm2, occupying an area of 0.36 mm2. Under reverse-bias condition and at room temperatures, each detector’s junction has a dark current typically below 1 pA. The sensitivity characterisation of the CMOS BDJ detector using the statistical approach is shown in Fig. 3. Via an optic fibre, the BDJ detector under test receives a beam of light coming from a light source, which basically consists of
Fig. 3. Sensitivity estimation via statistical measurements.
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five LEDs of different colours: blue (470 nm), green (525 nm), yellow (590 nm), orange (623 nm) and red (660 nm). It is a programmable light source allowing LED selection and intensity setting. For further intensity attenuation, an attenuator consisting of neutral filters is inserted between the light source output and the detector active surface. At the two detector outputs, two piles serve to ensure reverse-biasing, and two identical pA-meters (KEITHLEY 485) are
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employed to measure output currents. Each pA-meters has a resolution of 0.1 pA for a full range of 2 nA. It includes an input amplifier stage (an inverting transimpedance amplifier), which gives an equivalent input noise voltage of about 5 AV p –p over a noise bandwidth between 0.1 and 10 Hz. The experimental method for statistical measurements consists of successive sampling to acquire a large number of points (e.g., 500 points). The average value of these points
Fig. 4. Photocurrent fluctuations DI1 (a) and DI2 (b) (which represent rms noise over a noise band width of 10 Hz) vs. incident flux.
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represents the signal, while fluctuations around the mean level represents superimposed noise. With the assumption that the noise has a Gaussian distribution, its root mean square (rms) value is equal to the standard deviation of the measured points. We can thus determine signals such as I1, I2 and I2/I1, and their fluctuations in rms values. When input intensity is very low, and unity signal-tonoise ratio is reached (i.e., SNR = 1), we consider the measured signal as minimum measurable level. We have thus obtained a value of 0.2 pA, which is close to the resolution limit of the employed pA-meters (0.1 pA). Measured results also show a minimum noise level between 0.04 and 0.1 pA (Fig. 4), which is intensityindependent. This minimum noise level is mainly due to noise contributions of the coupled input amplifier stage of each pA-meter. By assuming this minimum noise level to be 0.1 pA, which corresponds to a rms noise current of about 0.03 pA in a 1-Hz bandwidth, we can deduce the noise equivalent power (NEP) from the spectral responses of the BDJ detector, and calculate the detectivity defined by D = A1/2/NEP, where A is the active area of the detector (0.29 mm2). For input optical signals at wavelengths in the visible range, the obtained detectivity corresponding to both detector outputs is typically around 0.7 1012 cm Hz1/2 W 1.
3. Measurements with low-noise preamplifiers and synchronous detection As we have seen in the Statistical measurements section, the additional noise of measuring instruments coupled at the detector outputs may have major contributions and set a limit to detection sensitivity. One approach to minimise this additional noise effect is to combine the BDJ detector under test with low-noise preamplifiers. This has led us to design a testing circuit including the detector and preamplifiers. 3.1. Low-noise preamplifiers The testing circuit has two identical channels (A and B) connected to both outputs of the detector under test (Fig. 5). Each channel includes a low-noise transimpedance preamplifier and an ac-gain-variable amplifier. Each amplifier stage output is accessible for signal and noise measurement. The input stage of each channel is a transimpedance amplifier fulfilling several requirements. It converts the corresponding detector output current into a more readily observable voltage. Also, it has a low input impedance as a load for the detector, which is needed for the detector to be operated in photoconductive mode. In addition to that, lowimpedance loading reduces the effects of parasitic capacitances at detector outputs. The transimpedance amplifier is formed using a highgain, low-noise op-amp and a feedback resistor RfA,B (the
Fig. 5. Two-channel testing circuit associated with the detector under test.
subscript ‘A,B’ means ‘A’ or ‘B’). The noise characteristics of the op-amp represented by two equivalent input noise generators (1 noise voltage and 1 noise current) will be presented in the following section. At low frequencies, the transfer function of the stage is given by: AzA;B ¼ RfA;B =ð1 þ jxx1 c Þ
ð2aÞ
with xc ¼ RfA;B CfA;B
ð2bÞ
where CfA,B ( c 2 pF) is an equivalent capacitor in parallel with RfA,B. The value of RfA,B ( = 5 1010 V) is chosen with the consideration of its noise effect. Even at the chosen value, RfA,B generates a thermal noise comparable to a shot noise from a 1-pA dark junction current in the detector ( < 1 pA at room temperatures). Apparently, increasing RfA,B increases noise voltage at the amplifier output, but the increase in signal amplitude is much faster. This leads to a higher signal-to-noise ratio (SNR). However, the consequence of raising Rf is the decrease of cut-off frequency ( fc c 1.5 Hz for the chosen value of RfA,B). It limits the operating frequency and measuring rate in application cases. Another problem is that when 1/f noise is dominant, operation at a lower frequency means the presence of a higher noise level. One solution to this latter problem is to use selective filters for noise rejection, in which case the increase in noise density with frequency lowering will be compensated by the decrease in noise bandwidth. The second stage in each channel is a band-pass gainvariable amplifier consisting of a low-noise op-amp and a RC network. This stage performs further signal amplification to a sufficient level so as to minimise SNR deterioration when the output is connected to measuring instruments. The bandwidth of the stage is limited to a frequency range of interest. This bandwidth narrowing serves to reduce noise
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level for avoiding amplifier saturation induced by overwhelming noise. The transfer function of the stage is: 2 1 1 AvA;B ¼ Avo jxx1 cl =½1 þ jxðxcl þ xch Þ þ ðjxÞ
ðxcl xch Þ1
ð3Þ
where Avo is the maximum voltage gain in the bandwidth, xcl and xch are cutoff angular frequencies determining the bandwidth. This stage has a signal bandwidth ranging from 0.3 to 6 Hz, with a gain selection between 22 and 42 dB through a switch control. This two-channel testing circuit is realised in a shielded board incorporating the detector for the test. 3.2. Measurements with synchronous detection The scheme of sensitivity characterisation using preamplifiers and synchronous detection technique is shown in Fig. 6. The input optical signal is provided by a programmable light source, which can be controlled for signal chopping (i.e., on/off modulation). For obtaining very low intensity levels required for the testing, an attenuator is added. The optical signal is fed to the detector surface via an optical fibre cable. The main measuring instruments are an oscilloscope and a lock-in amplifier. The oscilloscope serves to monitor signal and noise. The lock-in amplifier is employed for sensitive signal and noise measurements. This latter instrument enables us to use the synchronous detection technique for weak signal recovery. To do so, the reference oscillator output of the instrument is linked to the light source for the control of signal chopping. Such a lock-in system has a SNR improvement factor [8]: SNRo =SNRi ¼ Bi =Bo
ð4Þ
where SNRo and SNRi represent, respectively, its output and input signal-to-noise ratios, and Bi and Bo are, respectively, its input and output noise bandwidths. An adequate output noise bandwidth can be chosen via time constant setting. There is a trade-off between detection sensitivity and measurement time.
Fig. 6. Sensitivity evaluation using low-noise preamplifiers and synchronous detection technique.
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When used for signal measurement, the lock-in amplifier performs amplitude demodulation, and provides a quasistatic output signal proportional to the amplitude of input modulated signal. Also, it can be operated to measure noise spectral density. A noise spectrum can be obtained by measuring rms noise over a transmission window ( = 2Bo) centred at a reference frequency, and by scanning the reference frequency over the frequency range of interest. Automatic measurement can be made via PC-control with a data-acquisition software. 3.3. Detectivity evaluation The detectivity corresponding to each detector output is estimated as follows. First, the noise spectral density at each preamplifier output is measured. Then the measured noise voltage is reflected to the preamplifier input. This gives a total noise current (ItA for the detector output A, or ItB for its output B) contributed by the detector and the preamplifier. We can derive the noise equivalent input (optical) power (NEP) referring, respectively, to the detector outputs A and B: ðNEPÞA ¼ 4:7ItA =S1
ð5aÞ
and ðNEPÞB ¼ 4:7ItB =ðS1 þ S2 Þ
ð5bÞ
where the coefficient value 4.7 represents a ratio between on/off signal magnitude and rms noise in the case of unity signal-to-noise ratio, i.e., SNR = 1. Finally, the corresponding detectivity is determined according to D = A1/2/(NEP). Let us first consider the BDJ detector as a photodetector, with the chosen output current (I1 + I2) as its response. In this case the corresponding detectivity of the detector can be derived from noise measurement at the preamplifier outputs of channel B. Fig. 7 shows the measured noise spectral density of the second amplifier stage output VoB2. The results are obtained with no optical signal input. Two curves correspond, respectively, to two chosen gain of the second amplifier stage: Av0 = 11 and 110. They have a fairly flat portion at frequencies around 1 or 2 Hz, and a faster attenuation starting at a few hertz. By reflecting these noise voltages to the input of channel B (by dividing them by corresponding transfer functions), we can determine the level of the total noise current ItB ( c 1.9 fA Hz 1/2) which is practically a white noise over the observed frequency range (0.5 – 100 Hz). The level of the total noise current ItB determines the detectivity of the detector referring to channel B (see Eq. (5b)). For an optical signal centred at 525 nm, for example, the corresponding responsivity of the detector is about 0.39 A/W. Accordingly, the obtained detectivity is 2.4 1012 cm Hz1/2 W 1.
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Fig. 7. Output noise voltages of channel B (in mV Hz 1/2).
Similarly, the detectivity referring to channel A is evaluated by measuring the output noise of the second amplifier stage of the channel VoA2. Measured results of VoA2 (in mVHz 1/2) for Av0 = 11 and 110 are shown in Fig. 8. The estimated detectivity referring to this channel is about 1.9 1012 cm Hz1/2 W 1. For the detector to be used as a wavelength-sensing device, its corresponding detectivity will also be 1.9 1012 cm Hz1/2 W 1. These results show higher detectivity performance compared with those previously obtained from statistical measurements. This improvement is due to the use of low-noise preamplifiers, which have lower additional noise contributions. 3.4. Dominant noise identification The total noise currents at both detector outputs ItA and ItB are contributed either by noise generators of the detector and by those of the coupled transimpedance amplifiers. Let us first consider noise in the detector. A noise model can be built with two noise current generators Ij1 and Ij2 related, respectively, to the two buried junctions (see Fig. 9).
Both noise generators of the detector Ij1 and Ij2 are a superimposition of several principal types of noise: shot noise, generation – recombination (G – R) noise and 1/f noise. Shot noise in the detector is produced when a current flows through a junction. Such a junction current may include a photocurrent component and a dark current. This shot noise can be estimated by Ish1,2=[2q(Iph + Idark)Df]1/2. It becomes intensity-dependent when Iph > Idark. G – R noise IG – R1,2 is due to random mechanism of generation, recombination, and trapping of carriers. It is often the dominant noise mechanism in photodetectors [9]. The spectral distribution of G –R noise is white up to a frequency determined by the lifetime of the carriers in the photodetector [10]. Obviously, 1/f noise also has a characteristic frequency distribution. Both G –R noise and 1/f noise are proportional to junction current flow, and thus intensity-dependent. Compared with the shot noise which is proportional to the square root of junction current, G – R noise and 1/f noise have a faster increase with input signal intensity. On the other hand, each coupled transimpedance amplifier has three noise sources: two equivalent input noise generators EnA,B and InA,B for the op-amp and a thermal noise generator IfA,B for the feedback resistor RfA,B. The
Fig. 8. Output noise voltages of channel A (in mV Hz 1/2).
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input noise voltage generator EnA,B is a superimposition of a 1/f noise and a white noise. According to the characteristics of the op-amp, this noise voltage has a typical value of 60 nV/Hz1/2 at 1 Hz, with a corner frequency at 100 Hz. The other equivalent input noise generator of the op-amp is a noise current InA,B, which is a white-noise at low frequencies (before starting a steady increase with frequency due to Miller effect). This noise current is a shot noise due to leakage current of input transistor’s gate of the op-amp, given by InA,B=(2qIdc)1/2. This noise current has a typical value of 0.6 fA/Hz1/2. The third noise generator from the feedback resistor can be calculated by IfA,B=(4kT/RfA,B)1/2. It gives a comparable noise current of 0.6 fA/Hz1/2. It is noted that all the noise generators in each transimpedance amplifier are intensity-independent. From the obtained noise spectral densities ItA and ItB (deduced from measured results of VoA2 and VoB2) which show no frequency dependence over the observed frequency range, we can first rule out the possibility of 1/f noise preponderance. The remaining noise generators contributing to ItA and ItB can be written as: 2 2 2 2 2 2 2 ¼ Ij12 þ InA þ IfA ¼ IGR1 þ Ish1 þ InA þ IfA ItA
ð6aÞ
2 2 2 ¼ Ij12 þ Ij22 þ InB þ IfB ItB 2 2 2 2 2 2 ¼ IGR1 þ Ish1 þ IGR2 þ Ish2 þ InB þ IfB
ð6bÞ
Apart from the G –R noise sources IG – R1 and IG – R2, the other terms in the above expression can be calculated by taking the typical values of dark junction currents and the input bias current of the op-amp. Summing the calculated terms (not including IG – R1 and IG – R2) gives noise levels below 1.1 fAHz 1/2. They are distinctly lower than the experimentally obtained total noise currents (e.g., ItB = 1.9 fA Hz 1/2). Such differences may be explained by assum-
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Fig. 10. Noise current ItB as a function of optical signal intensity.
ing that G – R noise may have a major contribution, because it is not included in the calculations. In order to confirm this assumption, we have observed the dependence of noise on signal intensity. Taking channel B as an example, we can see from Fig. 10 that the measured noise current ItB (in fA Hz 1/2 at 1 Hz) has a steady increase with signal intensity, which indicates the preponderance of noise generators in the detector. Furthermore, the noise level seems to have a linear intensity-dependence, which accounts for the presence of dominant G – R noise. 3.5. Minimum detectable signal By using the synchronous detection technique, we can recover a signal buried in noise. According to Eq. (4), provided that the output bandwidth of the lock-in amplifier is much narrower than its input bandwidth, i.e., BobBi, a signal in the case where SNRib1 is still detectable. This condition can be met by increasing the time constant of the instrument sc, which is inversely proportional to Bo. This will lead to sensitivity improvement at the expense of measurement rate. Experimentally, we have obtained a minimum detectable optical signal of 9.6 fW with sc = 1 s, and 3.8 fW with sc = 5 s.
4. Conclusion
Fig. 9. Schematic diagram of the CMOS BDJ detector for noise analysis [4].
We have estimated the sensitivity of the CMOS optical BDJ detector, which is a key performance aspect for its applications to biochemical analysis. The detector sensitivity is closely related to noise effects, and this performance will deteriorate if the coupled measuring instruments or associated electronics have major noise contributions. To minimise additional noise, we have combined low-noise preamplifiers with the detector. This has led to the sensitivity estimation much closer to the achievable performance of the device. We have thus evaluated the detectivity of the BDJ detector, which depends on its operation and the use of its two outputs (which provide, respectively, two currents, I1
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and (I1 + I2)). For photodetection with the use of (I1 + I2) as the detector response, the corresponding detectivity is 2.4 1012 cm Hz1/2 W 1. When choosing the other output signal I1, or using both outputs for wavelength detection, the obtained detectivity is 1.9 1012 cm Hz1/2 W 1. The measured noise seems to be mainly contributed by the G – R noise sources of the detector. This assumption is confirmed by observation of its intensity-dependence. We have also used synchronous detection for weak signal recovery. At a reasonable measurement rate, the minimum detectable optical signal is lower than 10 fW (10 14 W).
[3]
[4]
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[7]
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