Trends in Electrical Drives for Low Cost Automation

Copyright © IFAC Low Cost Automation 1989 Milan , italy, 19 89

TRENDS IN ELECTRICAL DRIVES FOR LOW COST AUTOMATION D. Schroeder Institute for Electrical Drives, Technical University of Munchen Arcisstaresse 21, D 8000 Munchen, F.R.G.

Abstract . Electrical drives - controlled in torque , speed and position - had been dc drives in the past. These drives have three major components : the electrical machine , the power converter and the signal processing for the control . In all three areas occured major changes , thus meanwhile a variety of solutions are available even for low power drives at a reasonable cost level. Nowadays ac- machines tend to replace dc - machines . This trend is supported by a dramatic development of new power semi conductors which can be switched off , too. Moreover new techniques are available to simplify the mechanical construction of the converters , thus reducing space and weight. Furthermore new converter topologies are in discussion , enabling higher switching frequencies and reducing the harmonics. The signal processing has shifted from analogue to digital, given rise to the implementation of more complex control strategies on the one hand, but decoupling hard- and software production on the other hand . These trends will be described in detail in this paper. Keywords . dc - motors; ac - motors; drives - control; digital contro~i power suppl y _

INTRODUcrION Electrical drives are a main component in different industria l- and home applications. Up to 60% of the e l ectrica l energy produced is transformed into mechanica l energy in industr ialized countries . The greater portion of this energy is used for drives working directly from the mains without any energy conversion of the electrical energy and without contro l . The other part of the drives are con trolled drives with electrical energy conversion. This part will increase rapidly in the future due to different aspects as energy saving , higher or changing demands in technologies or new applica tions .

speed

contro l ler

current l Im i tatIon

If we concentrate on controlled drives we must con sider different aspects which influenced the development. The first aspect are the new power semiconductors, the second are new converte r and inverter topologies, the third new control strate gies, the fourth the introduction of the microcomputer and digital signal processing and last but not least the general system approach to design and optimize the whole system. I n the following chapters the trends in the dif ferent areas will be shown and how they in fluenced each other.

cur rent contro l le r

1
four Quadrant converter

flrlnl) atl9 1e llr.l l tatlon

Fig . 1 Standard solution for dc - drives bridge had to take into consideration different requiremen t s as electrical insulation of the thyri stors against each other and thermal aspects. Furthermore due to the topology of the bridge har monics and reactive power had to be accepted resulting in noncontinuous current at the dc - side on the one hand and on the other hand in distor tions and a low power factor on the ac - side. Another time - and cost-consuming factor was the putting into operation.

STATE OF THE ART AT THE BEGINNING OF 80th

The situation was much more complicated in the field of ac - drives. Due to the fact that only t he thyristor was available, a wide variety of dif ferent solutions for different applications and ac - motors were used . It wou l d be too ambitious to show all the lines of the development of ac drives during these times and all the efforts of the engineers. Only the most important so l utions with some influence up to now should be ment i oned.

At the beginning of the development the thyristor was the standard component of the converte r and inverter circuits. For example the standard solution for dc - drives was a four quadrant bridge converter without circulating current and cascaded control for current, speed and position (fig . 1). This solution had some advantages , for example it was simple i n the design , not to costly and the control strategies were well known. Of course there are some disadvantages, too. Due to the discrete component thyristor the mechanical design of the

A very simple and low cost- solution was t he subsynchronous cascade (fig . 2 ) , which is used for drives with quadratic torque characteristic and a

317

D. Schroeder

31X

restricted ra nge of ope r ati on of the speed as fo r example pumps or test sta nds for combustion engines.

LI--H

L2--~

ASM SDeeo

sreed

current

(ur l ent

refe rence

cont roller

reference

controller

L3--~

Fig . 2 Subsynch r onous cascade Fig. 4 The advantage of t.his solution is the limited ratings of the diode bridge and the natural commutated inverter , limited in respect to the limited operation range . The disadvan t ages are for example the s liprings of the mac hine, the starting equipment , the harmon i cs in the ma in s whi ch produ c e

flicker at some operat i ng points and the power factor of the inverter. To avoid the drawbacks of these solutions a va riet y of othe r inverter topologies we r e p r oposed f o r the

Ac -actuator

This solution is used up to now but its application is restric ted t o l o w power drives due t o the losses in the ac -machine and fo r low performance. A much bet ter solution which is used in a wide range of applications up to now is the curre nt source inverter for ac - drives (fig . 5).

subsynchronous solution .

10

Anothe r solution with naturally commutated conve r ters is the cycloconver ter. Thi s solution wa s

accepted ve ry often , because the bridge converte r was a solution well known from the dc - drives. Th e r e is a ve r y wide variety of solutions whi ch are shown in / 1/ consider in g only topologies f or

mains 11

11- - -+--+

Id ,Vd

sidering the topology of th e i nverter and the ac mach ine again to reduce costs . But the general disadvantage of the cycloconverter limited frequency

vOflob ~1

variable

the inverter to reduce costs or for example - con -

LJ-- - + - - t - - i

range, low power factor and harmonics in the mains remained.

A solut i on used up to now for medium and high powe r drives is the synChronou s machine drive (f~g. 3).

K fmsp~ls ung .

n~f19~fuhrf~r Stromfl(hf~r

und

6ldttufll}sdross~1

Lo

cur rent source: l i ne cOITITlutated converter and choke

Ulbsf9~(uhrf~ Stromn,hf~r

sel fCOIlmutated Inverter

drehl0hlvo rlObl~

ASYfKhronmoschm.

SDeedcont ro ll ed Induc t Ion machine

ma ins

Fig. 5 Cur rent source inverter for ac-drives

11

" 11--+-+-,<

current source

load (O'l1TIutated

Inverter

Fig. 3 Synchronous machine drive The solution shown in fig.

This so lutio n consists of a controDable current source (s ix- pulse - thyristor bridge and choke) and the selfcommutated inverter with very simple and r obust selfcommutating circui ts. The advantages of this solution are four quadrant operation of t he drive , very r obust behaviour . Furthermo re very simple control strategies can be used, if it is desired - so this so lution is up to now a r eal low cost solution.

3 or modified solutions

with twelve - pulse load commutated inverters and two three phase winding - systems with 30 0 phase shift are a common solution up to now . The disad -

vantages of this solution are the requir ement to use synchronous machines (b rushless e xcitation) , the restricti ons of the dynamic behaviour at zero speed and the harnonics and the powe r factor in the

mains .

Therefore the engineeri ng efforts conc entrated on the development of ac - drives. The problem at the beginning of this development was the lack of ele ments that can be s~itched on and of f. Therefore selfcornmutat i r:g topologies \o;e~ssary , but these selfcorr~utating topologies had been very costly . The most si~ple solution - wit hout using selfcommutating circuits - for ac - drives is the

But the most desired solution wa s the voltage source inverter (fig. 6 ) because it was belie ved that this solution had no limita tions - except the cost. Th e main disadvantages of this system in the past years were the relati vely high expenditure in the commutation circuits or if the cost in these cir cuits was reduced the limitations in the s witching frequency , the high switching losses , the sensiti vi t y against variation of the dc - vol tage, the sensitivity aga i ns t commutation failures - a common p r ob l em of all voltage source inverters - and the high cost fo r the information processing - on the one side fo r the inverter itself and on the other side for the ac-machine .

TRENDS IN ELECfRICAL DRIVES FOR LOW COST AUTOMATION

319

C • catho<1e

G • gat e

Sym b ol

ma ins

LT

M +--_,,3-

L2---+---4 L3----1-+_~

uN,r N

const.

H od~11

--------------

A • ano<1e

mod.1

se l fcolilutated

voltage source

Inverter

Fig. 8 Principle structure of a GTO (~ate Turn Off thyristor)

Fig . 6 General vOltage source inverter

r-+----~ R

IMlvlC:ual turnJff Circuit

structure so that the npn+ - transistor has a gain cC. npn ~l. Therefore the positive feedback condi t~on oc. pnp . o£,npn ~ 1 in the two tra n s i sto r structure ~s ~nfluenced mainly by t he npn - transistor at the gate - cathode side and t h u s the GTO can be swi~ ched off by a negative gate current . But the = I /I ~ 3 . .. 5 so a high negative gate g&irentG~§e Anodenecessa ; y /2/- /7/. The range of these elements started f r om elements 1200 V at the beginning, so that direct coupling to the mains was possible. Now these e l e ments are used mainly in the high power range . For low power applications other elements start ing from the unipolar or bipolar transistors are mo r e common.

r,s

b

]IE '::9H-]3I C

d

e

le<;J tL;rn off circuits

Power - fie id- effect transistors are vertical fie l d effect-transistors (FET) (fig . 9).

Go~ ""~

1 ~,. Sym bol

cormor. :urn off c!rC':..llt

Fig.

7 Different commutation circuits

Thus voltage source inverters had been only a limi -

ted application range in the past, due to the situation described. This changed dramatically in the last years . TRENDS IN THE FIELD OF POWER ELECTRONICS In the 80th a series of new power semiconductors were developed. Some of them disappeared meanwhile againl other elements had been introduced success -

fully. There is the c lass of elements which started from the design of the well known thyristors. These elements had been the ASCR (asymmetrical thyristor ) and the RCT (an ASCR with integrated antiparallel diode). These elements used during optimization of the date and the design the effect, that these elements should not block negative voltages. But inverters with these elements required the discrete commutation circuits too, thus these elements were not very successfull . A standard element starting from the thyristor is the GTO (fig. 8). The GTO has a NPNP structure as the thyristor a nd can be s . .·:itched cn by a positive gate current impulse as a thyristor. But due to the anode shorts the co(. of the anode side pnp - transistor is low. pnp Furthermore the gate - cathode side has a fine

Fig . 9 Vertical FET, type SIPMOS The power-FET ' s can be switched on by a pos i tive gate - source voltage and off by a zero or small negative gate - source voltage. The switching speed depends from the loading and de loading of the gate source capacitor and the drain - gate - capacitor , transients of U in 50ns are possible, so the power - FET is a ~~ry fast switching element, a l lowi n g high switching frequencies without expensive snubber - circuits. Therefore the gate circuit for FET ' s must load capacities only , this results i n a low power requirements if the switching f r eque n cy is not too high. Furthermore vertical FET ' s possess an inherent antiparallel diode N+N- P+ this is ad vantegeous for voltage - source inverters, if the dynamic characteristics of this diode are adequa t e. The essential drawback of FET ' s is the high on-resistance relatively to the blocking voltage / 8/ - /1 1!. Paralleling of FET ' s at high switching frequencies must be considered carefully . Another field - effect - transistor is the SIT (Static Induct~on !ransistor). This is a short - channel - fet

D. Schroeder

320

with a triode characteristic, which is normally on.

Another hybrid element

is the MCT /20/ - /2 1/.

Fig. 10 shows the structure of a bipolar power transistor. Similar to the GTO modern bipolar transistors have finger structures for emitter and base / 12 / .

.,-

Symbol

B~C"""''' baN

E -"er

E .. 6111 tter

Al-

(I,'''rod.

.,..,...

Symbol

~

f------7.-----\~AI

~C

c: : "

I"

p+

R,.

W. Plall,

B • baN

C ., collector E

A A/lODE

anode

.""rt..

Fig. 12 Structure of a Static Induction Thyristor SITh Fig.

10 Structure of a bipolar power transistor

The advantage of the bipolar - transistor compared to the FET is its lower on resistance, thus bipolar transistor can carry much higher currents, and to some extend higher blocking voltage can be realized. The disadvantages are the continuous base current during the on-time, the relatively small collectorbase gain when the blocking voltage is high and the dynamic behaviour . Another solution are transistors which avoid the parasitic two-dimensional effects / 13 /. The most interesting trend is the combination of unipolar and bipolar devices to gain a combination of the advantages of both: simple gate circuit, high blocking voltage, low on-resistance, high collector currents and good dynamic behaviour. From fig. 11 it can be derived easily, that for example the IGBT (Isolated Gate Bipolar Transisto r ) is such a hybrid pOwer switch. The structure of the IGBT is at the upperside in fig. 11 similar to a vertical FET in fig. 9. Additionally a p+layer is added at the drainside of the FET / 14 / - /11/ .

The following list gives a general overview, the paper /22/ and fig. 14 FET

IGBT

GTO

BJT

5000 V

1200 V

1000 V

current

3000 A

300 A

200 A

10 A

t

3-5"s

2"s

200ns

l OOns

blocking voltage

1000 V

max.

on 10 -25"s toff switch. 1kHz frequency

l"s

0 , 3- 0,5 "s

20kHz

50kHz

600 V

500 V

15-25"s 2 -5kHz

typical parameters blocking voltage

1200 V

800 V

max. current

600 A

100 A

40 A

20 A

r,;!~~S~ncy

1kHz

3kHz

20kHz

20kHz

t.

Thyrtstors for Dhase control GTO-thyrtstor

--------, I

I I

Symbol sylllbol.

1

Fast-thyrlstor : "--"---'.1

1 10 ... p+ IAV-

collector cathode v~

:: fort'iarC

of~-state

voltage

IAV :: maxl il'H..tn average on -state current Fig.

f = switch ing fr eQue ncy

11 Structure of an IGBT

A similar approach was used for another hybrid element, the SITh (Static Inducti on Thyristorl/18/. A FCT (Field Controlled Thyristor) /19/ is an element as-the S1Th. The difference is the mesastructure of the gate-cathode region, which results in a different behaviour.

Fig.

14 Maximum ratings of power semiconductors (Courtesy ABB)

TRENDS IN ELECfRICAL DRIVES FOR LOW COST AUTOMATION

321

Pa r allel to the development of the power semiconductor devices was the development of the power modules . Power modules are combinations of d i ffe rent power devices - for example switch i ng element

and antiparallel diode or a full bridge - which are insulated electrical l y against each other in the mo dul but could use the same heatsink . By th i s technique t he mechanical construction of the power circuit was simplified very much, reducing costs ,

volume and weight too. A further development in this direction a re smar t

devices . These devices include not only the power switches but the gate circuit, overcurrent detec -

tion and other features / 23/ . TRENDS IN CONVERTER AND INVERTER TOPOLOGIES With the availability of power semiconductors, which can be switched on and off , all converterand inverter topologies with selfcommutating re sonant cicuits we~obsolete for general app l ica tions. Due to this situation two trends can be

detected. The first trend was to use these elements in a

" hard swit c hed " mode , thus most of the o l d topo l o g i es can be used with small modifications . The second trend was to find new topologies and to avoid or reduce the switching losses in semicon ductors. One objective is an increase i n the swit -

Fig. 16 Inte grated cont r o l un it ( 120x 60x 90 mm ) f o r dc - drives i n c luding t h e powe r c ir c uit and the con tro l ci r cuit s (Courtes y ABB )

ching frequency above 20 kHz to reduce the audi b l e noise . If we consider dc - drives, fig . 15 shows a very simp l e and low cost solution, which nowadays i s

used often .

Lt

L2 L 3 ---j-

f--i

~~

vol taQe

brack lo;

sour(~

-$1 L

Fig.

bri dge

res i stor

J

i ll

~

l )J

15 Mo d e rn power circuit for dc - drives

The di ode bridge pro duce a dc vo ltage , which is con s tan t . By swit c hing the transist o rs crosswise ( for example T) a ndT o n o r o ff; T2 and T4 off always 3 fo r o n e d lre c tl o n o f r o tatl onl

the armature voltage

c a n b e varie d by p ulse width modulation. If current co ntro l i s the most inner c ontrol loop this holds f o r the t o rque , too. By integrating the diode bridge into on e mo dul e and the H- bridge for the dc - motor into a second modu l e , th i s results in a very smal l po wer ci rcui t . The further i ntegration of both

mod u les in o ne mod u le and in the next step of the co ntrol t oo will result in even smaller units.

Fig. 17 Power module

(Courtesy ABB)

Disadvantages of the solution shown in fig.

15 are the b r acking mode and the "hard - switching " o f t h e

transistors. If bracking occurs the ene r gy is fed back to the dc - link capacitor C and the dc - vo ltage will increase , because the energy can ' t be re covered by the mains due to the diode br i dge . The r e f o r e

Fig. 16 s h o ws s uc h a comp ac t unit which includes two thyr i s to rs - mod u les, f i ve d iode - modules and the

a bracking resistor with another swit c h i s neces -

who le con tr ol f o r a dc - moto r up t o an armature cur 36 A. ren t fo r lA

exceeds a maximum limit . The "hard switching" of the t r ansis t or s

Fi g .

drawback , because during the on - and o f f - swi t c h ing , switching losses are produced.

1 7 s ho ws a po we r mod ule with lGBT, which is

sary, this switch is activa t ed, if t he d c - vo lta ge is anoth e r

used whe n higher ratings are con sidered. "Hard switching " means the fo l low i ng s ituation .

When transistor TI fig . 15 the diodes carry the armature the transistors TI

and T3 are switched off i n of the switches T2 and T 4 must current of the dc- machi ne . Whe n and T3 are swi tched on f o r the

D. Schroeder

322

n e x t switching cycle of the pulse - width- mod u lation , the transistors carry the full dc-voltage before switching on, because the diodes of swi t ch T and T4 carry the armature current. Furthermore the transistors Tl and T are switched on, they must carry a short circui~ current for a limited time due to the recovery effect of the diodes , which carried the current before. After the recovery ef fect of the diodes the transistors carry the armature current. Thus the t ransistors were stressed very much during the switching process . A similar situation occurs during the switching off transient . This is a general drawback of voltage source in verters (for dc - and dc - drives) . Mo r eover these systems are very sensitive against misgating , if for e x ample the transistors Tl and T4 are switched on simultaneously by a mistake.

i!

because th e y a r e designed for high switchi ng speed and s mal l snubbers. If an asynchronous machine is considered , different solutions are available. F i g . 19 shows a simple solut i on with a variable dc voltage U . d

lhl -CVJ Mai ns

Lt

c

L2

The losses produced during the switch i ng process limits the maximum s witching frequency - for examp l e less than 5 kHz for bipolar trans i stors normally. This disadvantage can be avoided if MOSFET or IGBT were used. But the list and the graphic in the l ast chapter show that the achievable rated power of the IGBT is smaller tha n of a bipolar transistor a n d the achievable rated power with MOS FET switches is lower than the rated power with an IGBT. A well - known solution to reduce the specific swit ching losses in power semiconductors is the use of special snubber circuits /24/ - /27/. T h ese special snubbers circuits reduce the d~dt during switching off, limit the maximum dynamic b l ocki n g voltage and avoid high di/dt during swit ching on. Furthermore the energy stored in the snubber circuit is recovered . A typical example of such a snubber circuit shows fig. 18.

'" l,

•

r;~ !I:

ud

D., c,

~

R

c..

-' ""

R

'"

l3--j'-+-4 o,..f N cons t .

-------vorioblt! voiiag~ 'sourct! Ud

Vd" UdioCOsa

Fig . 19 Voltage source inverter with PAM The dc - voltage U of the thristor - bridge with na d tural commu ta t ion is controlled by the firing ang l es ~ , this producing a variable link vo ltage U . The mach i ne side inverter is controlled wi t h tge fundame n tal f requency on l y , so the switch ing frequency of the inverter and the output fre quency fl are the Same. The ampl i tude of U i s control l ed by the dc - voltage U , therefore l this d procedure i s called pu l se amplltude modulat i on (PAM) . So this solution avoids high switching fre quencies for t h e inverter. The draw backs of t h is solut i on are the relatively low dynamic response o f the li n eside thyristor - bridge and no rege n e r a tive bracking . The wellknown draw backs of cont ro lled t h y r istor - bridges - reactive power and harmo nics at the l ineside - has to be accepted . Regenerative bracking can be achieved by an add itional antipara l lel thyristor bridge (four qua d r ant br i dge). Another solution is shown in fig. 20 .

-

~ L mains

1. SlnQle snllt>ber caoacito r ' 1JX lllary volu~ source

~n d

2. Grourlded aux il iary voltage source. Su1tabl e for lOOdul e aoo llcatl on

,. Svmretrl ctll ultt.-hl~

a rran~n t for PON!r .ooll cltlon

.-.i

11

l2 L 3 --t--t----t

v"

r L

UN, f H

Fig.

18 GTO snubber circult

(Co urtesy Marquardt )

const.

Vd'

Such snubber circuits are very effective, the swit ching frequency for example of bipolar transistor can be increased from maximum 5 kHz to 100 kHz , if such circuits are used /28 / . This result is very valuable , because the dynamic response of such systems compared to systems without such snubber circuits can be much better. Furthermore the harmonics in the l o ad are reduced , so the discontinuous current range is reduced and thus the loss of the dynamic response in the dis continuous current range is avoided / 29 / . Moreover additiona l losses in the dc - machine are reduced. A draw back of these special snubber circuits are the additional costs, so one design object i ve for power semiconductor manufacturer is to avoid these snubber circuits (s nubber less power semiconductors). The hybrid power semiconductors - as the IGBT , FCT and the SITh - are another possible solution ,

U,. ',

variQb/~

consf

chopper

Ud ]

vQnab/~

Fig. 20 Inverter with intermediate circuit chopper The variable dc - link - voltage is produced by an termediate circuit chopper , thus the amplitude of the output voltage of the machineside inverter is controlled by U again. The output frequency fl d2 is controlled by the switching frequency of the inverter switching frequency again (f undamental frequency ) . This solution avoids the reactive power in the mains nearly , because a diode bridge has a firing angle IX. Ill: O. The most common solution for ac - drives had been shown i n fig . 6 in the last chapter . But now we can use the different power semiconductors which can be s Wl tched on and off, thus avoiding the additional

TRENDS IN ELEcrRICAL DRIVES FOR LOW COST AUTOMAnON r esonant circuits for selfcommutation shown in fig. 7 . The dc - link voltage is constant now, be cause we use a diode b r idge at the mains side , so regenerative bracking is not possible and a bracki ng chopper is necessary if bracking occurs. The amplitude U and frequency f1 are controlled by 1 pulse - width - modulating (PWM). A standard solution for PWM is the synchronized PWM ,where a sinussoidal modul ating wave and a synchronized triangul ar car r ier wave are compared to fix at the intersections the switching instants of the six switches of the i n verter in fig . 6. The amplitude of the output vo l tage of the inverter is varied by changing the amplitude relation between the carrier s i gnal and modulating wave. The PWM procedure was not straigh t forward in the past , because of the restricted swi t ching frequency of the inverter which had not exceeded 400 Hz normally. This restriction resulted in different pulse pattern i n different operati ng points of the ac - machine . When the pulse patte r n were switched undesired transients occured . Due to the much higher switching frequencies of low power devices this can be avoided. The pulse generation is achieved now with single chip hardware solutions /30/ or on - line optimized pulse pattern. The voltage source of the dc link can be used for different inverters , when a solution shown in fig . 6 is used , so the costs are reduced a g ain . A very common idea is to use the advantages of the solution in fig . 19 and fig . 6. The advantage of the solution in fig . 6 was the low harmonic content, if t h e switching frequency in the inverter is high , but this results in high switching losses in the power - semiconductors. The advantage of the solution i n fig. 19 were the low switching losses, because t h e fundamental switching frequency is used only , but the harmonics are high compared to the PWM so l ution . So a general idea is to use the solution s h own in fig . 19 and to use PAM at higher dc - link vo l tages and PWM with a constant dc - link vo l tage be l ow a certain operating point - for example 10% of t h e nominal rating .

323

I f a better dynamic response is necessa r y decoup ling control /31/ or field o r iented control can be used. Up to now we cons i dered frequency changers without ene r gy recovery only. But the utility companies are sensitive against the pollution of the mains with reactive power and harmonics meanwhile. A general object i ves are consumer with cos = 1 and even during t ransient operat i on. In f i g . 22 a solution for voltage source i nve rters /32/ - /35/ and in fig . 23 for current source inver ters /36/ - 37/ is shown. All these solutio n s use " hard switched " e l ements .

'I

A,..l ,

Mains

L1 ---------1 L 2'---_--j---1

L3' -~"--r----+----i

------------voltage source

machine side inv erter

Fig. 22 Structure of vol t age source inverter with cos 1 and ,{JI< l .

Y=

L1---__"""1"-"""1"--4 L2-~

__~~-+~

L3--~"~~--t--+~

Mains

The higher switching frequency in the inverter re su l ts in another advantage, if PWM is used . The higher the switching frequency the l ower are the harmonics in the current at a given load inductivity. A lower load inductivity in the ac - machine results in a higher ma x imum torque and a lower speed variation during load . This effect had been used in the past in current source inverters too to avoid speed sensors. Meanwhile this simple control strategy is used for VOltage source inverters too , fig . 21 shows the signal flow graph o f this solution. The realization can be analogue or digital.

f

mains

occelerafion

ret.

v~ odd.

odd.

re'

ret

vo/u ~

K1/u ~

Fig. 21 Signalflow graph of a sensorless control for induction machines

Lin e- side inverter

Load - side inver ter

voltages and currents are Slnusso lda l

Fig. 23 Structure of a current source inverter with cos 1 and 1.

Y=

itc:

The result of these developments is a remarkable reduction in volume weight and cost. For e x ample the volume and costs of drives reduced from 100% in 1960 to 4 0% in 197 0 to 14 % in 1980 and will be reduced to 8% in 1990 for power sources of 100 W; the volume of the converters for dc-drives reduced f r om 100% in 1975 to 30% in 1980 to 12% in 1989, the costs from 100% to 70, in 1989. for ac - drives the following results hold 100%/ 4 0 %/ 10% for the volume and 100%/ 95% /70% for the costs. This holds for dc and ac- drives with the rated power from 10 - 20 kW . A different concept for drive applications are quasi- , multi - and distinct - resonant topologies for converters and inverters. Such topologies had been investigated for power supplies :or computers / 38/ - / 40/ f o r example. Another approach had been used ~or res onant l1nk inverters ,' 41 / . With ~he new pov.er se::J.iconductors hard switching can be avoided. Fig . 24 shov;s a zero - voltage switch (ZVS ) and a zero current s~itch (ZCS;. With these resonant switches the specific switching losses can be reduced very much, thus very high switching frequencies are possible. Fig. 25 shows the structure of a resonant inverter , the switching

324

D. Schroeder

$

"11 .......

now control strategies were used which used appro-

ximations for the highly nonlinear dynamic behaviour of converters and inverters. The general idea

"

"

of these new approaches is to generate control strategies which take into consideration the nonlinear dynamic behaviour without approximations

Fig. 24 Structure of ZVS- and ZCS-switches

/53/-/55/. Fig. 26 shows the control structure of a predictive control strategy for dc-drives /54/. The basic idea is a precalculation of the desired current trajectory for the next switching state. By comparing this precalculated trajectory with the real current, the next optimal switching instant is determined.

1-+-+lI--++---<> R H-++--<> 5

I

II

IrfColculolfd

t

(\Trent

'p',lIt

I ~,~ ~-------B

II

___ l _______J

Fig. 25 Resonant inverter topology frequency of the GTO's is 20 kHZ, this is a remarkable result /42/.

Fig. 26 Structure of the precalculating control strategy

But the design of these very complex topologies is really complicated, especially if high switching frequencies of 100 kHz and more should be achieved and if all the parasitic effects are considered. Up to now there are no closed solutions to design these topologies, so CAE is a very helpful tool. This tool will gain more impcrtance in the future to avoid bredboard experiments which are costly and time consuming. But the models of these modern

The result of this strategy is the best dynamic response (in only one switching state normally) and the highest possible cross-over-frequency for superimposed control loops. The same strategy can be used for ac-drives, too. The cost function for the control strategy consi-

ders lowest possible switching frequency during steady state (to reduce switching losses) and best possible dynamic response /43/-/46/; /55/.

power semiconductors are just under development,

so this tool will be available soon /43/-/47 / .

A practical result shows the fig.

27. Fig. 27 shows

a reversion of the power flow in the mains from There are many other ideas under investigation, so new topologies with new characteristics and new

positive rated torque to negative rated torque. The system has one GTO-inverter at the lineside and

control strategies can change the situation in the near future very much. Further activities are in the field of electrical machines for example electronic commutated dc-

another inverter at the machineside.

machines, permanent-magnet synchronous

machines or

reluctance-machines /48 ' . A far topic are HDmachines or machines with high-temperature superconducting windings, but may be this is not so far

eXp"erimenlal resulls transient QP..-;'ralion: reversion 01 power Ilow \uz=6S0V. HYS=20%. LWIS mHo fp

=1 kHZ\

line volfag!t, (phase a)

as we believe nowadays.

TRENDS IN THE CONTROL OF DRIVES At the end of the 70th the cascade, adaptive control of dc-drives was common, it's realization was

analoguous normally /49/. Furthermore the theory for the control for ac-drives was established /50/, /51/, /31/. But a general industrial application of these theories for ac-motors - especially for low power drives - failed to appear. The microcomputer and its digital signal processing changed the situation. Two trends could be observed. The first trend was to implement the well-known strategies for dc- and ac-drives into a microcomputer / 52/. Thus the specific advantages and disadvantages of microcomputer generations and the digital signal processing in combination with different sensors

could be studied. The industry followed this approach soon, because the production of the hardand the software was decoupled, an impcrtant advantage. But a much more important feature of the microcomputer is the ability to implement novel openand closed-loop control strategies. A typical example for this approach are the "on-line optimized pulse pattern" for dc- and ac-drives. Up to

Fig. 27 Experimental results at the l ineside inverter of an ac-machine drive The maximum switchi n g frequency was set to 1 kHz,

the pulse pattern was on-line o ptimized . From the results in the three phases the e xcel l ent stati c and dynamic respo nse can be seen. The strategy works well, even if the u tility vol tage i s disto rted very much. Another approach fix the switching frequency t o a desired sampling frequency / 56/ . All the se strate-

1RENDS IN ELECfRlCAL DRNES FOR LOW COST AUTOMATION

325

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Hicrocomputer Fig. 28 Control in field coordinates by microcomputer for an ac - servo drive gies are not used in industry up to now . There is a long and fruitful discussion about the superimposed control loops for speed and position for dc- and ac- drives . So it seems not very useful to discuss it here again. It is assumed that in the references discussed in the last chapters all aspects of the control of these loops are considered.

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Fig. 28 shows the typical structure of the control for an ac - servodrive with current source inverter . But there are other activities which had not been mentioned yet. One activity is on-line parameter identification . This on- line parameter identifica t ion was not so important , when cascaded control with PI-contro llers were used. But for modern cont r ol concepts parameter identification is essential. Therefore research concentrated on on-l ine para meter adaption for ac-machines, because the well known theories / 5J/, /5 1/ , /31/ failed , whe n the parameters for the estimation of the orientation of the rotor- or stator-flux was unknown. There are two important parameters if we co n side r to use a rotor-flux oriented coordinate system , this is the rotor resistance R2 and the inductivi ty M. R2 will vary because of thermal heating and M will vary if field weaking is used. There is a wide variety of different suggestions /57/- , 59 / . At the beginning models of the acmachine were used, the parameter sensitivity was investigated and suitable models were chosen. In the next step of the research observer were used , which had been designed to be robust against the parameter variation. These observers had been full or reduced observers . Other approaches used the model reference adaptive control strategy (MRAC). These research activities show very clearly that now very modern control strategies were used not only to estimate the parameters on- line but furthermore to put the drive into operation auto matically. This is up to now a time - and costconsuming activity for many engineers. But in the near future this will be done by the control unit itself or by an additional unit during the putting into-operation - time / 6 0/ - / 61 / . Furthermore the wellknown control strategies will be counter - checked against modern control strategies like optimal control , robust control , active adaptive control, passive adaptive control, learning (repetitive ) and intelligent (expert) control. A very g o od o verview are given in / 62 / and / 6 3/ . Fig . 29 shows the principle structure of the modern control theory. Another trend is the fault diagnosis for complex systems. This development has very strong relations to the parameter estimation and will be very

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Fig. 29 Structure of modern control theo r y important in the near future. An overview i s given in the papers / 64/ and / 65/. All these ac ti v i ties are highly complex and need a deep knowledge of the control theory on the one hand and hard - and software solutions on the other hand.

Up to now we discussed the drive as a system alone. Bu t a drive is never used as a stand alone unit, a drive or drives are used in an environment to produce. Thus the drive is onl y a compon ent in a mechanical system and the production process wi l l influence by the mechanical system and the cont rol the overall system behaviour and thus the rating, the design and the control of the drive, too. The consideration of this general system approach is another very important aspect in the industrial future. For example we must consider that the con trol of a two mass system with one sensor at the drive side , elastic coupling and two nonlinearities (backlash and friction ) is an unsolved control problem up to now . If we consider production processes , the situation is much more complex and difficult to solve. In / 66 / -/70/ a few examples are shown.

D. Schroeder

326 CONCLUSION

Electrical drives are regarded by many people as a classical subject, where few, if any, innovations take place. It has been the aim of this paper to show that in the last ten years a remarkable development started. This development took place in all components of a drive: the electrical machine, the converters and inverters and their power semiconductors, signal processing hardware (mic r ocomputer, ASIC .. ) , the tools to design the whole systems or parts and the theory to design the control. All these developments are in full progress now. So many challenging problems are waiting for the engineers in the future . REFERENCES / 1/ /2/ /3/ /4/

/5/ /6/

/7/ /8/

/9/

/ 10 /

/ 11 / / 1 2/

/ 1 3/

/14/ / 15 /

/ 16 /

/ 17/

/ 18 /

/ 19 /

/20/ /21/

/22/

B . .R. Pelly. Thyristor phase controlled con verters and cycloconverters. Wiley N.Y., 1971. E.D. Wolley. Gate turn off in pnpn- devices. IEEE Trans . Electron. Devices , 1966 . p.590-597. M. Azuma; M. Kurata. GTO-thyristors, an overview. IEEE Pro ceedings 4 / 1988, p.419 -4 27. O. Hashimoto et.al. 4, 5 kV, 3 kA high power reverse conducting gate turn off thyristor. PESC 88 , p.915-920. T. Katsuo et .al. Design consideration for large current GTO (4,5 kV, 4 kA). PESC 88 , p.895-902. M. Kekura et.al. 8000 v, 1000 A Gate Turn off Thyristor with low o n-s tate voltage and low switching loss. PESC 88 , p.330-336. O. Hashimoto et.a l. Light triggered thyristor (6 kV, 2,5 kA). PESC 88 , p . 928-933 . M.J. Declerq; J.D. Plummer. Avalanche break down in high voltage D-MOSdevices . IEEE Elec. Devices 1976, p. 1-6. J. Tihanyi. A qualitative study of the dcperformance of S IPMOS -transistors. Siemens R&D report 1980 , p. 181-189. T. Saki; N. Murakami. Low reverse transfer capacitance VDMOS transistor. PESC 88 , p. 689 - 694. Y. Yoshida et . al . Low on-resistance and high reliability power MOSFET. PESC 88, p.674-680. H. Nishiumi et.al. High voltage , high power transistor modules for 440 Vac - line voltage inverter application. IPEC 83, p. 297 - 305. G. Muller; A. Porst; H. Strack , SIRET, a 1000V bipolar transistor with no two dimensional parasitic effects. Siemens R&D report 1988, No. 1. N. Zommer. The monolithic HV BIPMOS. Int. Electr. Devices 1981 , p. 263 - 266. H. Yilmoz et .al . Recent advances in Insulated Gate Bipolar Transistor Technology. IEEE / IAS 1986, p. 345 - 349. B.J. Baliga. Evolution in MOS Bipolar Power Semiconductor Technology. IEEE Proceedings 4 / 1988, p. 409-4 18 . G. Muller; J. Sack . A new concept for a non punch through IGBT with MOSFET like switching characteristic. PESC 89 , p. 21 - 25 . J . Nishizawa. Junction field effect Devices for power conditioning. Plenum Press . p. 241 - 270 . H. GrUning et. al. Properties of a high power field controlled thyristor. IEEE, IEDl1 1986, p. 110-11 3. V.A.K. Temple. MOS controlled thyristor. Int. Electr. Devices, 1984 , p. 282 - 285. M. Stossik; H. Strack. MOS-GTO turn off thyristor with MOS controlled emitter shorts. IEEE / IEDM 1985, p. 158- 161. P. L. Hower . Power semiccnductor Devices: an overview. IEEE proceedings, 4 / 1988, p. 335 - 342 .

/23/ J. Tijanyi. Smart Sipmos Technology. Siemens R&D report 1/1988 , p. 35 - 42. /2 4 / A. Boehringer; H. Knoll. "Transistorschalter im Bereich haher Leistungen und Frequenzen" , etz Bd. 100 (1979), H. 1 3 , p. 664 - 670. /25/ A. Boehringer . "Vereinfachte Einrichtung ohne prinzipbedingte Verluste zur Befreiung elektrischer oder elektronischer Einwegschalter von hoher Verlustleistungsbeanspruchung wahrend ihres Ein - und Ausscha l tens sowie van uberhohter Sperrspannungsbeanspruchung im AnschluB an ihr Ausschalten". OS H 01 H 33 / 16 DE 3132512. /26/ R. Marquardt. "Untersuchung von Stromrichter schaltungen mit GTO - Thyristoren ". Dissertation Universitat Hannover, 1982. /27/ T. Undeland; F. Jenset ; A. Steinbakk ; T. Rogne; M. Hernes. "A Snubber Configuration for Both Power Transistors and GTO PWM Inverters". IEEE PESC, Gaithersburg/Maryland, June 1984,p.42-53. /28/ R. WUrslin. Transistor-Converter Operati ng on 380V -Thre e -Phase-Mains . Proceedings of the "S econd Annual European Power Conversion Con ference" 1980, Powerconversion International, Oxnard , USA. / 29 / D. Schr6der. Adaptive control of systems with controlled converters. 5th IFAC Congr ess on Sensitivity, Adaptivity and Optimality. p . 335 - 342. /30/ E . Kiel et.al. PWM Gate Array for ac - drives. EPE 1987, p. 653 - 658 . /31/ W. Flugel; R. Weninger: Con trol of InverterFed Asynchronous Motors via Decoupling Networks. IFAC Lausanne, 1983, pp. 305-312. /32/ D . Schr6der et.al. High dynamic four - quadrant ac- motor drive with improved power factor and on -l ine optimized pulse pattern with PROMC. EPE 1985, p. 173 - 178. /33/ D. Schroder et.al. GTO - pulse inverters with on line optimized pulse pattern for current control. ICEM 1986, p. 668 - 671. /3 4 / D. Schroder et . al. Control of double vo ltage source inverter coupling a three phase mains with an ac - drive. IEEE/IAS 1987, p. 593-599. /35/ D. Schroder et.al. Four quadrant ac - motor drive with line - and machineside inverter and predictive control. EPE 1989 /36/ D. Schr6der et . al. Current source inverter with GTO-thyristors and sinusoidal motorcurrents. ICEM 1986, p. 772-776 . /37/ D. Schr6der et.al . Four quadrant ac-motor drive with a GTO current source inverter with low harmonics and on line optimized pulse pat tern. IPEC 1990 / 38 / F.C. Lee et.al. Resonant switches - topologies and characteristics PESC 1985, p. 106 - 116 . /3 9 / R. Steigerwald. A comparison of half bridge resonant converter topologies. IEEE Transac . on power electronics, 1988, p. 174 - 182. / 40/ F. C . Lee et.al. Zero - voltage-switching multi resonant technique, a novel approach to impro ve performance of high-frequency quasi - resonant converters. PESC 1988, p. 9 -1 7. / 41 / F. C. Schwarz. An improved method of resonant current pulse modulation for power converters. PESC 1975, p. 194-204. / 42 / D.M. Divan et.al. Discrete pulse modulation strategies for high - frequency inverter systems . PESC 1989, p . 10 1 3-1 0 2 0 . / 43 / D. Schr6der et. al. Computing the switching be taviour of ~vwer MOS-FET tc optimize the cir cuit design . IPEC 1983, p . 778 -789 . / 44 / D. Schr6der et.al. Modelling and simulation of power MOS - FET and power diodes. PESC 88 , p. 76 - 83 . / 45 / D. Schroder et.al . A bipolar junction transi stor model describing the static anc dynamiC behaviour. PESC 1989. pp . 314-321 / 46 / D. Schroder et.al. A unified model for power MOS-FET including the inverse diode and the parasitic bipolar transistor . EPE 1989.

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327