Two lossy integrator loop based current-mode electronically tunable universal filter employing only grounded capacitors

Microelectronics Journal 59 (2017) 1–9

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Two lossy integrator loop based current-mode electronically tunable universal filter employing only grounded capacitors

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Remzi Arslanalp, Erkan Yuce , Abdullah T. Tola Department of Electrical and Electronics Engineering, Pamukkale University, Kinikli Campus, Denizli 20160, Turkey

A R T I C L E I N F O

A BS T RAC T

Keywords: BJT Current-mode Second-order Universal filter

In this paper, a novel single-input and multiple-output second-order current-mode universal filter is proposed where lossy integrator blocks, scaling blocks and summing blocks are employed. The proposed filter can simultaneously generate all the five universal filter responses namely; low-pass, high-pass, band-pass, notch and all-pass responses without requiring any critical passive component matching conditions. It uses only two grounded capacitors as passive components and BJTs as active elements; thus, it can be classified as an active-C filter. Both resonance frequency and quality factor of the proposed filter can be tuned electronically. Approximately 1.5 decades adjustable input frequency range for all the universal filter responses and high quality factor for band-pass response are obtained. Obtained PSpice simulation results confirm the theoretical design. Non-ideality and Monte Carlo analyses are also given. A practical realization of the single lossy integrator is accomplished. All the presented results are discussed.

1. Introduction In the design of analog filters, interests in flexibility and functionality have been growing interest for the last decades. Simultaneous realization of five fundamental second-order universal filter responses and electronically controllability can provide some advantages such as adjusting sensitive calibration, flexibility, etc. In addition to these, current-mode (CM) signal processing circuits in which some parameters can be controlled by external currents have low power consumption, greater linearity, a smaller number of components and larger dynamic range, wider bandwidth when compared to voltagemode counterparts such as operational amplifiers [1]. In the related open literature, a number of electronically tunable second-order CM filters [2]-[39] have been reported. Electronically tunable second-order CM filters can be mainly divided into four categories. The first one is use of active building blocks (ABBs) [2][26] such as current controlled current conveyors (CCCIIs) [2]-[18], electronically tunable current followers (CFs) [19], current controlled current conveyor transconductance amplifiers (CCCCTAs) [20], current differencing transconductance amplifiers (CDTAs) [21,22], Z-copy current follower transconductance amplifiers (ZC-CFTAs) [23]-[25] and operational transconductance amplifiers (OTAs) [26]. The second one is usage of log domain filter [27]-[34]. However, differential classAB log domain second-order filters [27,28] use four capacitors which occupy large chip area in integrated circuit (IC) fabrication. Also, the

⁎

circuit of [28] can provide only notch filter response. Class A log domain filter [33] has multiple current inputs; thus, it requires extra circuitry. The third one is use of square-root domain filters [35]-[39]. Nonetheless, they need blocks consisting of tens of MOS transistors. The topology of [36] can realize only all-pass filter response while one of [37] is not universal filter. Also, the configuration of [38] can provide only fifth-order low-pass filter response. The fourth one is direct design method. However, the best knowledge of the authors, second-order universal filters with direct design techniques have not been developed so far. The previously published ABB based second-order CM filters [2][26] have the following drawbacks:

• • • • • • • •

Do not provide all the five second-order universal filter responses simultaneously [2,3,5]-[9,13]-[15,19]-[22,24]-[26]. Do not provide second-order universal filter responses [9,21]. Do not employ identical active devices [2,4–6,8,10]-[12,14][18,20,23,24,26]. Use floating capacitor(s) [5,13,15]. The use of capacitor number is not canonical [6,8,9]. Output current(s) are obtained with complex combination of input current signals [3,7,14,19,20,22,26]. Do not provide high output impedance responses [21]. Capacitor(s) are connected in series to X terminal of the ABB [5,6,8,9,13]; accordingly, high frequency performances of the filters

Corresponding author. E-mail addresses: [email protected] (R. Arslanalp), [email protected] (E. Yuce), [email protected] (A.T. Tola).

http://dx.doi.org/10.1016/j.mejo.2016.11.005 Received 3 August 2016; Received in revised form 10 November 2016; Accepted 13 November 2016 0026-2692/ © 2016 Elsevier Ltd. All rights reserved.

Microelectronics Journal 59 (2017) 1–9

R. Arslanalp et al.

• •

a linear relationship which is multiplying factor among control currents can be obtained by using translinear design procedure [28]. After routine analysis of the block diagram given in Fig. 1, the relationships among outputs can be derived as given in matrix equation below.

are limited [8]. Use of OTAs as active devices [26]; thus, high frequency performances of the filters are limited [40]. Do not provide orthogonal control of quality factor (Q) and angular resonance frequency (ω0) [2]-[5,11,13]-[15,19], [21,26].

⎡ Y1 ⎤ ⎡ 1 ⎢ Y ⎥ ⎢0 ⎢ 2⎥ ⎢ ⎢ Ylp ⎥ ⎢ 0 ⎢Y ⎥ ⎢ ⎢ hp ⎥=⎢ 0 ⎢Ybp ⎥ ⎢ ⎢ Y ⎥ ⎢0 ⎢ n ⎥ ⎢0 ⎣Yap ⎦ ⎢⎣ 0

On the other hand, some CM filters [41–56] with lack of electronic tunability have been reported in open literature. Also, the filter of [41] simultaneously realizing all the standard second-order CM filter responses cannot have the feature of orthogonal control of Q and ω0. The proposed second-order single input multiple output (SIMO) CM universal filter configuration has following important features: 1) It can realize all the second-order CM universal filter responses simultaneously namely low-pass (lp), high-pass (hp), band-pass (bp), all-pass (ap) and notch (n) responses without changing structure and/or without using additional elements, 2) It can provide high output impedance responses; thus, it can be easily cascaded with other CM circuits, 3) Its pole frequency and quality factor can be controlled electronically and orthogonally, 4) It possesses advantages of all the CM circuits, 5) It employs a minimum number of only grounded capacitors as passive components; accordingly, it is suitable for IC process [57], 6) It does not need any critical passive component matching constraints, 7) Its resonance frequency can be obtained by small-valued capacitors when compared to log domain filters [27–34], 8) It can provide high Q for band-pass response with low component spread.

Ylp U

U Ybp U

Yap U

Σ

_ U+

+

Σ

Y1

Y2

0

+

Σ

+

0

+

0

+

_

+

0

_

Σ

+

Σ

2

ω0 s + ω0

1 0 0 0

− K2 1 K3 − K3 0 1 0 1

0

s2

=

=

+

ω02 ω0 s Q

0 0 0 1 1

(1)

+ ω02

(2a)

+ ω02

(2b)

s2 s2 +

s2 +

ω0 s Q

ω0 s Q ω0 s+ Q

ω02

s2 − =

s2 +

ω0 s Q ω0 s Q

(2c)

(2d)

+ ω02 + ω02

⎛ ωω0 ⎞ Q ⎟ ϕ (ω) = −2 tan−1⎜⎜ 2 ω − ω 2 ⎟⎠ 0 ⎝

(2e)

(3)

It is observed from Eq. (3) that phase angle varies from 0° to −360° as the frequency goes from zero to infinity. For all the cases in equations (2), ω0 and Q depending on values of the DC current sources are given by following equations:

ω 0=

Q=

IE VT C (β +1)

1 1 = 2−K1 K3

(4a)

(4b)

Here, C is capacitor value, IE depicts DC current of BJT located in the integrator, VT is thermal voltage defined as VT=kT/q and β is current gain [59]. The variable k represents Boltzmann constant, q is unit charge which is equal to 1.6×10−19C, and T is temperature in Kelvin. Also, VT is approximately equal to 25 mV at room temperature. It is shown from equations (4) that both of ω0 and Q can be controlled orthogonally. For instance, by adjusting the DC current of integrator, ω0 can be changed without disturbing Q and vice versa. Therefore, electronically tunable property is achieved. Using a single pole model [60], β can be represented as

Ybp

3

0

⎡ ⎤ 0 0⎤ ⎢ U ⎥ ⎥ Y1 0 0⎥ ⎢ ⎥ Y ⎥⎢ 2 ⎥ 0 0 0 ⎥ ⎢ Ylp ⎥ ⎢ ⎥ 0 0 0 ⎥⎥ ⎢Yhp ⎥ 0 0 0 ⎥ ⎢Ybp ⎥ ⎢ ⎥ 0 0 0 ⎥ ⎢ Yn ⎥ ⎥ ⎦ − 1 0 0 ⎢⎣Yap ⎥⎦ 0 0

From Eq. (2e), the following phase response is obtained:

_ +

− K1 0 0 0

s2 + ω20 Yn = U s2 + ω0 s+ω20 Q

Topology of the proposed filter shown in Fig. 1 employs two lossy integrator blocks in overall feedback loops, six summing blocks and three multiplying blocks. All the blocks can be classified in two categories: dynamic block which is an integrator block and the other is a static block. The key aspect of the proposed biquad filter architecture is lossy integrator block which is obtained by using the method presented in [58]. It should be also noted that static block designs of the diagram can be easily obtained due to CM process. Also,

+

=

Yhp

2. The proposed filter topology

Σ

K1 0

Coefficients of the first row of matrix equation given in (1) determine Q. The matrix equation in (1) also yields respectively the following well-known five universal filter transfer functions (TFs) namely; lp, hp, bp, n and ap ones:

The paper is organized as follows: After introduction is given in Section 1, design procedure of the proposed universal filter is presented in Section 2. In this section, architecture of the proposed circuit is described in detail by giving a block schema. Section 3 deals with nonideality analysis of the proposed filter structure. The simulations and performance analyses of the proposed circuit are presented in Section 4. In this section, PSpice simulations are performed in order to verify the theoretical results. In order to show the availability of the proposed structure, video filter application is presented in Section 5. An experimental test is achieved in Section 6. Finally, conclusions are given in Section 7.

1

0 ω0 s + ω0

Yap Ylp Yn

β (ω ) =

β0 jω

1+ ω

+ Yhp

β

(5)

Here, ωβ is an angular pole frequency which is ideally equal to infinity. Also, β0 is a DC current gain of the BJT. Therefore, Eq. (4a) turns,

+ Fig. 1. Topology of the proposed filter [58].

2

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Fig. 2. The proposed single input multi output second-order current-mode universal filter.

Fig. 3. Non-ideal case block diagram.

Fig. 5. Applied sinusoidal input current signals.

SKQ1 =

⎛ ω⎞ ⎜1+j ω ⎟ β⎠ ⎝

IE ⎞ CVT ⎛ ω ⎜1+j (β +1) ω ⎟ 0 β⎠ ⎝

(7)

It is seen from Eq. (7) that K1 should be chosen a bit smaller than two for high Q. BJT based SIMO CM universal filter is designed as given in Fig. 2. The proposed circuit mainly consists of forty one transistors, nine DC current sources and two grounded capacitors. In the circuit of Fig. 2, Q1, Q2, Q5 and Q6 and two grounded capacitors are parts of integrators while the other BJTs constitute multipliers and current mirrors. For the proposed filter in Fig. 2, multiplier factors which are controllable by currents are determined as given below. It is seen from the equations given below that each of the coefficients can be adjusted orthogonally.

Fig. 4. Simulated gain against frequency responses of the proposed CM universal filter.

ω0 (ω)=

K1 2 − K1

(6)

It is observed from Eq. (6) that the proposed filter can be operated properly for the frequencies f ≤0.1×ωβ /(2π) [8]. The sensitivity of K1 with respect to Q is evaluated as follows:

I f 1=I f 2

(8a)

I f 7=I f 8

(8b)

K1=

3

If 1 If 3

(8c)

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Fig. 8. Variations of the band-pass responses for various Q at fo =100 kHz.

Fig. 6. FFT of the input signal with bp, n and ap responses.

Fig. 9. Time domain performance of the proposed all-pass filter.

Fig. 7. Electronically tunable phase responses of the all-pass filter.

K2=2

K3=

(8d)

If 7 If 9

Fig. 10. THD variations with respect to applied sinusoidal input currents.

(8e)

second integrator block is reconstituted mainly by changing lossy integrator block with lossless integrator block whose angular pole frequency is ω01 and by adding frequency dependent error block. The new TFs due to non-ideal case marked with apostrophes are given as follows:

3. Non-ideality analysis In order to find out non-ideal case behavior of the proposed filter, a scenario is investigated that the angular pole frequencies of the integrators are not equal to each other. According to this case, an appropriate block model is developed as given in Fig. 3 and this model is analyzed to obtained non-ideal transfer functions. In this model, it is assumed that angular pole frequency of the first integrator is ω01 and angular pole frequency of the second one is ω02 which is different from ω01 by a mismatch error shown as ε . According to this situation, the

Ylp′ U

′ Yhp U

4

=

s2

=

+

s2 +

ω02n ω0n s+ Qn

ω02n

(9a)

s 2 + sε ω0n s + ω02n Q

(9b)

n

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R. Arslanalp et al.

Fig. 11. The deviations in phase angle at resonance frequency where both capacitor values are changed 5%. Fig. 14. A MC analysis to demonstrate phase angle variations of the proposed all-pass filter in which current gains of all the BJTs are changed 5%.

Fig. 12. A MC analysis to show time domain variations of the proposed all-pass filter.

Fig. 15. A MC analysis to demonstrate phase angle variations of the proposed all-pass filter where finite output resistors of all the BJTs are changed 5%.

Fig. 13. A MC analysis to demonstrate phase angle variations of the proposed all-pass filter where both capacitors are changed 5%.

′ Ybp U

=

ω0n s − εs Qn ω s 2 + Q0n s + ω02n n

Yn′ s 2 + ω02n + εs = U s 2 + ω0n s + ω02n Qn

′ Yap U

ω0n s+ Qn ω s 2 + Q0n s n

s2 − =

Fig. 16. Temperature analysis results for the band-pass filter.

⎛ ωω0n − 2ωε ⎞ ⎛ ωω0n ⎞ Q Qn ⎟ ⎟ − tan−1⎜ ϕ (ω) = −tan−1⎜⎜ n2 ⎟ ⎜ ω 2 − ω2 ⎟ 2 − ω ω ⎝ 0n ⎠ ⎠ ⎝ 0n

(9c)

In equations (9), ω0n, Qn and ε depict non-ideal angular resonance frequency, non-ideal quality factor and deviation from ideality, respectively. Also, ω0n, Qn and ε terms are respectively defined as in following equations:

(9d)

ω02n +2εs + ω02n

(10)

(9e) 2 ω0n= ω01 + εω01

From equation (9), the following phase response is obtained: 5

(11a)

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R. Arslanalp et al.

Table 1 Comparison of the proposed filter with previously published ABB based electronically tunable filters [2–26]. References

Number of active building blocks (type)

Number of capacitors (number of grounded capacitors)

Approximate dynamic range (dB)

Technology

Power dissipation (mW)

Providing five filter responsessimultaneously

[2] [3] [4] in Fig. 2 [5] [6] [7] [8] in Fig. 1 [8] in Fig. 2 [9] [10] [11] [12]

3 (CCCII) 3 (CCCII) 3 (CCCII) 4 (CCCII) 5 (CCCII) 5 (CCCII) 4 (CCCII) 5 (CCCII) 5 (CCCII) 4 (CCCII) 3 (CCCII) 1 (CCCA) and 2 (CCCII) 4 (CCCII) 3 (CCCII) 2 (CCCII) 2 (CCCII) 2 (CCCII) 3 (CCCII) 6 (CCCII) 3 (CCCII) 3 (CCCII) 2 (CF) 2 (CCCCTA) 2 (CDTA) 4 (CDTA) 4 (ZC-CFTA) 3 (ZC-CFTA) 4 (ZC-CFTA) 2 (OTA)

2 2 2 2 3 2 2 3 3 2 2 2

(2) (2) (2) (0) (3) (2) (2) (3) (3) (2) (2) (2)

NA NA 30 NA NA 30 NA NA 44 NA 50 NA

NA BJT BJT BJT BJT BJT BJT BJT BJT BJT BJT/ 0.35 µm 0.5 µm

NA NA 19.9 NA NA NA NA NA 17.1 NA NA NA

no no yes no no no no no no yes yes yes

2 2 2 2 2 2 2 2 2 2 2 2 2 2 2 2 2 2

(0) (2) (1) (2) (2) (2) (2) (2) (2) (2) (2) (2) (2) (2) (2) (2) (2) (2)

26 NA NA NA NA NA NA 46 NA NA NA NA 46 34 91 34 NA 28

BJT 0.35 µm BJT BJT BJT BJT BJT BJT BJT BJT BJT 0.5 µm 0.35 µm BJT 0.35 µm 0.35 µm LM13600 BJT

NA NA NA NA NA NA NA ≤ 45 NA NA NA NA 19 12.2 NA 9 NA 4.93

no no no no no no yes yes yes no no no no yes no no no yes

[13] in Fig. [14] [15] in Fig. [15] in Fig. [15] in Fig. [15] in Fig. [16] [17] [18] [19] [20] [21] [22] [23] [24] [25] [26] Proposed *

1 1 2 3 4

*

direct design, NA not available.

Fig. 17. Video filter configuration based on the cascaded two universal filters. Table 2 Specifications of the video filter in Fig. 17. Parameters

The first universal filter

The second universal filter

Q If1 If2 If3 If4 If5 If6 If7 If8 If9

1 11 µA 10 µA 10 µA 12 µA 10 µA 10 µA 10 µA 10 µA 10 µA

5 154.7 µA 140.4 µA 78 µA 93.6 µA 78 µA 78 µA 78 µA 78 µA 390 µA

Qn =

Fig. 18. Gain and phase responses of the video filter in Fig. 17 with quality factors Q1 =1 and Q2 =5.

called as ω0n and Qn, respectively. The second one is the extra terms in numerators and denominator of the TFs due to ε . In order to overcome the first non-ideal effect, electronically tunable property can be used. Thus, the desired angular resonance frequency and quality factor can be fixed. Beside to this, to reduce the influence of the extra terms in the denominators, determined useful frequency range should be obtained. It should be also noted that gain of the bp filter is given in Eq. (12).

2 ω01 + εω01 ω01 Q

+ε

ε=ω02 −ω01

(11b) (11c)

As it is seen from equations (9), if angular pole frequencies of the integrators are not equal to each other, two significant deteriorations are occurred. The first one is the changes in the characteristic parameters which are angular resonance frequency and quality factor

Kbp=

ω0n − Qn ω0n Qn

ε

For hp filter, the following constraint is obtained: 6

(12)

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R. Arslanalp et al.

ω 2 ≫2εω

(17b)

ω02n ≫2εω

(17c)

From equations (17), the following useful frequency range for ap filter is obtained:

0. 1ω02n 20ε ≤fap ≤ 2π 4πε

(18)

Finally, overall useful frequency range of the proposed universal filter is obtained as given in Eq. (18). On the other hand, dynamic range (DR) of a CM filter is computed as

⎛ io, max ⎞ DR = 20 log ⎜ ⎟ ⎝ io, min ⎠

(19)

where io, max and io, min are respectively maximum and minimum output currents when the total harmonic distortion (THD) is less than 4%. Also, when compared to log domain filters [27–34] using capacitor C connected to Emitter of the BJT, each capacitor of the proposed filter can be chosen as C/(β+1) yielding the same resonance frequency. Therefore, the proposed filter occupies less area in IC.

Fig. 19. THD variations of the video filter application in Fig. 17.

4. Simulation results In order to verify the theoretical synthesis, the proposed SIMO CM universal filter given in Fig. 2 is simulated using PSpice with CBIC-R real transistor models [61]. The DC power supply voltage is VCC =2.7 V. Values of both capacitors are chosen as C1 = C2 =50 pF in all the simulations. For the proposed universal filter, control currents If1 =68.3 µA, If4 =74.4 µA and If2 = If3 = If5 = If6 = If7 = If8 = If9 =62 µA resulting in f0 ≅100 kHz and Q ≅1 are chosen. Emitter areas of Q3, Q7 and Q28 are chosen 1.2 times greater, emitter area of Q12 is chosen 0.91 times greater and emitter area of Q26 is chosen 2 times greater than other BJTs. Fig. 4 represents the simulated gain versus frequency responses for the proposed CM universal filter designed with the mentioned parameters. To show the selectivity of the proposed filter, three sinusoidal input current signals in Fig. 5 whose frequencies are chosen as 5 kHz, 100 kHz and 1 MHz are applied simultaneously. Fast Fourier Transforms (FFTs) of the applied input with bp, n and ap responses are carried out as given in Fig. 6. As emphasized above in the design, ω0 and Q can be adjustable only by varying the values of the related current sources of the proposed filter. It should be noted that this benefit gives us wide area of usage without modification on the filter architecture. In Fig. 7, tunable phase response of the ap filter is given where f0 is tuned electronically approximately 1.5 decades. bp filter responses for various Q values at f0 =100 kHz are given in Fig. 8. As shown in Fig. 8, high selectivity is achieved by setting high Q values. 150 nA peak sinusoidal input current is applied to the proposed filter; thus, THD values of the bp filter are approximately found as 1.74% and 2.89% for Q =5 and Q =30, respectively. A sinusoidal current signal with 5 µA peak at resonance frequency (f0 =100 kHz) is applied to input of the proposed all-pass filter in order to show time domain performance. Also, applied input current signal and corresponding output current signal are demonstrated in Fig. 9. THD analysis of the proposed CM SIMO universal filter given in Fig. 10 is performed at 100 kHz for various sinusoidal peak input currents. It is observed from Fig. 10 that DR of the proposed filter is approximately evaluated as 28 dB. In order to observe the effect of tolerance variations of both capacitors, a Monte Carlo (MC) analysis with a hundred runs is performed with the mentioned capacitor values where %5 Gaussian deviations are used. Therefore, the phase angle at resonance frequency is depicted in Fig. 11. Similarly, time domain responses of the proposed all-pass filter at resonance frequency (f0 =100 kHz) are given in Fig. 12 while frequency domain phase angle responses of the proposed all-pass filter are given in Fig. 13. Also, frequency domain phase angle responses of the proposed all-pass filter are given in Figs. 14 and 15

Fig. 20. Experimental setup of the lossy integrator.

Fig. 21. Measured gain response of the lossy integrator.

ω 2 ≫εω

(13)

From Eq. (13), the following useful frequency range for hp filter is obtained:

fhp ≥

10ε 2π

(14)

For n filter, the following constraints are obtained:

ω 2 ≫εω

(15a)

ω02n ≫εω

(15b)

From equations (15), the following useful frequency range for n filter is obtained:

0. 1ω02n 10ε ≤fn ≤ 2π 2πε

(16)

For ap filter, the following constraints are obtained:

ω0n ≫2ε Qn

(17a) 7

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where current gains and finite output resistors of the BJTs are changed 5%, respectively. Temperature analysis results with various temperature values for the bp filter are given in Fig. 16. Total power dissipation of the proposed filter is found as 4.93 mW in PSpice simulations. A comparison of the proposed filter with previously published ABB based electronically tunable filters [2]-[26] is given in Table 1.

associate editor for their invaluable comments for improving the paper. References [1] C. Toumazou, F.J. Lidgey, D.G. Haigh, Analogue IC design: The Current-mode Approach, Peter Peregrinus, London, U.K, 1990. [2] N. Pandey, S.K. Paul, A. Bhattacharyya, S.B. Jain, A novel current controlled current mode universal filter: sito approach, IEICE Electron. Express 2 (17) (2005) 451–457. [3] W. Tangsrirat, W. Surakampontorn, High output impedance current-mode universal filter employing dual-output current-controlled conveyors and grounded capacitors, Intern. J. Electron. Commun. (AEU) 61 (2007) 127–131. [4] E. Yuce, Current-mode electronically tunable biquadratic filters consisting of only CCCIIs and grounded capacitors, Microelectron. J. 40 (2009) 1719–1725. [5] E. Yuce, S. Minaei, O. Cicekoglu, Universal current-mode active-C filter employing minimum number of passive elements, Analog Integr. Circuits Signal Process. 46 (2006) 169–171. [6] S. Minaei, S. Türköz, New current-mode current-controlled universal filter with single input and three outputs, Int. J. Electron. 88 (3) (2001) 333–337. [7] W. Tangsrirat, W. Surakampontorn, Electronically tunable current-mode universal filter employing only plus-type current-controlled conveyors and grounded capacitors, Circuits Syst. Signal Process. 25 (6) (2006) 701–713. [8] E. Yuce, S. Minaei, Universal current-mode filters and parasitic impedance effects on the filter performances, Int. J. Circuit Theory Appl. 36 (2008) 161–171. [9] S. Minaei, S. Turkoz, Current-mode electronically tunable universal filter using only plus-type current controlled conveyors and grounded capacitors, ETRI J. 26 (4) (2004) 292–296. [10] C. Wang, H. Liu, Y. Zhao, A new current-controlled universal filter based on CCCII( ± ), Circuits Syst. Signal Process. 27 (2008) 673–682. [11] N. Pandey, S.K. Paul, S.B. Jain, A new electronically tunable current mode universal filter using MO-CCCII, Analog Integr. Circuits Signal Process. 58 (2009) 171–178. [12] C. Wang, J. Xu, A.Ü. Keskin, S. Du, Q. Zhang, A new current-mode currentcontrolled SIMO-type universal filter, Intern. J. Electron. Commun. (AEU) 65 (3) (2011) 231–234. [13] S. Minaei, E. Yuce, Universal current-mode active-C filters employing only plustype current controlled conveyors, Frequenz 60 (7–8) (2006) 134–137. [14] H.-P. Chen, P.-L. Chu, Versatile universal electronically tunable current-mode filter using CCCIIs, IEICE Electron. Express 6 (2) (2009) 122–128. [15] E. Yuce, A. Kircay, S. Tokat, Universal resistorless current-mode filters employing CCCIIs, Int. J. Circuit Theory Appl. 36 (2008) 739–755. [16] M.T. Abuelma’atti, N.A. Tasadduq, A novel single-input multiple-output currentmode current-controlled universal filter, Microelectron. J. 29 (1998) 901–905. [17] M. Kumngern, F. Khateb, P. Phasukkit, S. Tungjitkusolmun, S. Junnapiya, ECCCIIbased current-mode universal filter with orthogonal control of ωo and Q, Radioengineering 23 (2) (2014) 687–696. [18] M. Kumngern, “A new current-mode universal filter with single-input five-output using trans-linear current conveyors, Aust. J. Electr. Electron. Eng. 9 (2) (2012) 177–184. [19] W. Tangsrirat, D. Prasertsom, Electronically tunable low-component-count current-mode biquadratic filter using dual-output current followers, Electr. Eng. 90 (2007) 33–37. [20] S.V. Singh, S. Maheshwari, Current-processing current-controlled universal biquad filter, Radioengineering 21 (1) (2012) 317–323. [21] A.U. Keskin, D. Biolek, E. Hancioglu, V. Biolkova, Current-mode KHN filter employing current differencing transconductance amplifiers, Int. J. Electron. Commun. (AEU) 60 (2006) 443–446. [22] W. Tangsrirat, T. Pukkalanun, Structural generation of two integrator loop filters using CDTAs and grounded capacitors, Int. J. Circuit Theory Appl. 39 (1) (. 2011) 31–45. [23] J. Satansup, W. Tangsrirat, Single-input five-output electronically tunable currentmode biquad consisting of only ZC-CFTAs and grounded capacitors, Radioengineering 20 (3) (2011) 650–655. [24] B. Singh, A.K. Singh, R. Senani, New universal current-mode biquad using only three ZC-CFTAs, Radioengineering 21 (1) (2012) 273–280. [25] J. Satansup, W. Tangsrirat, Realization of current-mode KHN-equivalent biquad filter using ZC-CFTAs and grounded capacitors, Indian J. Pure Appl. Phys. 49 (12) (2011) 841–846. [26] C.M. Chang, S.-K. Pai, Universal current-mode OTA-C biquad with the minimum components, IEEE Trans. Circuits Syst.-I: Fundam. Theory Appl. 47 (8) (2000) 1235–1238. [27] A.T. Tola, R. Arslanalp, S.S. Yilmaz, Current mode high-frequency KHN filter employing differential class AB, Log. Domain Integr.,” Int. J. Electron. Commun. (AEU) 63 (7) (2009) 600–608. [28] A. Kircay, U. Cam, Differential type Class-AB second-order log-domain notch filter, IEEE Trans. Circuits Syst.-I: Regul. 55 (5) (2008) 1203–1212. [29] D.R. Frey, Exponential state space filters: a generic current mode-design strategy, IEEE Trans. Circuits Syst. I: Fundam. Theory Appl. 43 (1) (1996) 34–42. [30] N. Duduk, A.T. Tola, Log-domain universal biquad filter design using lossy integrators, Elektron. Ir. Elektro. 22 (3) (2016) 56–59. [31] D. Perry, G.W. Roberts, The design of log-domain filters based on the operational simulation of LC ladders, IEEE Trans. Circuits Syst. II: Express Briefs 43 (11) (1996) 763–774. [32] C. Psychalinos, Log-domain SIMO and MISO low-voltage universal biquads, Analog Integr. Circuits Signal Process. 67 (2) (2011) 201–211.

5. An application example In this section, as an application of the discussed above SIMO universal filter, a video filter obtained by cascade of the proposed notch filter and low-pass filter is constructed. lp output of the first filter is connected as input of the second n filter illustrated in Fig. 17. TF of the video filter application is given in Eq. (20). The specifications of the video filter are detailed as given in Table 2. Gain and phase responses of the video filter application in Fig. 17 are depicted in Fig. 18 where Q1 =1, Q2 =5, f01 ≅55 kHz and f02 ≅82 kHz. THD variations of the video filter application in Fig. 17 at 10 kHz versus applied peak sinusoidal input currents are demonstrated in Fig. 19. 2 2 ω01 s 2 + ω02 Yv = 2 ω01 ω02 2 2 2 U s + s Q + ω01 s + s Q + ω02 1

2

(20)

6. An experimental test result As mentioned above, the most important part of the designed filter circuit is lossy integrator. In order to confirm the circuit operation of the integrator in practice, this part is implemented on breadboard. As shown in Fig. 20, to accomplish experimental test of the introduced integrator circuit, two commercially available active devices such as AD844s [62] with symmetrical DC power supply voltages ± 6 V, two BJTs, two resistors and one capacitor are used. The supply voltage, VCC in Fig. 20 is set to 1 V. The passive components are chosen as C =1nF and R1 = R2 =1kΩ resulting in a pole frequency of f0 ≅3 kHz. As shown in Fig. 21, the experimental test result confirms both the theoretical and simulated results of the proposed lossy integrator especially at low frequencies. However, due to the effects of parasitic resistors and capacitors of the board, a bit difference between ideal and experimental results occurs. 7. Conclusion A novel SIMO CM universal filter simultaneously realizing five fundamental filter responses is proposed in this paper. In the design of the circuit, two lossy integrator loop based block diagram is employed. All the blocks in the diagram are designed by using essential electronic components which are BJTs, current sources, DC power supply and two grounded capacitors. All the steps of the process are given in detail in the related sections. In order to show validity and feasibility of the proposed filter, PSpice simulations including frequency domain and time domain ones are performed. Non-ideality analysis of the proposed universal filter is investigated. As an application of the proposed universal filter, a video filter configuration based on the cascade of the proposed notch filter and low-pas filter is given. Moreover, practical realization of the lossy integrator block which plays a leading role in the process is accomplished. Ideal, simulation and experimental results verify the claimed theory well. It is expected that the proposed universal configuration will be useful in communications, instrumentation, etc. Acknowledgments The circuit of this paper was derived from the PhD thesis given in [63]. Also, we would like to thank the anonymous reviewers and 8

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with current feedback op-amps, Frequenz 51 (1997) 203–208. [49] R.K. Sharma, R. Senani, Universal current mode biquad using a single CFOA, Int. J. Electron. 91 (3) (2004) 175–183. [50] H.-P. Chen, Versatile current-mode universal biquadratic filter using DO-CCIIs, Int. J. Electron. 100 (7) (2013) 1010–1031. [51] E.O. Güneş, A. Toker, S. Özoğuz, Insensitive current-mode universal filter with minimum components using dual-output current conveyors, Electron. Lett. 35 (7) (1999) 524–525. [52] S. Özoguz, C. Acar, Universal current-mode filter with reduced number of active and passive components, Electron. Lett. 33 (11) (1997) 948–949. [53] A.M. Soliman, Voltage mode and current mode Tow Thomas bi-quadratic filters using inverting CCII, Int. J. Circuit Theory Appl. 35 (4) (2007) 463–467. [54] A.M. Soliman, Current mode filters using two output inverting CCII, Int. J. Circuit Theory Appl. 36 (7) (2008) 875–881. [55] H.-P. Chen, Current-mode dual-output ICCII-based tunable universal biquadratic filter with low-input and high-output impedances, Int. J. Circuit Theory Appl. 42 (4) (2014) 376–393. [56] A.Ü. Keskin, U. Cam, Insensitive high-output impedance minimum configuration SITO-type current-mode biquad using dual-output current conveyors and grounded passive components, Int. J. Electron. Commun. (AEU) 61 (2007) 341–344. [57] M. Bhushan, R.W. Newcomb, Grounding of capacitors in integrated circuits, Electron. Lett. 3 (4) (1967) 148–149. [58] R. Arslanalp, A.T. Tola, E. Yuce, Novel resistorless first-order current-mode universal filter employing a grounded capacitor, Radioengineering 20 (3) (. 2011) 656–665. [59] A.S. Sedra, K.C. Smith, Microelectronics Circuits, 6th edition, Oxford University Press, 2011. [60] E. Yuce, S. Minaei, O. Cicekoglu, Limitations of the simulated inductors based on a single current conveyor, IEEE Trans. Circuits Syst.-I: Regul. 53 (12) (2006) 2860–2867. [61] D.R. Frey, Log-domain filtering: an approach to current-mode filtering, IEEE Proc.G Circuits Devices Syst. 140 (6) (1993) 406. [62] Analog Devices, AD844 Datasheet (Rev. F), Doc. Number: D00897-0-2/09(F), 〈http://www.analog.com/static/imported-files/data_sheets/AD844.pdf〉. Accessed 01 January 2013. [63] R. Arslanalp, Synthesis of Electronically Tunable Analog Functional Blocks,”blocks, PhD thesis, Pamukkale University, 2011.

[33] N.A. Shah, F.A. Khanday, Multiple-input-multiple-output log-domain universal biquad filter, Indian J. Pure Appl. Phys. 50 (2012) 928–934. [34] P. Premmee, K. Dejhan, Single-input multiple-output tunable log-domain currentmode universal filter, Radioengineering 22 (11) (2013) 474–484. [35] C. Laoudias, C. Psychalinos, E. Stoumpou, 1.5V square-root domain universal biquad filters, Int. J. Circuit Theory Appl. 41 (11) (2011) 307–318. [36] S.S. Yilmaz, A.T. Tola, R. Arslanalp, A novel second-order all-pass filter using square-root domain blocks, Radioengineering 22 (1) (2013) 179–185. [37] G.-J. Yu, C.-Y. Huang, B.-D. Liu, J.-J. Chen D, Design of square-root domain filters, Analog Integr. Circuits Signal Process. 43 (2005) 49–59. [38] S.S. Yilmaz, A.T. Tola, Fifth order butterworth low pass square-root domain filter design, Elektron. Ir. Elektro. no. 9 (2011) 55–58. [39] F.A. Khanday, C. Psychalinos, N.A. Shah, Universal filters of arbitrary order and type employing square-root-domain technique, Int. J. Electron. 101 (7) (2014) 894–918. [40] A. Fabre, O. Saaid, F. Wiest, C. Boucheron, High frequency applications based on a new current controlled conveyor, IEEE Trans. Circuits Syst.: Fundam. Theory Appl. 43 (1996) 82–91. [41] J. Jerabek, K. Vrba, SIMO type low-input and high-output impedance currentmode universal filter employing three universal current conveyors, Int. J. Electron. Commun. (AEU) 64 (2010) 588–593. [42] T. Tsukutani, Y. Sumi, N. Yabuki, Novel current-mode biquadratic circuit using only plus type DO-DVCCs and grounded passive components, Int. J. Electron. 94 (12) (2007) 1137–1146. [43] H.-P. Chen, Tunable versatile current-mode universal filter based on plus-type DVCCs, Int. J. Electron. Commun. (AEU) 66 (2012) 332–339. [44] M.A. Ibrahim, S. Minaei, H. Kuntman, A 22.5 MHz current-mode KHN-biquad using differential voltage current conveyor and grounded passive elements, Int. J. Electron. Commun. (AEU) 59 (2005) 311–318. [45] S. Minaei, M.A. Ibrahim, A mixed-mode KHN-biquad using DVCC and grounded passive elements suitable for direct cascading, Int. J. Circuit Theory Appl. 37 (2009) 793–810. [46] R.K. Sharma, R. Senani, On the realization of universal current mode biquads using a single CFOA, Analog Integr. Circuits Signal Process. 41 (2004) 65–78. [47] R.K. Sharma, R. Senani, Multifunction CM/VM biquads realized with a single CFOA and grounded capacitors, Int. J. Electron. Commun. (AEU) 57 (5) (2003) 301–308. [48] R. Senani, S.S. Gupta, Universal voltage-mode/current-mode biquad filter realised

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