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A current integrator for use with low variable currents
NUCLEAR
INSTRUMENTS
AND METHODS
60
([968)
323-325; © NORTH-HOLLAND
PUBLISHING
CO.
A CURRENT I N T E G R A T O R FOR USE W I T H L O W VARIABLE CURRENTS* S. K. G U P T A t
Tara Institute of Fundamental Research, Bombay, India Received 6 November 1967 A transistorized current integrator using a very high input impedance has been designed. It is found to be linear in the range 1 nA to
10/~A. The leakage has been reduced to ~ 5 x 10-12 A and reproducibility of the calibration is less than + 0.3%.
1. Introduction A current integrator is a widely used instrument for the measurement of the electric charge when the current is fluctuating. For example it can be used as such for measuring photoelectric and thermionic currents and accelerator beam currents. With slight modifications it can be used as analog to digital converter for the pulse height analysis. In nuclear measurements the integrator measures the total charge of the incident particles falling on the target. A good current integrator should have low input leakage, linearity, stability and preferably a digital output. The design of the instrument is therefore, to be decided by the following considerations. 1. The leakage in the integrator can be due to the integrating capacitor, higher triggering voltage, the grid current of the first stage and the discharging circuit. For the currents handled in nuclear experiments the leakage current should be less than 10-11 A. 2. To measure fluctuating currents the integrator should be linear over its whole range. 3. The stability of the integrator should be good so as to give reproducible results for the same amount of the charge collected. 4. The dead time of the integrator should be small to integrate fast fluctuating currents accurately. 5. For occasional surge currents the integrator should not stop working. * Work done at Van de Graaff Laboratory, Nuclear Physics Division, Bhabha Atomic Research Centre, Trombay, Bombay.
t Present address: Van de Graaff Laboratory, Bhabha Atomic Research Centre, Trombay, Bombay.
6. The accuracy of the charge measurement need not be better than other accuracies involved in a particular measurement. In most of the nuclear experiments the statistical accuracy is ~ 0.3%. Therefore an integrator accurate to 0.3% may serve the purpose well. Many integrator circuits have been described in the literature1-4), but either they involve complicated circuitry or they do not satisfy the above requirements. We here describe a simple circuit which has satisfied the above requirements.
2. Principle of operation A block diagram of the circuit is shown in fig. 1. The dc amplifier acts as a charging circuit. When its output reaches the trigger level of the Schmitt trigger, it operates and the gated multivibrator becomes free running. Every pulse of the multivibrator actuates the charge pump to take away a constant charge Q0 from the input of the amplifier. Therefore by counting the number n of the pulses coming out of the multivibrator through the emitter follower, the measured charge q over a period is given by (1)
q = nQo+(QR2-QR,),
where QR, is the charge at the input of the amplifier at the start and QR2 is the charge at the end of the period. QR2--QR, is always less than Qo and can be made negligible by increasing n. The stability of the instrument depends on the constancy of Qo. Qo is determined by the supply voltage Vcc of the collector and the saturation voltage Vs, both
C!
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J D.C.AMPLIFIER
°1
I
J JlSCHMITT TRIGGER ~ I J
]
GATED
FREERUNNING JMULTIVIBRATER
FOLLOWER1 3VOLT5
CHARGE PUMP Fig. 1. Block diagram of the current integrator.
323
324
s.K.
CHARGE PUMP
GUPTA
SCHMITT TRIGGER
D.C . A M P L I F I E R
1 GATED 'IFREE RUNNING IIMULTIVIBRATER
EMffTER FOLLOWER
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o4
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~o (2"Z 7 0 /
Fig. 2. Complete circuit diagram of the current integrator. of the discharging transistor Ta0 (fig. 2). If the voltage at the input of the amplifier is v, Qo is given by
Qo = c 2 ( V c c - Vs + v)
c=(Vcc- vs), provided Vcc >> v.
(2)
The variation Vs is small with temperature and therefore A Q o / Q 0 = { ( A C 2 / C 2 ) 2 + ( A V c c / V c c ) 2 } I-. (3) Therefore the stability of the instrument is dependent on the stability of the capacitor C2 and the voltage Vco C2 and Vcc can change due to temperature and aging. In practice one should be able to achieve a stability better than 0.01% by choosing a temperature compensated capacitor and power supply and also by regulating the temperature of the equipment. 3. Description of the circuit The complete circuit diagram is given in fig. 2. The electrometer tube Ta (CK 5886) has been chosen as the input stage due to its low grid current ~ 10-aSA. T z has been used to minimise the changes in the cathode voltage of T1 due to the changes in the power supply voltage. The tube T 1 and the transistors T 3 and T4 form
the direct coupled amplifier. The feedback from the collector of T 4 to the screen of T~ has been used to reduce the drift in the output dc level of the amplifier. The feedback capacitor C~ from the collector of T4 to the grid of T t decides the output voltage V = Q / C a , where Q is the charge at the input.
T5 and T 6 form a direct coupled Schmitt trigger circuit. Effectively T,, W3, T 4 , T 5 and T 6 form a trigger circuit and its trigger level V0 can be adjusted by varying the screen grid voltage of Tx. When V rises to V0, T 5 starts conducting and T 6 gets cut off. T 7 and T8 form a circuit of a free-running multivibrator which is normally inoperative because of the conducting diode D 3. A s T 6 starts conducting, D 3 stops conducting and the multivibrator runs freely till the diode D 3 starts conducting again. The free-running frequency of the multivibrator is slightly above 10 kc/s. For non-overload conditions this multivibrator acts as a univibrator. The positive output from the collector of T 7 sets the normally cut-off T, o into heavy conduction and thereby reduces the voltage of the collector of T~0 almost to the ground level. The diodes D~ and D2 are normally nonconducting due to the insufficient voltage across them for conduction. The negative pulse at the collector of Tlo makes D 1 conduct and the charge gets transferred
A CURRENT INTEGRATOR
from the grid of T 1 to the cathode of D 1 and when D 1 stops conducting, the charge is conducted away to the ground through D 2. Thus, with every pulse of the multivibrator, a charge equal to Qo is subtracted. This in turn reduces the output of the amplifier and in case the charge subtracted is equal to the charge collected, the voltage V falls to zero and the multivibrator is switched off. The collected and the subtracted charges can be equated by putting a Vacuum tube voltmeter at the input and by setting the potentiometer in the screen circuit of T1 to get zero volt at the grid of T1 while counting non-overload currents. When the circuit receives overload currents, the multivibrator becomes free-running and goes on subtracting the charge until the output voltage of the amplifier fails below Vo. The output from the emitter follower T 9 c a n be counted on a scalar. By connecting the output to a frequency meter, a digital reading for the current is obtained. If a count rate meter is used to sense the output pulses of the integrator, an analogue deflection proportional to the input current will be obtained. It can be easily seen that the dead time of the instrument is small. D1 conducts due to the discharge pulse appearing on the capacitor C 2 and the charge transferred is Qo and is bound to it for the duration of the pulse. Any charge coming during this period can, therefore, charge only the capacitors Ct and C3. When the discharge pulse on the capacitor C2 dies away, the bound charge becomes free to get conducted to the ground. During this period, Dx should be off due to reverse bias across it. The only dead time of instrument occurs only when both D 1 and D2 are conducting. The upper limit for this condition is the switching time of the diodes D 1 and D2. In our circuit it should be less than 1/~s. The capacitors C1, C 2 and C3 should be of the low leakage type and especially C1 and C2 are to be carefully selected by counting zero signal counts and estimating the leakage. Some polystyrene and silvered mica capacitors have been found to be satisfactory. Both the diodes D 1 and D 2 are low leakage silicon diodes 1N 3579. The transistors T3, T 4 and Txo are silicon transistors and are chosen to be of the low leakage type. The values of C1 and C2 can be chosen
325
appropriately depending upon the calibration factor desired. With the values shown in the figure, one count is equal to 1.I nC. As the maximum counting rate is 10 kc/s, we, therefore, can handle currents upto 10 pA. The transistors T 4 and Tlo can be replaced by 2N1711 and 2N914 respectively and the diode D 3 by 1N3579. The power supply has been derived by using 10 and 16V zener diodes from a readily available + 300 V dc power supply. 4. Performance and tests The circuit has been tested using a mercury cell to generate different currents by using different resistors. The reproducibility of the calibration of the circuit is limited to + 0.3% over one week of continuous operation. This limitation is due partly to the zener diodes used which had a temperature coefficient of 0.07% / ° C and partly to the drift in the input dc level. The second factor should not contribute in the actual measurement involving ideal currents sources e.g. positive ion currents from the accelerators. The input leakage of the circuit is estimated to be less than 5 x 10 -12 A. It is found necessary to shield the electrometer tube from light to reduce the leakage. The linearity for currents upto 0.1 #A is found to be better than 0.2% and is 0.2% for the currents upto 10/~A. The precision of the instrument can be further improved by using temperature compensated Zener diodes and by putting the whole instrument in a constant temperature bath. The author is thankful to Dr. A. S. Divatia and Dr. M. K. Mehta for their interest in this work and for encouragement, and to Dr. N. Sarma and Mr. S. S. Kerekatte for many stimulating discussions during the progress of this work. References 1) p. j. M. Smulders and P. B. Smith, Nucl. Instr. and Meth. 8 (1960) 40. 2) I. Pelah and D. Maeden, IRE Trans. Nucl. Sci. NS-9, no. 5 (1962) 27. a) E. J. Rogers, Rev. Sci. Instr. 34 (1963) 660. 4) E. Blignaut and J. J. Kritzinger, Nucl. Instr. and Meth. 36 (1965) 176.
INSTRUMENTS
AND METHODS
60
([968)
323-325; © NORTH-HOLLAND
PUBLISHING
CO.
A CURRENT I N T E G R A T O R FOR USE W I T H L O W VARIABLE CURRENTS* S. K. G U P T A t
Tara Institute of Fundamental Research, Bombay, India Received 6 November 1967 A transistorized current integrator using a very high input impedance has been designed. It is found to be linear in the range 1 nA to
10/~A. The leakage has been reduced to ~ 5 x 10-12 A and reproducibility of the calibration is less than + 0.3%.
1. Introduction A current integrator is a widely used instrument for the measurement of the electric charge when the current is fluctuating. For example it can be used as such for measuring photoelectric and thermionic currents and accelerator beam currents. With slight modifications it can be used as analog to digital converter for the pulse height analysis. In nuclear measurements the integrator measures the total charge of the incident particles falling on the target. A good current integrator should have low input leakage, linearity, stability and preferably a digital output. The design of the instrument is therefore, to be decided by the following considerations. 1. The leakage in the integrator can be due to the integrating capacitor, higher triggering voltage, the grid current of the first stage and the discharging circuit. For the currents handled in nuclear experiments the leakage current should be less than 10-11 A. 2. To measure fluctuating currents the integrator should be linear over its whole range. 3. The stability of the integrator should be good so as to give reproducible results for the same amount of the charge collected. 4. The dead time of the integrator should be small to integrate fast fluctuating currents accurately. 5. For occasional surge currents the integrator should not stop working. * Work done at Van de Graaff Laboratory, Nuclear Physics Division, Bhabha Atomic Research Centre, Trombay, Bombay.
t Present address: Van de Graaff Laboratory, Bhabha Atomic Research Centre, Trombay, Bombay.
6. The accuracy of the charge measurement need not be better than other accuracies involved in a particular measurement. In most of the nuclear experiments the statistical accuracy is ~ 0.3%. Therefore an integrator accurate to 0.3% may serve the purpose well. Many integrator circuits have been described in the literature1-4), but either they involve complicated circuitry or they do not satisfy the above requirements. We here describe a simple circuit which has satisfied the above requirements.
2. Principle of operation A block diagram of the circuit is shown in fig. 1. The dc amplifier acts as a charging circuit. When its output reaches the trigger level of the Schmitt trigger, it operates and the gated multivibrator becomes free running. Every pulse of the multivibrator actuates the charge pump to take away a constant charge Q0 from the input of the amplifier. Therefore by counting the number n of the pulses coming out of the multivibrator through the emitter follower, the measured charge q over a period is given by (1)
q = nQo+(QR2-QR,),
where QR, is the charge at the input of the amplifier at the start and QR2 is the charge at the end of the period. QR2--QR, is always less than Qo and can be made negligible by increasing n. The stability of the instrument depends on the constancy of Qo. Qo is determined by the supply voltage Vcc of the collector and the saturation voltage Vs, both
C!
II
IN PUT
J D.C.AMPLIFIER
°1
I
J JlSCHMITT TRIGGER ~ I J
]
GATED
FREERUNNING JMULTIVIBRATER
FOLLOWER1 3VOLT5
CHARGE PUMP Fig. 1. Block diagram of the current integrator.
323
324
s.K.
CHARGE PUMP
GUPTA
SCHMITT TRIGGER
D.C . A M P L I F I E R
1 GATED 'IFREE RUNNING IIMULTIVIBRATER
EMffTER FOLLOWER
i
r,,,
r~
re c, ,oooer ra
~
: rs
II
r~
I
: i
rz
r~
r~
m
t ,~,TK
Do~ ~ _.,2, o o / ~
h~'~
"°~$.-~g 11;oo.I ',, ,,"-~ g ~ . ~ 5 ~' . : [ I
',
L
I
4
L-----~÷eay i
zg/, z)z
//¢357.q
7-/ C K 5 8 8 6
~
C/l. s2,~
z)3
c z) e 1
zz o c z,e
r s . r9 za,~o~
o4
//¢ ~4
7A z , , , / , z / r
~o (2"Z 7 0 /
Fig. 2. Complete circuit diagram of the current integrator. of the discharging transistor Ta0 (fig. 2). If the voltage at the input of the amplifier is v, Qo is given by
Qo = c 2 ( V c c - Vs + v)
c=(Vcc- vs), provided Vcc >> v.
(2)
The variation Vs is small with temperature and therefore A Q o / Q 0 = { ( A C 2 / C 2 ) 2 + ( A V c c / V c c ) 2 } I-. (3) Therefore the stability of the instrument is dependent on the stability of the capacitor C2 and the voltage Vco C2 and Vcc can change due to temperature and aging. In practice one should be able to achieve a stability better than 0.01% by choosing a temperature compensated capacitor and power supply and also by regulating the temperature of the equipment. 3. Description of the circuit The complete circuit diagram is given in fig. 2. The electrometer tube Ta (CK 5886) has been chosen as the input stage due to its low grid current ~ 10-aSA. T z has been used to minimise the changes in the cathode voltage of T1 due to the changes in the power supply voltage. The tube T 1 and the transistors T 3 and T4 form
the direct coupled amplifier. The feedback from the collector of T 4 to the screen of T~ has been used to reduce the drift in the output dc level of the amplifier. The feedback capacitor C~ from the collector of T4 to the grid of T t decides the output voltage V = Q / C a , where Q is the charge at the input.
T5 and T 6 form a direct coupled Schmitt trigger circuit. Effectively T,, W3, T 4 , T 5 and T 6 form a trigger circuit and its trigger level V0 can be adjusted by varying the screen grid voltage of Tx. When V rises to V0, T 5 starts conducting and T 6 gets cut off. T 7 and T8 form a circuit of a free-running multivibrator which is normally inoperative because of the conducting diode D 3. A s T 6 starts conducting, D 3 stops conducting and the multivibrator runs freely till the diode D 3 starts conducting again. The free-running frequency of the multivibrator is slightly above 10 kc/s. For non-overload conditions this multivibrator acts as a univibrator. The positive output from the collector of T 7 sets the normally cut-off T, o into heavy conduction and thereby reduces the voltage of the collector of T~0 almost to the ground level. The diodes D~ and D2 are normally nonconducting due to the insufficient voltage across them for conduction. The negative pulse at the collector of Tlo makes D 1 conduct and the charge gets transferred
A CURRENT INTEGRATOR
from the grid of T 1 to the cathode of D 1 and when D 1 stops conducting, the charge is conducted away to the ground through D 2. Thus, with every pulse of the multivibrator, a charge equal to Qo is subtracted. This in turn reduces the output of the amplifier and in case the charge subtracted is equal to the charge collected, the voltage V falls to zero and the multivibrator is switched off. The collected and the subtracted charges can be equated by putting a Vacuum tube voltmeter at the input and by setting the potentiometer in the screen circuit of T1 to get zero volt at the grid of T1 while counting non-overload currents. When the circuit receives overload currents, the multivibrator becomes free-running and goes on subtracting the charge until the output voltage of the amplifier fails below Vo. The output from the emitter follower T 9 c a n be counted on a scalar. By connecting the output to a frequency meter, a digital reading for the current is obtained. If a count rate meter is used to sense the output pulses of the integrator, an analogue deflection proportional to the input current will be obtained. It can be easily seen that the dead time of the instrument is small. D1 conducts due to the discharge pulse appearing on the capacitor C 2 and the charge transferred is Qo and is bound to it for the duration of the pulse. Any charge coming during this period can, therefore, charge only the capacitors Ct and C3. When the discharge pulse on the capacitor C2 dies away, the bound charge becomes free to get conducted to the ground. During this period, Dx should be off due to reverse bias across it. The only dead time of instrument occurs only when both D 1 and D2 are conducting. The upper limit for this condition is the switching time of the diodes D 1 and D2. In our circuit it should be less than 1/~s. The capacitors C1, C 2 and C3 should be of the low leakage type and especially C1 and C2 are to be carefully selected by counting zero signal counts and estimating the leakage. Some polystyrene and silvered mica capacitors have been found to be satisfactory. Both the diodes D 1 and D 2 are low leakage silicon diodes 1N 3579. The transistors T3, T 4 and Txo are silicon transistors and are chosen to be of the low leakage type. The values of C1 and C2 can be chosen
325
appropriately depending upon the calibration factor desired. With the values shown in the figure, one count is equal to 1.I nC. As the maximum counting rate is 10 kc/s, we, therefore, can handle currents upto 10 pA. The transistors T 4 and Tlo can be replaced by 2N1711 and 2N914 respectively and the diode D 3 by 1N3579. The power supply has been derived by using 10 and 16V zener diodes from a readily available + 300 V dc power supply. 4. Performance and tests The circuit has been tested using a mercury cell to generate different currents by using different resistors. The reproducibility of the calibration of the circuit is limited to + 0.3% over one week of continuous operation. This limitation is due partly to the zener diodes used which had a temperature coefficient of 0.07% / ° C and partly to the drift in the input dc level. The second factor should not contribute in the actual measurement involving ideal currents sources e.g. positive ion currents from the accelerators. The input leakage of the circuit is estimated to be less than 5 x 10 -12 A. It is found necessary to shield the electrometer tube from light to reduce the leakage. The linearity for currents upto 0.1 #A is found to be better than 0.2% and is 0.2% for the currents upto 10/~A. The precision of the instrument can be further improved by using temperature compensated Zener diodes and by putting the whole instrument in a constant temperature bath. The author is thankful to Dr. A. S. Divatia and Dr. M. K. Mehta for their interest in this work and for encouragement, and to Dr. N. Sarma and Mr. S. S. Kerekatte for many stimulating discussions during the progress of this work. References 1) p. j. M. Smulders and P. B. Smith, Nucl. Instr. and Meth. 8 (1960) 40. 2) I. Pelah and D. Maeden, IRE Trans. Nucl. Sci. NS-9, no. 5 (1962) 27. a) E. J. Rogers, Rev. Sci. Instr. 34 (1963) 660. 4) E. Blignaut and J. J. Kritzinger, Nucl. Instr. and Meth. 36 (1965) 176.