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A Novel Approach towards the Designing of an Antenna for Aircraft Collision Avoidance System
Int. J. Electron. Commun. (AEÜ) 71 (2017) 53–71
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International Journal of Electronics and Communications (AEÜ) journal homepage: www.elsevier.com/locate/aeue
Regular paper
A Novel Approach towards the Designing of an Antenna for Aircraft Collision Avoidance System Debajit De ⇑, Prasanna Kumar Sahu Department of Electrical Engineering, National Institute of Technology, Rourkela, India
a r t i c l e
i n f o
Article history: Received 21 September 2016 Revised 31 October 2016 Accepted 2 November 2016
Keywords: TCAS/ACAS Monopole antenna Microstrip Antenna S Parameter VSWR Impedance E-Field Gain Beamwidth Side lobe
a b s t r a c t TCAS/ACAS (Traffic/Aircraft Collision Avoidance System) is an airborne system designed to increase cockpit awareness of nearby aircraft and to service as a last defense against mid-air collisions between the aircrafts. In the existing system, four monopole stub elements are used as TCAS directional antenna and one blade type element is used as TCAS Omni-directional antenna. The transmission and reception frequencies of the TCAS antenna are 1.03G Hz and 1.09G Hz respectively. The existing TCAS antenna has some drawbacks such as low gain, large beamwidth, frequency and beam tuning/scanning issues etc. Antenna issues like unwanted signals reception may create difficulties in identifying the possible threats. In this paper, the focus is on the design and development of a novel Microstrip Antenna Array which can be used for TCAS/ACAS application. Two proposed antenna models are presented here – a Unit Element Dual Feed Microstrip Dual Patch Slotted Antenna and a Compact Microstrip Antenna Array. These are designed in CST tool to meet the current needs of aircraft Collision Avoidance System and to overcome the possible limitations associated with the existing techniques. The performance and other antenna characteristics have been explored from the simulation results followed by the antenna fabrication and measurement. A good Reflection Coefficient and VSWR with proper 50 X Impedance Matching, narrow Beamwidth, perfect Directional Radiation Pattern, high Gain and Directivity at the operating frequencies make this proposed antenna a good candidate for TCAS application. The proposed antenna would be expected to meet the requirements of the advanced avionics standards in terms of design simplicity, lightweight and high performance. Ó 2016 Elsevier GmbH. All rights reserved.
1. Introduction The Aircraft Collision Avoidance System provide a solution of reducing the risk of mid-air collisions between aircraft. TCAS is a family of airborne systems that function independently of ground-based air traffic control (ATC) to provide collision avoidance protection. The TCAS concept makes use of the radar beacon transponders carried by aircraft for ground ATC purposes and provides no protection against aircraft that do not have an operating transponder [1]. After many years of extensive analysis, development, and flight evaluation by the Federal Aviation Administration (FAA), Civil Aviation Authorities (CAAs), and the aviation industries, Traffic Collision Avoidance System (TCAS) was developed to reduce the risk of mid-air collisions between aircraft [2]. The main feature of TCAS, which was first proposed by Dr. John S. Morell in 1955, is that the function is according to time criteria and not ⇑ Corresponding author. E-mail addresses: (P.K. Sahu).
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(D.
http://dx.doi.org/10.1016/j.aeue.2016.11.002 1434-8411/Ó 2016 Elsevier GmbH. All rights reserved.
De),
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the distance. From several successive replies, TCAS calculates a time to reach the CPA (Closest Point of Approach) with the intruder, by dividing the range by the closure rate [3]. In 1983, J. D. Welch and V. A. Orlando of Lincoln Laboratory in M.I.T, USA had published first official report on functional overview of TCAS system [4]. In this report, the presented a review of the functions performed by any collision avoidance system and then a definition of the way in which these functions are implemented in the TCAS II. This section concludes with a summary of TCAS II design parameters. TCAS/ACAS employs radio signals for surveillance of nearby aircraft, and in dangerous encounters, it warns the aircraft pilot by means of cockpit displays and auditory alarms. To detect the presence of nearby aircraft, TCAS transmits interrogations at a steady rate, nominally once per second, and employs a receiver for detecting the replies to these interrogations from the transponders on nearby aircraft as shown in Fig. 1. Fig. 2 shows the basic block diagram of overall TCAS system. TCAS consists of the Mode S/TCAS Control Panel, the Mode S Transponder, the TCAS Computer, Antennas, Traffic and Resolution Advisory Displays, and an Aural
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Fig. 1. Air-to-air surveillance.
Fig. 2. TCAS/ACAS system block diagram.
Annunciator. Control information from the Mode S/TCAS Control Panel is provided to the TCAS Computer via the Mode S Transponder [5]. In 1991, K. S. Sampath, R. G. Rojas and W. D. Burnside of ElectroScience Laboratory in Ohio State University, USA had discussed 2 different implementations of the TCAS Antenna [6]. In this paper, 8 elements circular antenna array with a monopulse principle and a 4 element antenna with an amplitude comparison scheme are presented. The purpose of this paper is to study the effect of aircraft itself on the performance of TCAS in estimating the bearing angle of the intruder aircraft. The performance of the 4 elements TCAS was not as good as the 8 elements but the 4 elements amplitude system is much simpler in its implementation than the 8 elements monopulse system. In 2011, W. H. Harman and M. L. Wood of Lincoln Laboratory in M.I.T, USA have proposed a Triangle TCAS Antenna [7]. This report describes an antenna of 3 elements in the shape of a triangle. It illustrates the concept of the triangle antenna as compared with the conventional TCAS antenna. The triangle antenna offers an advantage in angle of arrival accuracy as affected by receiver noise. Presently, TCAS/ACAS uses a directional antenna, mounted on top of the aircraft. An Omni-directional transmitting and receiving antenna is mounted at the bottom of the aircraft to provide the range and altitude data to TCAS from traffic that is below the aircraft. In addition to the two TCAS antennas, two antennas are also
required for the Mode S transponder. These antennas enable the Mode S transponder to receive interrogations at 1.03 GHz and to reply for the received interrogations at 1.09 GHz [1]. Fig. 3 shows the Directional Antenna, which contains an array of four passive, steerable, radiating elements mounted at 0°, 90°, 180°, and 270° in relationship with the vertical axis of the antenna. This whole assembly is mounted directly to the fuselage of the aircraft. During Mode S and C interrogation message transmission, the directional antenna transmits 1030 MHz pulses on the four radiating elements. During TCAS receptions, each of the four directional
Fig. 3. Directional antenna.
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TCAS application with better performance than the existing TCAS antenna. The important properties like lightweight, low power consumption, low cost, high reliability, and mobility capability make a microstrip antenna more reliable than other antennas. In this work, Microstrip Patch Antenna has been used due to its extensively uses in commercial and communication systems [9– 10]. Simple patch antennas are well-suited to this application because of its narrow bandwidth characteristics. Frequency Tuning [11], Radiation Beam Shaping, Tuning and Scanning are possible in Microstrip Antenna. These antennas are very much mechanically robust and that is a significant advantage in this application [12]. 2. Antenna configuration and design aspects Fig. 4. Omni-directional antenna.
Table 1 Specifications of existing TCAS antenna. Specifications
Values
Transmitting frequency Receiving frequency V.S.W.R Impedance Power Directional antenna
1.03 GHz 1.09 GHz 1–1.5 50 O 10 Watt null, 1000 W/Peak Polarization : Linear, Gain : 3.6 dB, Side Lobe : 8 dB, Beamwidth : More than 100° Gain : 1 dB
Omni-directional antenna
antenna elements detects 1090 MHz RF signals that are present. The phasing of this received signals are determined by the direction from which the RF energy is received [8]. Fig. 4 shows the Omni-Directional Antenna, which is mounted directly to the underside of the aircraft. It consists of L-band blade type, dipole element. It connects to the TCAS Processor via a single coaxial cable. The other three bottom antenna connectors to the processor are left open. Cable resistance is 50 O or less. The Omni antenna must be certified to C66b, C74, C112 and C119 () [8]. Table 1 summarizes the features of existing TCAS antenna [8]. Fig. 5 shows the radiation pattern of the existing directional and omni-directional TCAS antenna. The 4 monopole stubs are oriented in such a way that the overall radiation pattern can cover all the surveillance area around the aircraft. In this paper, two novel Microstrip Antennas are designed, investigated and presented. The first proposed antenna is a unit element Microstrip Dual Patch Slotted Antenna which is having two Co-axial feeds while the other proposed antenna is a Microstrip Antenna Array, which has total eight Co-axial feeds. The main motive of this work is to implement a new antenna technology in
The proposed Dual Feed Microstrip Dual Patch Slotted Antenna and the Microstrip Antenna Array have been shown in Fig. 6(a) and in Fig. 7(a). These proposed antennas are fabricated on FR4 substrate having dielectric constant of 4.4, thickness 4 mm, and loss tangent 0.002 as shown in Fig. 6(b) and in Fig. 7(b). 2.1. Dual Feed microstrip dual patch slotted antenna In this proposed antenna, two separate copper patches are designed and used as radiators. The conventional Co-axial feeding technique is used here through which patch 1 and 2 is excited and by the mutual coupling of electric fields between each other, the resultant electric fields are generated. The feeds which are applied at the patch 1 and at patch 2 are abbreviated as Port 1 and Port 2 respectively. In this antenna, patch 1 and patch 2 are excited separately but simultaneously. During the simulation, it is observed that, the electric fields are in same phase at both the patches for 1.03 GHz and for 1.09 GHz. Hence, this proposed antenna radiation beam is always similar for both the frequencies. One Shorting Pin or Inductive Post made of copper has been put inside the substrate under the patch 2 and patch 2 also consists of two narrow slots which are identical but opposite in position to each other. The design aspect behind the shorting pin and the slots is the fine frequency tuning of the antenna. Shorting pin provides inductance and slots having length less than k0/4 provide capacitance to the overall antenna. As the frequency goes high, inductive nature dominates and as the frequency goes low, capacitive nature dominates. So, by tuning the diameter of the shorting pin and the length of the slots, proper resonating frequency is achieved here. In this case, these two are actually tuning parameters. A triangular shaped metallic cap made of thin copper foil has been proposed here and it is placed above the patch 2. Here, this triangular cap is used to reduce the back radiation so that, the antenna can radiate properly along the azimuth plane. Conventional Microstrip Patch Antenna always radiates in the broadside direction which is along the elevation plane. In this work, it has
Fig. 5. Radiation pattern of 4 monopoles directional and blade type omni-directional TCAS antenna.
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D. De, P.K. Sahu / Int. J. Electron. Commun. (AEÜ) 71 (2017) 53–71 Table 2 Dimensions of the designed single feed microstrip dual patch slotted antenna. Sl. No.
Section
Dimensions (in millimeters)
1 2 3 4 5 6 7
Substrate (length width) Patch 1 (length width) Patch 2 (length width) Slot 1 & Slot 2 (length width) Diameter of the shorting pin Height of the metallic cap Gap between patch 1 & patch 2
157 110.1 63.5 82.1 62.5 88.1 3 12 7 65 2
Fig. 6a. Simulated dual feed antenna..
Fig. 6b. Fabricated antenna.
Fig. 8. Patch numbering in the antenna.
Length (L) and Width (W) of both the patches have been calculated using the formulae available in the literature [10]. The dimensions of the antenna are given in Table 2.
2.2. Microstrip antenna array model
Fig. 7a. Simulated antenna array.
Fig. 7b. Fabricated antenna.
been proposed that, due to the dual patch radiation and the triangular shaped metallic cap, Microstrip Patch Antenna can also radiate in the end-fire direction [13] which is along the azimuth plane.
This proposed antenna consists of total five patches out of which patch 1 is treated as the main patch and rest of the four patches are the array elements. In this antenna, total eight Coaxial feeds are used to excite all those array elements separately and hence this proposed antenna can also be assumed as a Multiple-Input-Multiple-Output (MIMO) antenna [14]. The numbering of patches and ports in this antenna is very much essential and hence, those have been shown in Fig. 8 and in Fig. 9 respectively. In this design, total five Shorting Pins/Inductive Posts have been used and those are put inside the substrate under all the five patches separately. The patch 1 is also having one square ring slot. Since, patch 1 is responsible for 1.03 GHz while patch 2, 3, 4 and 5 are responsible for 1.09 GHz, the purpose of the slot and the shorting pin is to tune the frequency 1.03 GHz and 1.09 GHz respectively. The tuning concept has been already described in the previous section. 4 Triangular shaped Metallic Caps are placed above those 4 side patches – patch 2, 3, 4 and 5 with the orientation of 0°, 90°, 180° and 270°. As this proposed structure has total eight ports, the antenna radiation beam can be continuously steered in the horizontal plane with 45° step and by exciting proper port combinations. Table 3 shows the details of it. To make an antenna array, the previously designed dual feed microstrip antenna can be used as the unit element in the array.
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D. De, P.K. Sahu / Int. J. Electron. Commun. (AEÜ) 71 (2017) 53–71 Table 4 Dimensions of the designed antenna array. Sl. No.
Section
Dimensions (In millimeters)
1 2 3 4 5 6 7 8
Substrate (length width) patch 1 (length width) patch 2, 3, 4, 5 (length width) Square ring slot (length width thickness) Diameter of the shorting pin under patch 1 Diameter of the shorting pin under patch 2–5 Height of the metallic cap Gap between patch 1 & 4 side patches
224 224 74.1 74.1 88.1 63.95 18 18 2 7 3 65 2
and testing purpose, Agilent E5071C Vector Network Analyzer (VNA) is used. The antenna radiation pattern measurement is performed in a Tapered Anechoic Chamber. 3.1. Dual feed microstrip dual patch slotted antenna Fig. 9. Port numbering in the antenna.
Table 3 Antenna radiation beam orientation with its corresponding port excitations. Ports excitation
Antenna radiation beam orientation
1 and 5 1, 5, 2 and 2 and 6 2, 6, 3 and 3 and 7 3, 7, 4 and 4 and 8 4, 8, 1 and
0° 45° 90° 135° 180° 225° 270° 315°
6 7 8 5
But in that case, the array will be quite large in size and the overall structure will not be compact. So, to make it compact and precise, this antenna array structure is proposed. Now, if the whole antenna is assumed to be a cavity, then the cavity size should be remain same in all aspects so that it can resonate at its proper frequencies. But, in this proposed case, the antenna size is reduced which causes the size reduction of the cavity in terms of length and width. So, to make the proper cavity size, the depth of the structure can be increased. Hence, an additional 4 mm. layer of FR4 substrate is added above the main substrate and under the patch 1. Hence, the total height of the substrate area under patch 1 becomes 8 mm, while the substrate height in the rest of the antenna part remains 4 mm as shown in Fig. 10. The basic Length (L) and Width (W) of all the patches have been calculated using the formulae available in the literature [15]. After that, some of the dimensions are modified little bit as per requirement during designing of the antenna. The dimensions of this proposed antenna are provided in Table 4.
3. Results and discussions These two proposed antennas are simulated, fabricated and measured in order to validate the design technology. The antennas are simulated using CST Studio Suite tool. For the measurement
The simulated and measured S Parameter of this proposed antenna are shown in Fig. 11. In this case, the antenna is designed in such a way that, Port 1 is responsible for 1.03 GHz and Port 2 is responsible for 1.09 GHz. From the simulated results, it is observed that, at 1.04 GHz S11 is 28.8 dB and S22 is 15 dB at 1.095 GHz. Similarly from the measured S parameter plots, it is shown that, S11 is 26.4 dB and S22 is 14.8 dB at 1.045 GHz and at 1.1 GHz respectively. A reasonable matching between the simulated and measured S parameters can be seen. Fig. 12 shows the simulated and measured VSWR of this designed antenna. In simulation, VSWR is 1.08 and 1.5 at 1.04 GHz and at 1.095 GHz respectively. At 1.045 GHz, measured VSWR is 1.2 and at 1.1 GHz, it is 1.55. The simulated and measured antenna impedance characteristics have been shown in Fig. 13. From the simulated result, it can be seen that, impedance is 51 O and 47 O at 1.04 GHz and at 1.095 GHz respectively. At 1.045 GHz, the measured impedance is 54 O and at 1.1 GHz, it is 46 O. Figs. 14 and 15 show the Electric Field distribution in the patches at 1.03 GHz and at 1.09 GHz respectively. From these results, it is understood that, the electric fields are in same phase for both the patches at 1.03 GHz and at 1.09 GHz. Hence, both the patches are having almost equal E-Field distribution at those two resonating frequencies. The electric path length between electric-field-zero lines corresponding to the blue colored ones in the two patches as seen from the Figs. 14 and 15, have been calculated to be 0.26 k at 1.03 GHz and 0.27 k at 1.09 GHz (approx.). Since, the path length is almost same for these two frequencies, it explains that both the patches together will radiate in such a way that the power patterns at two said frequencies will be nearly same. From the Insertion-loss plot as shown in Fig. 16, it has been observed that, S12 is around 5 dB at both the frequencies. As S12 is close to 0 dB, it is understood that the two ports are well coupled, as it is expected as the orientation of the two patches are 180° to each other with regard to the feed locations. The phase values of S12 phase plot in Fig. 17, are 17° at 1.03 GHz and 60° at 1.09 GHz. These phase values imply that the port offers two reactances at two different frequencies. The feed point is located at 0.05 k for 1.03 GHz and at 0.83 k for 1.09 GHz (approx.)
Fig. 10. Side view of additional 4 mm FR4 substrate layer under patch 1.
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Fig. 11. Simulated and measured S parameters of the Dual Feed Microstrip Antenna.
Fig. 12. Simulated and measured VSWR of the Dual Feed Microstrip Antenna.
Fig. 13. Simulated and measured impedance of the Dual Feed Microstrip Antenna.
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Fig. 14. E-Field distribution of the Dual Feed Microstrip Antenna at 1.03 GHz.
Fig. 15. E-Field distribution of the Dual Feed Microstrip Antenna at 1.09 GHz.
with respect to the electric-field-zero line at patch 1, whereas, the feed point is physically located at 0.08 k for the design frequency. So, the patch resonator works in such a way that, there are phase lead and phase lag at 1.03 GHz and at 1.09 GHz respectively. For S21, the discussion will remain same. The simulated 3D Radiation Pattern of the proposed antenna at both frequencies are shown in Fig. 18 and in Fig. 19. It is observed
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that, at 1.03 GHz and at 1.09 GHz, the antenna is radiating in the end-fire direction that is along the azimuth plane. The simulated cum measured 2D Polar Plots at two resonating frequencies 1.03 GHz and 1.09 GHz of the proposed antenna have been presented in Fig. 20. Fig. 21 shows the simulated and measured Cartesian Power Pattern Plots for both the frequencies. As from the simulation results, it is observed that, this proposed antenna provides the end-fire radiation, hence, during the antenna pattern measurement for 1.03 GHz and for 1.09 GHz, the antenna is mounted horizontally that is along the azimuth plane. From these plots, it can be seen that, the measured radiation patterns are in quite proper match with the simulated results. At both the frequencies, the peak of the radiation beam is at around 0° in the azimuth plane. The measured radiation pattern of this proposed antenna has little bit of wide beamwidth and high side lobe level as compared to that of simulated ones. The simulated antenna beamwidth is around 79° while the measured beamwidth is around 84°. Since, this proposed antenna provides narrow beamwidth in comparison to the existing TCAS antenna, the high Directivity is also achieved. Fig. 22 shows the simulated and measured peak gain of the proposed Dual Feed Microstrip Dual Patch Slotted antenna. From these results, it is shown that, at 1.03 GHz the simulated and measured maximum gain is 6.04 dB and 6.08 dB respectively. Similarly, for 1.09 GHz, the simulated and measured maximum gain is 6.34 dB and 6.1 dB respectively. The simulated radiation efficiency of the antenna is shown in Fig. 23. It is understood that, the radiation efficiency is 57% and 52% for 1.03 GHz and for 1.09 GHz respectively. The simulated temperature distribution in the antenna substrate for both the frequencies has been presented in Fig. 24. During the simulation, 10 W is given to the both antenna port 1 and 2 as input power. It is observed that, the maximum raised temperature in that circumstances is 2.7 °C for 1.03 GHz and for 1.09 GHz. In general, the temperature index of FR4 Substrate is 140 °C. So, in this case, the maximum temperature is well under the threshold level. A CAD Model of Boeing-787 Aircraft has been designed here. The proposed antenna is mounted at the top of the aircraft and then it has been simulated with the aircraft model using CST Studio Suite tool and the results are shown in Fig. 25 and in Fig. 26. The purpose of this study is to observe the antenna installed performance on a large electrical structure. To simulate this large aircraft structure, the Asymptotic Solver is used instead of FIT (Finite Integration Technique) Solver. Because, this numerical technique can
Fig. 16. Simulated and measured insertion loss of the Dual Feed Microstrip Antenna.
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Fig. 17. Phase of simulated and measured insertion loss in the Dual Feed Microstrip Antenna.
Fig. 18. 3D radiation pattern of the Dual Feed Microstrip Antenna at 1.03 GHz.
Fig. 19. 3D radiation pattern of the Dual Feed Microstrip Antenna at 1.09 GHz.
simulate large problems with the available hardware resources by using acceleration techniques like multi core/CPU computing or distributed computing. In addition, the separation of complex problems into multiple parts, with the link between them being defined by equivalent near or far field sources, can dramatically speed up installed antenna performance analysis. From the results, it is observed that, the beam pattern has not been distorted due to this large electrical structure. 3.2. Microstrip antenna array structure The simulated and measured S Parameter of this proposed antenna are shown in Fig. 27. In this case, the antenna is designed in such a way that, Port 1, 2, 3, 4 are responsible for 1.03 GHz and Port 5, 6, 7, 8 are responsible for 1.09 GHz. From the simulated results, it is observed that, at 1.03 GHz S11, S22, S33, S44 are 48.1 dB and S55, S66, S77, S88 are 25.8 dB at 1.09 GHz. Similarly from the measured S parameter plots, it is shown that, S11, S22, S33, S44 are 28.3 dB and S55, S66, S77, S88 are 33.4 dB at
Fig. 20. Simulated and measured polar plots at both the resonating frequencies of the Dual Feed Microstrip Antenna.
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Fig. 21. Simulated and measured Cartesian power pattern plots at both the resonating frequencies of the Dual Feed Microstrip Antenna.
Fig. 22. Antenna gain vs. frequency plots of the Dual Feed Microstrip Antenna.
Fig. 23. Antenna radiation efficiency plot of the Dual Feed Microstrip Antenna.
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Fig. 24. Temperature distribution in the substrate of the Dual Feed Microstrip Antenna at 1.03 GHz and at 1.09 GHz.
1.035 GHz and at 1.1 GHz respectively. A reasonable matching between the simulated and measured S parameters can be seen. Fig. 28 shows the simulated and measured VSWR of this designed antenna. In simulation, VSWR is 1.007 and 1.2 at 1.03 GHz and at 1.09 GHz respectively. At 1.035 GHz, measured
VSWR is 1.15 and at 1.1 GHz, it is 1.06. The simulated and measured antenna impedance characteristics have been shown in Fig. 29. From the simulated result, it can be seen that, impedance is 50.4 O and 48.5 O at 1.03 GHz and at 1.09 GHz respectively. At 1.035 GHz, the measured impedance is 52.6 O and at 1.1 GHz, it is 49.3 O. Figs. 30 and 31 show the Electric Field distribution in the patches at 1.03 GHz and at 1.09 GHz respectively. From these results, it can be seen that, how the electric fields are shifting from patches to patches as the port excitations are changed simultaneously. It is also understood that, E-Field distribution in the antenna is same for both the frequencies since there is no issue of phase differences between the patches. The simulated 3D Radiation Pattern of the proposed antenna at both frequencies are shown in Fig. 32 and in Fig. 33. It is observed that, at 1.03 GHz and at 1.09 GHz, the antenna is radiating in the end-fire direction that is along the azimuth plane. As, the feed excitation are changed from one combination to another, the antenna radiation beam is continuously steered from 0° to 360° in the azimuth plane. Hence, the proposed antenna can cover its whole around 360° surveillance area. The simulated 2D Polar Plots at two resonating frequencies 1.03 GHz and 1.09 GHz of the proposed antenna have been presented in Fig. 34 and in Fig. 35 respectively. Figs. 36 and 37 show
Fig. 25. Dual Feed Microstrip Antenna installed performance at 1.03 GHz.
Fig. 26. Dual Feed Microstrip Antenna installed performance at 1.09 GHz.
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Fig. 27. Simulated and measured S parameters of the proposed antenna array.
Fig. 28. Simulated and measured VSWR of the proposed antenna array.
Fig. 29. Simulated and measured impedance of the proposed antenna array.
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Fig. 30. E-Field distribution in the proposed antenna array at 1.03 GHz.
Fig. 31. E-Field distribution in the proposed antenna array at 1.09 GHz.
the simulated Cartesian Power Pattern Plots for both the frequencies. As from the simulation results, it is observed that, this proposed antenna provides the end-fire radiation, hence, during the antenna pattern measurement for 1.03 GHz and for 1.09 GHz, the antenna is mounted horizontally that is along the azimuth plane. At both the frequencies, the peak of the radiation beam is at around 0° in the azimuth plane when port 1 and 5 are simultaneously excited. After that when, port 1, 5, 2 and 6 are simultaneously excited, the radiation beam is shifted by 45° in the azimuth plane. Like this way, the beam is steered in the azimuth plane as the port excitations are changed. Hence, this proposed antenna cover the whole 360° surveillance region. The simulated antenna beamwidth is around 52° and side lobe level is 15.8 dB. As this proposed antenna provides narrow beamwidth in comparison to the dual feed antenna and the existing TCAS antenna, much high Directivity is also achieved in this antenna.
From the simulation results, it is observed that, to get the diagonal radiation pattern at both the frequencies, four antenna ports need to be excited simultaneously. But this facility was not available in the laboratory where the antenna power pattern measurement has been performed. Hence, the conventional radiation pattern measurement was not possible for this antenna and therefore, an alternative way of power pattern measurement has been executed to validate all those simulated results. First, the proposed antenna and a transmitting double-ridge horn antenna is kept in the Line of Sight (LOS) to each other. The arrangement has been done in such a way that, port 1 and 5 i.e.; patch 1 and 2 are in the LOS with the transmitting antenna. The transmitting antenna is radiating 10 dBm power. A pictorial representation of this arrangement is shown in Fig. 38. In this situation, without rotating the proposed antenna, the received power by all the ports have been measured, which is shown in Fig. 39.
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Fig. 32. 3D radiation patterns of the proposed antenna array at 1.03 GHz.
Fig. 33. 3D radiation patterns of the proposed antenna array at 1.09 GHz.
Fig. 34. Simulated polar plots of the proposed antenna array at 1.03 GHz.
Fig. 35. Simulated polar plots of the proposed antenna array at 1.09 GHz.
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Fig. 36. Simulated Cartesian power patterns of the proposed antenna array at 1.03 GHz.
Fig. 37. Simulated Cartesian power patterns of the proposed antenna array at 1.09 GHz.
Fig. 38. First arrangement during antenna power reception measurement.
These received powers are converted to the Electrical Fields using the formulas available in the literature [16] [17]. All these values have been tabulated in Table 5. From the Table, it can be seen that, the maximum electrical field is received by port 1 and 5. The total received E-Field by port 1 and 5 can be calculated by doing vector addition of E1 and E5. As, vector
E1 and E5 are in same direction, simple addition can be done. Hence, the resultant vector is given by, E1–5 = E1 + E5 = 3.81 V/m and it is maximum as compared to other received E-Fields. The resultant E-Field E1–5 is in the same direction as that of E1 and E5. The vector representation of these electrical fields is shown in Fig. 40.
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Fig. 39. Measured received power by all the ports when the proposed antenna is at 0° with the transmitting antenna.
Table 5 Measured received power and electric fields by the antenna ports for the first arrangement. Port number
Received power (in dBm)
Electric field (in V/m)
1 5 2 6 3 7 4 8
P1 = P5 = P2 = P6 = P3 = P7 = P4 = P8 =
E1 = 1.898 E5 = 1.912 E2 = 0.182 E6 = 0.192 E3 = 0.010 E7 = 0.014 E4 = 0.188 E8 = 0.193
5.44 5.38 22.96 22.65 39.88 39.17 22.78 22.65
The corresponding power of the electric field E1–5 is denoted by P1–5 and that can be calculated by using the same formulas as used before. The calculated value of P1–5 is 0.4 dBm. Hence, the total received power by the antenna port 1 and 5 is 0.4 dBm. The same measurement procedure has been repeated by keeping port 2 & 6, 3 & 7 and 4 & 8 in the LOS with the transmitting antenna. For all these cases, the total received power is around 0.4 dBm. After this measurement process, the proposed antenna is then rotated 45° clockwise with respect to LOS. In this case also, the transmitting antenna is radiating 10 dBm power. A pictorial representation of this arrangement is shown in Fig. 41. In this situation, without rotating the antenna, the received power by all the ports have been measured which is shown in Fig. 42.
Fig. 40. E-Field vector addition and its representation for the first arrangement.
As similar with the first case, these received powers are converted to the Electrical Fields using the formulas available in the literature [16] [17]. All these values have been tabulated in Table 6. From the Table, it can be seen that, the maximum electrical field is received by port 1, 5, 2 and 6. The total received E-Field by these ports can be calculated by doing vector addition of E1–5 and E2–6. E1–5 is the total electric field, received by port 1 and 5 and it is given by, E1–5 = E1 + E5 = 1.893 V/m. Similarly, we will get E2–6
Fig. 41. Second arrangement during antenna power reception measurement.
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Fig. 42. Measured received power by all the ports when the proposed antenna is at 45° with the transmitting antenna.
Table 6 Measured received power and electric fields by the antenna ports for the second arrangement. Port number
Received power (in dBm)
Electric field (in V/m)
1 5 2 6 3 7 4 8
P1 = P5 = P2 = P6 = P3 = P7 = P4 = P8 =
E1 = 0.937 E5 = 0.956 E2 = 0.999 E6 = 0.989 E3 = 0.086 E7 = 0.098 E4 = 0.092 E8 = 0.099
14.57 14.31 14.92 14.70 29.96 29.12 29.56 29.04
Fig. 43. E-Field vector addition and its representation for the second arrangement.
which is E2–6 = E2 + E6 = 1.989 V/m. The resultant E-Field vector of these two E-Fields E1–5 and E2–6 is denoted by E1–5, 2–6 and that can be calculated by applying the vector addition rule of parallelogram. The vector representation of these electrical fields is shown in Fig. 43. E1–5, 2–6 can be written as, E1–5, 2–6 = ((E1–5)2 + (E2–6)2)1/2 = 2.75 V/m and it is maximum as compared to other received E-Fields. According this vector addition rule, the direction of the resultant E-Field vector E1–5, 2–6 is in the diagonal direction with respect to the antenna structure as shown in Fig. 43. The corresponding power of the E-Field E1–5, 2–6 is denoted by P1–5, 2–6 and that can be calculated by using the same formulas as used before. The calculated value of P1–5, 2–6 is 3.23 dBm. Hence, the total received power by the antenna ports 1, 5, 2 and 6 is 3.23 dBm. P2–6, 3–7, P3–7, 4–8 and P4–8, 1–5 have been also measured by repeating the same measurement procedure with proper antenna orientation and for all these case the total received power is around 3.23 dBm in the diagonal direction. The value of the resultant power vectors P1–5, 2–6, P2–6, 3–7, P3–7, 4–8 and P4–8, 1–5 should be close to 0.4 dBm. But, as their value is 3.23 dBm, which means, 2.83 dBm power loss will be produced when, there will be simultaneously 4 ports excitation in the antenna. As a whole, by performing this power pattern testing of the proposed antenna, it can be understood that, the antenna radiates in the diagonal direction along the azimuth plane when four proper antenna ports are excited simultaneously and it somehow validates the simulation results.
Fig. 44. Simulated antenna gain vs. frequency plots of the proposed antenna array.
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Fig. 45. Antenna gain when 2 ports are excited.
Fig. 48. Temperature distribution with 4 ports excitation in the proposed antenna array.
Fig. 46. Antenna radiation efficiency plots of the proposed antenna array.
for 1.09 GHz respectively. When, four antenna ports are simultaneously excited, the radiation efficiency is 69% and 67% for 1.03 GHz and for 1.09 GHz respectively. When two ports simultaneously excited, the simulated temperature distribution in the antenna substrate for both the frequencies has been presented in Fig. 47. During the simulation, 10 W is given to the both antenna port 1 and 5 as input power. It is observed that, the maximum raised temperature in that circumstances is 3.7 °C for 1.03 GHz and for 1.09 GHz. Fig. 48 shows the simulated temperature distribution at both the frequencies, when four ports are simultaneously excited. In this case, the maximum raised temperature is 4.7 °C for 1.03 GHz and for 1.09 GHz. For this antenna also, general, the maximum temperature is well under the temperature index of FR4 Substrate. The similar CAD Model of Boeing-787 Aircraft has been also used here. The proposed antenna is placed at the top of the aircraft and then it is simulated along the aircraft model using CST Studio Suite tool and the results are shown in Fig. 49 and in Fig. 50 with proper port excitation combination. From the results, it is observed that, in this case also, the beam pattern has not been distorted due to this large electrical aircraft body. 3.3. Performance comparison between the proposed and existing TCAS antenna
Fig. 47. Temperature distribution with 2 ports excitation in the proposed antenna array.
Fig. 44 shows the simulated peak gain of the proposed antenna. Fig. 45 shows the simulated and measured peak gain of the antenna when two ports are excited. From these results, it is shown that, at 1.03 GHz the simulated and measured maximum gain is 7.46 dB and 7.3 dB respectively. Similarly, for 1.09 GHz, the simulated and measured maximum gain is 6.93 dB and 6.8 dB respectively. The simulated radiation efficiency of the antenna is shown in Fig. 46. It is understood that, when two ports are simultaneously excited, the radiation efficiency is 65% and 59% for 1.03 GHz and
Table 7 lists the comparison of the proposed antennas to the existing TCAS/ACAS antenna. From this table, it is found that, the proposed antenna provides better performance compared to the existing monopole TCAS antenna and hence, this design technique can be applied with some further development in the microstrip antenna to make it a good candidate for this application. Comparing the S parameters as seen from the Table 7, it is clear that the frequency sensitivity is better for the proposed antenna. The VSWR and impedance values of the proposed antenna are well matched like that of existing antenna. It is seen that the Gain of the proposed antenna is roughly doubled which implies the power expenditure during signal transmission is economized. From Table 7, it is observed that the Beamwidth of the proposed antenna is roughly reduced to half as compared to that of the existing TCAS antenna. Due to this narrow Beamwidth, resolving the location of the intruder aircraft in terms of bearing angle is much better and, hence, the resolution power of this proposed antenna is better than the existing one. The current TCAS antenna which is fitted in the aircraft has the Side Lobe Level of 8 dB, as shown in Table 7, while the Side Lobe
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Fig. 49. Antenna Array installed performance at 1.03 GHz.
Fig. 50. Antenna Array installed performance at 1.03 GHz.
Table 7 Performance comparison between proposed antennas and conventional TCAS antenna. Parameters
Dual feed antenna
Antenna array With 2 ports
Tx. Freq. & Rx. Freq. (In GHz) S Parameter at Tx. Freq. & Rx. Freq. (In dB) VSWR at Tx. Freq. & Rx. Freq. Impedance at Tx. Freq. & Rx. Freq. (In Ohm) Gain at Tx. Freq. & Rx. Freq. (In dB) Beamwidth at Tx. Freq. & Rx. Freq. Side Lobe Level at Tx. Freq. & Rx. Freq. (In dB) Radiation Pattern Polarization Efficiency at Tx. Freq. & Rx. Freq. Max. Temp. for 10 Watt I/P Power
Simulated Measured Simulated Measured Simulated Measured Simulated Measured Simulated Measured Simulated Measured Simulated Measured
1.04 & 1.095 1.045 & 1.1 28.8 & 15 26.4 & 14.8 1.08 & 1.5 1.2 & 1.55 51 & 47 54 & 46 6.04 & 6.34 6.08 & 6.1 79° & 77° 84° & 81° 14.3 & 13.9 10.4 & 9.5 Directional Linear 54% & 40% 1.67 °C
1.03 & 1.09 1.035 & 1.1 48.1 & 25.8 28.3 & 33.4 1.007 & 1.2 1.15 & 1.06 50.4 & 48.5 52.6 & 49.3 7.5 & 6.9 7.3 & 6.8 52° & 53° – 15.2 & 15.0 – Directional Linear 65% & 59% 3.7 °C
Existing TCAS antenna With 4 ports 1.03 & 1.09 15 1–1.5 50 7.1 & 6.7 – 53° & 54° – 14.9 & 14.2 –
69% 67% 4.7 °C
3.6 >100° 8 Directional Linear – –
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Level of the proposed antenna is around 15 dB which is almost half of the said magnitude of the existing antenna. Hence, the sensitivity of the proposed antenna is expected to be better. Due to this low level of side lobe in the proposed antenna and if this low level is considered to be the threshold value of return signal, the covering distance will likely to become more compared to the existing antenna. As the covering distance is larger, more number of aircraft which were hither to unseen in the existing system, can be detected. So, larger number of aircraft can be tracked for collision probability with respect to the own aircraft. 4. Conclusions This paper presents a designing technique for Microstrip Antenna which can be used as an aircraft tracking antenna for TCAS/ACAS application. It is shown that, the proposed microstrip antenna array provides a better performance like high gain, good directivity and narrow beamwidth as compared to that of existing monopole TCAS/ACAS antenna. The whole antenna array structure can cover the 360° surveillance region around the aircraft with proper excitation of the corresponding ports. Hence, TCAS system can measure the bearing angle of an intruder aircraft while tracking. This proposed antenna is also very much light in weight and cost effective. It has been designed using the dielectric most suited for the intended frequency of operation. The length and width of the patch has been calculated comply with the space bracket available on the fuselage of the aircraft. The above designed antenna exhibits the parameters similar to the original antenna. The proposed antenna will be fitted on the fuselage at the top of the aircraft. This fitting is aided by 8 round headed bolts so that the antenna should not fall of. The antenna has been also simulated with those 8 bolts and it has been observed that there is no additional effect in the power pattern, S parameter, VSWR and Impedance plots. The proposed antenna will not be left open at the top of the aircraft as shown in the paper. It will be encapsulated inside an aerodynamically shaped enclosure, known as ‘Radome’. For a high speed carrier a radome or an antenna of lower profile must be used to reduce the air drag. The existing TCAS Antenna is also enclosed inside a radome. Yet, the enclosure has some air-drag, however small it may be. Originally the TCAS antenna elements were having large height, however by loading the elements, the height of the elements and consequently the height of the encapsulation above the aircraft surface have been reduced so as to improve the airdrag performance [2]. The current research focus is on the designing of the antenna element. The 65 mm. height of the metallic cap will cause the height of its encapsulation to be at least of the same amount. Presently there are various antennas, i.e. Automatic Direction Finder (ADF) antenna [18], GPS antenna etc. which are fitted at the top of the aircraft like TCAS antenna. All these are encapsulated with radomes having antenna elements within each of these. Properly designed shaped-encapsulation that hides the metallic cap of 65 mm. will optimize the drag performance. Apart from the design
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considerations it also saves on the maintenance front as it is not directly exposed in the line of the airflow. Hence, this Antenna can be embedded in the body of the aircraft, so that it will offer minimum resistance to the airflow. Research on the designing of such shaped encapsulation is still going on. Therefore, with these proposed implementations in the system, this TCAS antenna would be expected to meet the requirements of the advanced avionics standards in terms of design simplicity, lightweight and high performance. Acknowledgements The authors would like to thank Space Application Centre (SAC), ISRO for proving the chance to explore their excellent measurement facilities during this research work. The authors are also very much glad to the college authority of National Institute of Technology, Rourkela, India for various financial support during this research work. References [1] Steve Henely, Rockwell Collins, The avionics hand book, 18-TCAS II, CRC Press LLC, 2001. [2] Harman WH. TCAS: a system for preventing midair collisions. Lincoln Lab J 1989;2:437–58. [3] U.S. Department of Transportation (Federal Aviation Administration), ‘‘Introduction to TCAS II”, Version 7.1, February 28, 2011 [4] Welch JD, Orlando VA. Traffic alert and collision avoidance system (TCAS): a functional overview of minimum TCAS II. USA: Lincoln Laboratory, M.I.T.; April 8, 1983. Project Report ATC-119. [5] Kuchar James K, Drumm Ann C. The traffic alert and collision avoidance system. Lincoln Lab J 2007;16. [6] Sampath KS, Rojas RG, Burnside WD. Analysis & simulation of collision voidance TCAS antennas mounted on aircraft. In: IEEE Conference, London; 1991, vol. 2, p. 948–51. [7] Harman WH, Wood ML. Triangle TCAS antenna. USA: Lincoln Laboratory, M.I. T.; 2011. Project Report ATC-380. [8] TCAS S72-1735-25, AIRNC 735. Sensor Systems Inc., Aircraft antennas since 1961. [9] Umar Khan Qasim, Bin Ihsan Mojeeb. Higher order mode excitation for high gain microstrip patch antenna. AEU Int J Electron Commun November 2014;68 (11):1073–7. [10] Shanmuganantham T, Raghavan S. Design of a compact broadband microstrip patch antenna with probe feeding for wireless applications. AEU Int J Electron Commun 2009;63(8):653–9. [11] Soon-Soo Oh, Park Wook-Ki, Jung Young-Bae, Choi Tael-II, Lee Young-Hwan. Frequency-tunable open-ring microstrip antenna with optimally-positioned varactors for radiated-power in situ measurements. AEU Int J Electron Commun September 2014;68(9):841–5. [12] Orban D, Moernaut GJK. The basics of patch antennas, orban microwave products, September 2005. [13] Wang Chien-Jen, Li Simon C, Sun Te-Liang, Lin Chun-Min. A wideband steppedimpedance open-slot antenna with end-fire directional radiation characteristics. AEU Int J Electron Commun 2013;67(3):175–81. [14] Kumar Ghosh Chandan. A compact 4-channel microstrip MIMO antenna with reduced mutual coupling. AEU Int J Electron Commun 2016;70(7):873–9. [15] Balanies CA. Antenna theory: analysis & design. 2nd ed. John Wiley & Sons Inc; 1997. [16] Fontolliet PG. Telecommunications systems, Section 3.9. PPUR Presses Polytechnics; 1996. [17] Godlewski Philippe, Tabbane Sami, Lagrange Xavier. Reseaux GSM-DCS. 4th ed. Hermes Science Publications; 1999. [18] Honeywell. Installation manual Bendix/King KR 87 automatic direction finder, May 6, 2006. Manual Number 006-00184-0006.
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International Journal of Electronics and Communications (AEÜ) journal homepage: www.elsevier.com/locate/aeue
Regular paper
A Novel Approach towards the Designing of an Antenna for Aircraft Collision Avoidance System Debajit De ⇑, Prasanna Kumar Sahu Department of Electrical Engineering, National Institute of Technology, Rourkela, India
a r t i c l e
i n f o
Article history: Received 21 September 2016 Revised 31 October 2016 Accepted 2 November 2016
Keywords: TCAS/ACAS Monopole antenna Microstrip Antenna S Parameter VSWR Impedance E-Field Gain Beamwidth Side lobe
a b s t r a c t TCAS/ACAS (Traffic/Aircraft Collision Avoidance System) is an airborne system designed to increase cockpit awareness of nearby aircraft and to service as a last defense against mid-air collisions between the aircrafts. In the existing system, four monopole stub elements are used as TCAS directional antenna and one blade type element is used as TCAS Omni-directional antenna. The transmission and reception frequencies of the TCAS antenna are 1.03G Hz and 1.09G Hz respectively. The existing TCAS antenna has some drawbacks such as low gain, large beamwidth, frequency and beam tuning/scanning issues etc. Antenna issues like unwanted signals reception may create difficulties in identifying the possible threats. In this paper, the focus is on the design and development of a novel Microstrip Antenna Array which can be used for TCAS/ACAS application. Two proposed antenna models are presented here – a Unit Element Dual Feed Microstrip Dual Patch Slotted Antenna and a Compact Microstrip Antenna Array. These are designed in CST tool to meet the current needs of aircraft Collision Avoidance System and to overcome the possible limitations associated with the existing techniques. The performance and other antenna characteristics have been explored from the simulation results followed by the antenna fabrication and measurement. A good Reflection Coefficient and VSWR with proper 50 X Impedance Matching, narrow Beamwidth, perfect Directional Radiation Pattern, high Gain and Directivity at the operating frequencies make this proposed antenna a good candidate for TCAS application. The proposed antenna would be expected to meet the requirements of the advanced avionics standards in terms of design simplicity, lightweight and high performance. Ó 2016 Elsevier GmbH. All rights reserved.
1. Introduction The Aircraft Collision Avoidance System provide a solution of reducing the risk of mid-air collisions between aircraft. TCAS is a family of airborne systems that function independently of ground-based air traffic control (ATC) to provide collision avoidance protection. The TCAS concept makes use of the radar beacon transponders carried by aircraft for ground ATC purposes and provides no protection against aircraft that do not have an operating transponder [1]. After many years of extensive analysis, development, and flight evaluation by the Federal Aviation Administration (FAA), Civil Aviation Authorities (CAAs), and the aviation industries, Traffic Collision Avoidance System (TCAS) was developed to reduce the risk of mid-air collisions between aircraft [2]. The main feature of TCAS, which was first proposed by Dr. John S. Morell in 1955, is that the function is according to time criteria and not ⇑ Corresponding author. E-mail addresses: (P.K. Sahu).
[email protected]
(D.
http://dx.doi.org/10.1016/j.aeue.2016.11.002 1434-8411/Ó 2016 Elsevier GmbH. All rights reserved.
De),
[email protected]
the distance. From several successive replies, TCAS calculates a time to reach the CPA (Closest Point of Approach) with the intruder, by dividing the range by the closure rate [3]. In 1983, J. D. Welch and V. A. Orlando of Lincoln Laboratory in M.I.T, USA had published first official report on functional overview of TCAS system [4]. In this report, the presented a review of the functions performed by any collision avoidance system and then a definition of the way in which these functions are implemented in the TCAS II. This section concludes with a summary of TCAS II design parameters. TCAS/ACAS employs radio signals for surveillance of nearby aircraft, and in dangerous encounters, it warns the aircraft pilot by means of cockpit displays and auditory alarms. To detect the presence of nearby aircraft, TCAS transmits interrogations at a steady rate, nominally once per second, and employs a receiver for detecting the replies to these interrogations from the transponders on nearby aircraft as shown in Fig. 1. Fig. 2 shows the basic block diagram of overall TCAS system. TCAS consists of the Mode S/TCAS Control Panel, the Mode S Transponder, the TCAS Computer, Antennas, Traffic and Resolution Advisory Displays, and an Aural
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Fig. 1. Air-to-air surveillance.
Fig. 2. TCAS/ACAS system block diagram.
Annunciator. Control information from the Mode S/TCAS Control Panel is provided to the TCAS Computer via the Mode S Transponder [5]. In 1991, K. S. Sampath, R. G. Rojas and W. D. Burnside of ElectroScience Laboratory in Ohio State University, USA had discussed 2 different implementations of the TCAS Antenna [6]. In this paper, 8 elements circular antenna array with a monopulse principle and a 4 element antenna with an amplitude comparison scheme are presented. The purpose of this paper is to study the effect of aircraft itself on the performance of TCAS in estimating the bearing angle of the intruder aircraft. The performance of the 4 elements TCAS was not as good as the 8 elements but the 4 elements amplitude system is much simpler in its implementation than the 8 elements monopulse system. In 2011, W. H. Harman and M. L. Wood of Lincoln Laboratory in M.I.T, USA have proposed a Triangle TCAS Antenna [7]. This report describes an antenna of 3 elements in the shape of a triangle. It illustrates the concept of the triangle antenna as compared with the conventional TCAS antenna. The triangle antenna offers an advantage in angle of arrival accuracy as affected by receiver noise. Presently, TCAS/ACAS uses a directional antenna, mounted on top of the aircraft. An Omni-directional transmitting and receiving antenna is mounted at the bottom of the aircraft to provide the range and altitude data to TCAS from traffic that is below the aircraft. In addition to the two TCAS antennas, two antennas are also
required for the Mode S transponder. These antennas enable the Mode S transponder to receive interrogations at 1.03 GHz and to reply for the received interrogations at 1.09 GHz [1]. Fig. 3 shows the Directional Antenna, which contains an array of four passive, steerable, radiating elements mounted at 0°, 90°, 180°, and 270° in relationship with the vertical axis of the antenna. This whole assembly is mounted directly to the fuselage of the aircraft. During Mode S and C interrogation message transmission, the directional antenna transmits 1030 MHz pulses on the four radiating elements. During TCAS receptions, each of the four directional
Fig. 3. Directional antenna.
D. De, P.K. Sahu / Int. J. Electron. Commun. (AEÜ) 71 (2017) 53–71
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TCAS application with better performance than the existing TCAS antenna. The important properties like lightweight, low power consumption, low cost, high reliability, and mobility capability make a microstrip antenna more reliable than other antennas. In this work, Microstrip Patch Antenna has been used due to its extensively uses in commercial and communication systems [9– 10]. Simple patch antennas are well-suited to this application because of its narrow bandwidth characteristics. Frequency Tuning [11], Radiation Beam Shaping, Tuning and Scanning are possible in Microstrip Antenna. These antennas are very much mechanically robust and that is a significant advantage in this application [12]. 2. Antenna configuration and design aspects Fig. 4. Omni-directional antenna.
Table 1 Specifications of existing TCAS antenna. Specifications
Values
Transmitting frequency Receiving frequency V.S.W.R Impedance Power Directional antenna
1.03 GHz 1.09 GHz 1–1.5 50 O 10 Watt null, 1000 W/Peak Polarization : Linear, Gain : 3.6 dB, Side Lobe : 8 dB, Beamwidth : More than 100° Gain : 1 dB
Omni-directional antenna
antenna elements detects 1090 MHz RF signals that are present. The phasing of this received signals are determined by the direction from which the RF energy is received [8]. Fig. 4 shows the Omni-Directional Antenna, which is mounted directly to the underside of the aircraft. It consists of L-band blade type, dipole element. It connects to the TCAS Processor via a single coaxial cable. The other three bottom antenna connectors to the processor are left open. Cable resistance is 50 O or less. The Omni antenna must be certified to C66b, C74, C112 and C119 () [8]. Table 1 summarizes the features of existing TCAS antenna [8]. Fig. 5 shows the radiation pattern of the existing directional and omni-directional TCAS antenna. The 4 monopole stubs are oriented in such a way that the overall radiation pattern can cover all the surveillance area around the aircraft. In this paper, two novel Microstrip Antennas are designed, investigated and presented. The first proposed antenna is a unit element Microstrip Dual Patch Slotted Antenna which is having two Co-axial feeds while the other proposed antenna is a Microstrip Antenna Array, which has total eight Co-axial feeds. The main motive of this work is to implement a new antenna technology in
The proposed Dual Feed Microstrip Dual Patch Slotted Antenna and the Microstrip Antenna Array have been shown in Fig. 6(a) and in Fig. 7(a). These proposed antennas are fabricated on FR4 substrate having dielectric constant of 4.4, thickness 4 mm, and loss tangent 0.002 as shown in Fig. 6(b) and in Fig. 7(b). 2.1. Dual Feed microstrip dual patch slotted antenna In this proposed antenna, two separate copper patches are designed and used as radiators. The conventional Co-axial feeding technique is used here through which patch 1 and 2 is excited and by the mutual coupling of electric fields between each other, the resultant electric fields are generated. The feeds which are applied at the patch 1 and at patch 2 are abbreviated as Port 1 and Port 2 respectively. In this antenna, patch 1 and patch 2 are excited separately but simultaneously. During the simulation, it is observed that, the electric fields are in same phase at both the patches for 1.03 GHz and for 1.09 GHz. Hence, this proposed antenna radiation beam is always similar for both the frequencies. One Shorting Pin or Inductive Post made of copper has been put inside the substrate under the patch 2 and patch 2 also consists of two narrow slots which are identical but opposite in position to each other. The design aspect behind the shorting pin and the slots is the fine frequency tuning of the antenna. Shorting pin provides inductance and slots having length less than k0/4 provide capacitance to the overall antenna. As the frequency goes high, inductive nature dominates and as the frequency goes low, capacitive nature dominates. So, by tuning the diameter of the shorting pin and the length of the slots, proper resonating frequency is achieved here. In this case, these two are actually tuning parameters. A triangular shaped metallic cap made of thin copper foil has been proposed here and it is placed above the patch 2. Here, this triangular cap is used to reduce the back radiation so that, the antenna can radiate properly along the azimuth plane. Conventional Microstrip Patch Antenna always radiates in the broadside direction which is along the elevation plane. In this work, it has
Fig. 5. Radiation pattern of 4 monopoles directional and blade type omni-directional TCAS antenna.
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D. De, P.K. Sahu / Int. J. Electron. Commun. (AEÜ) 71 (2017) 53–71 Table 2 Dimensions of the designed single feed microstrip dual patch slotted antenna. Sl. No.
Section
Dimensions (in millimeters)
1 2 3 4 5 6 7
Substrate (length width) Patch 1 (length width) Patch 2 (length width) Slot 1 & Slot 2 (length width) Diameter of the shorting pin Height of the metallic cap Gap between patch 1 & patch 2
157 110.1 63.5 82.1 62.5 88.1 3 12 7 65 2
Fig. 6a. Simulated dual feed antenna..
Fig. 6b. Fabricated antenna.
Fig. 8. Patch numbering in the antenna.
Length (L) and Width (W) of both the patches have been calculated using the formulae available in the literature [10]. The dimensions of the antenna are given in Table 2.
2.2. Microstrip antenna array model
Fig. 7a. Simulated antenna array.
Fig. 7b. Fabricated antenna.
been proposed that, due to the dual patch radiation and the triangular shaped metallic cap, Microstrip Patch Antenna can also radiate in the end-fire direction [13] which is along the azimuth plane.
This proposed antenna consists of total five patches out of which patch 1 is treated as the main patch and rest of the four patches are the array elements. In this antenna, total eight Coaxial feeds are used to excite all those array elements separately and hence this proposed antenna can also be assumed as a Multiple-Input-Multiple-Output (MIMO) antenna [14]. The numbering of patches and ports in this antenna is very much essential and hence, those have been shown in Fig. 8 and in Fig. 9 respectively. In this design, total five Shorting Pins/Inductive Posts have been used and those are put inside the substrate under all the five patches separately. The patch 1 is also having one square ring slot. Since, patch 1 is responsible for 1.03 GHz while patch 2, 3, 4 and 5 are responsible for 1.09 GHz, the purpose of the slot and the shorting pin is to tune the frequency 1.03 GHz and 1.09 GHz respectively. The tuning concept has been already described in the previous section. 4 Triangular shaped Metallic Caps are placed above those 4 side patches – patch 2, 3, 4 and 5 with the orientation of 0°, 90°, 180° and 270°. As this proposed structure has total eight ports, the antenna radiation beam can be continuously steered in the horizontal plane with 45° step and by exciting proper port combinations. Table 3 shows the details of it. To make an antenna array, the previously designed dual feed microstrip antenna can be used as the unit element in the array.
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D. De, P.K. Sahu / Int. J. Electron. Commun. (AEÜ) 71 (2017) 53–71 Table 4 Dimensions of the designed antenna array. Sl. No.
Section
Dimensions (In millimeters)
1 2 3 4 5 6 7 8
Substrate (length width) patch 1 (length width) patch 2, 3, 4, 5 (length width) Square ring slot (length width thickness) Diameter of the shorting pin under patch 1 Diameter of the shorting pin under patch 2–5 Height of the metallic cap Gap between patch 1 & 4 side patches
224 224 74.1 74.1 88.1 63.95 18 18 2 7 3 65 2
and testing purpose, Agilent E5071C Vector Network Analyzer (VNA) is used. The antenna radiation pattern measurement is performed in a Tapered Anechoic Chamber. 3.1. Dual feed microstrip dual patch slotted antenna Fig. 9. Port numbering in the antenna.
Table 3 Antenna radiation beam orientation with its corresponding port excitations. Ports excitation
Antenna radiation beam orientation
1 and 5 1, 5, 2 and 2 and 6 2, 6, 3 and 3 and 7 3, 7, 4 and 4 and 8 4, 8, 1 and
0° 45° 90° 135° 180° 225° 270° 315°
6 7 8 5
But in that case, the array will be quite large in size and the overall structure will not be compact. So, to make it compact and precise, this antenna array structure is proposed. Now, if the whole antenna is assumed to be a cavity, then the cavity size should be remain same in all aspects so that it can resonate at its proper frequencies. But, in this proposed case, the antenna size is reduced which causes the size reduction of the cavity in terms of length and width. So, to make the proper cavity size, the depth of the structure can be increased. Hence, an additional 4 mm. layer of FR4 substrate is added above the main substrate and under the patch 1. Hence, the total height of the substrate area under patch 1 becomes 8 mm, while the substrate height in the rest of the antenna part remains 4 mm as shown in Fig. 10. The basic Length (L) and Width (W) of all the patches have been calculated using the formulae available in the literature [15]. After that, some of the dimensions are modified little bit as per requirement during designing of the antenna. The dimensions of this proposed antenna are provided in Table 4.
3. Results and discussions These two proposed antennas are simulated, fabricated and measured in order to validate the design technology. The antennas are simulated using CST Studio Suite tool. For the measurement
The simulated and measured S Parameter of this proposed antenna are shown in Fig. 11. In this case, the antenna is designed in such a way that, Port 1 is responsible for 1.03 GHz and Port 2 is responsible for 1.09 GHz. From the simulated results, it is observed that, at 1.04 GHz S11 is 28.8 dB and S22 is 15 dB at 1.095 GHz. Similarly from the measured S parameter plots, it is shown that, S11 is 26.4 dB and S22 is 14.8 dB at 1.045 GHz and at 1.1 GHz respectively. A reasonable matching between the simulated and measured S parameters can be seen. Fig. 12 shows the simulated and measured VSWR of this designed antenna. In simulation, VSWR is 1.08 and 1.5 at 1.04 GHz and at 1.095 GHz respectively. At 1.045 GHz, measured VSWR is 1.2 and at 1.1 GHz, it is 1.55. The simulated and measured antenna impedance characteristics have been shown in Fig. 13. From the simulated result, it can be seen that, impedance is 51 O and 47 O at 1.04 GHz and at 1.095 GHz respectively. At 1.045 GHz, the measured impedance is 54 O and at 1.1 GHz, it is 46 O. Figs. 14 and 15 show the Electric Field distribution in the patches at 1.03 GHz and at 1.09 GHz respectively. From these results, it is understood that, the electric fields are in same phase for both the patches at 1.03 GHz and at 1.09 GHz. Hence, both the patches are having almost equal E-Field distribution at those two resonating frequencies. The electric path length between electric-field-zero lines corresponding to the blue colored ones in the two patches as seen from the Figs. 14 and 15, have been calculated to be 0.26 k at 1.03 GHz and 0.27 k at 1.09 GHz (approx.). Since, the path length is almost same for these two frequencies, it explains that both the patches together will radiate in such a way that the power patterns at two said frequencies will be nearly same. From the Insertion-loss plot as shown in Fig. 16, it has been observed that, S12 is around 5 dB at both the frequencies. As S12 is close to 0 dB, it is understood that the two ports are well coupled, as it is expected as the orientation of the two patches are 180° to each other with regard to the feed locations. The phase values of S12 phase plot in Fig. 17, are 17° at 1.03 GHz and 60° at 1.09 GHz. These phase values imply that the port offers two reactances at two different frequencies. The feed point is located at 0.05 k for 1.03 GHz and at 0.83 k for 1.09 GHz (approx.)
Fig. 10. Side view of additional 4 mm FR4 substrate layer under patch 1.
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Fig. 11. Simulated and measured S parameters of the Dual Feed Microstrip Antenna.
Fig. 12. Simulated and measured VSWR of the Dual Feed Microstrip Antenna.
Fig. 13. Simulated and measured impedance of the Dual Feed Microstrip Antenna.
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Fig. 14. E-Field distribution of the Dual Feed Microstrip Antenna at 1.03 GHz.
Fig. 15. E-Field distribution of the Dual Feed Microstrip Antenna at 1.09 GHz.
with respect to the electric-field-zero line at patch 1, whereas, the feed point is physically located at 0.08 k for the design frequency. So, the patch resonator works in such a way that, there are phase lead and phase lag at 1.03 GHz and at 1.09 GHz respectively. For S21, the discussion will remain same. The simulated 3D Radiation Pattern of the proposed antenna at both frequencies are shown in Fig. 18 and in Fig. 19. It is observed
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that, at 1.03 GHz and at 1.09 GHz, the antenna is radiating in the end-fire direction that is along the azimuth plane. The simulated cum measured 2D Polar Plots at two resonating frequencies 1.03 GHz and 1.09 GHz of the proposed antenna have been presented in Fig. 20. Fig. 21 shows the simulated and measured Cartesian Power Pattern Plots for both the frequencies. As from the simulation results, it is observed that, this proposed antenna provides the end-fire radiation, hence, during the antenna pattern measurement for 1.03 GHz and for 1.09 GHz, the antenna is mounted horizontally that is along the azimuth plane. From these plots, it can be seen that, the measured radiation patterns are in quite proper match with the simulated results. At both the frequencies, the peak of the radiation beam is at around 0° in the azimuth plane. The measured radiation pattern of this proposed antenna has little bit of wide beamwidth and high side lobe level as compared to that of simulated ones. The simulated antenna beamwidth is around 79° while the measured beamwidth is around 84°. Since, this proposed antenna provides narrow beamwidth in comparison to the existing TCAS antenna, the high Directivity is also achieved. Fig. 22 shows the simulated and measured peak gain of the proposed Dual Feed Microstrip Dual Patch Slotted antenna. From these results, it is shown that, at 1.03 GHz the simulated and measured maximum gain is 6.04 dB and 6.08 dB respectively. Similarly, for 1.09 GHz, the simulated and measured maximum gain is 6.34 dB and 6.1 dB respectively. The simulated radiation efficiency of the antenna is shown in Fig. 23. It is understood that, the radiation efficiency is 57% and 52% for 1.03 GHz and for 1.09 GHz respectively. The simulated temperature distribution in the antenna substrate for both the frequencies has been presented in Fig. 24. During the simulation, 10 W is given to the both antenna port 1 and 2 as input power. It is observed that, the maximum raised temperature in that circumstances is 2.7 °C for 1.03 GHz and for 1.09 GHz. In general, the temperature index of FR4 Substrate is 140 °C. So, in this case, the maximum temperature is well under the threshold level. A CAD Model of Boeing-787 Aircraft has been designed here. The proposed antenna is mounted at the top of the aircraft and then it has been simulated with the aircraft model using CST Studio Suite tool and the results are shown in Fig. 25 and in Fig. 26. The purpose of this study is to observe the antenna installed performance on a large electrical structure. To simulate this large aircraft structure, the Asymptotic Solver is used instead of FIT (Finite Integration Technique) Solver. Because, this numerical technique can
Fig. 16. Simulated and measured insertion loss of the Dual Feed Microstrip Antenna.
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Fig. 17. Phase of simulated and measured insertion loss in the Dual Feed Microstrip Antenna.
Fig. 18. 3D radiation pattern of the Dual Feed Microstrip Antenna at 1.03 GHz.
Fig. 19. 3D radiation pattern of the Dual Feed Microstrip Antenna at 1.09 GHz.
simulate large problems with the available hardware resources by using acceleration techniques like multi core/CPU computing or distributed computing. In addition, the separation of complex problems into multiple parts, with the link between them being defined by equivalent near or far field sources, can dramatically speed up installed antenna performance analysis. From the results, it is observed that, the beam pattern has not been distorted due to this large electrical structure. 3.2. Microstrip antenna array structure The simulated and measured S Parameter of this proposed antenna are shown in Fig. 27. In this case, the antenna is designed in such a way that, Port 1, 2, 3, 4 are responsible for 1.03 GHz and Port 5, 6, 7, 8 are responsible for 1.09 GHz. From the simulated results, it is observed that, at 1.03 GHz S11, S22, S33, S44 are 48.1 dB and S55, S66, S77, S88 are 25.8 dB at 1.09 GHz. Similarly from the measured S parameter plots, it is shown that, S11, S22, S33, S44 are 28.3 dB and S55, S66, S77, S88 are 33.4 dB at
Fig. 20. Simulated and measured polar plots at both the resonating frequencies of the Dual Feed Microstrip Antenna.
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Fig. 21. Simulated and measured Cartesian power pattern plots at both the resonating frequencies of the Dual Feed Microstrip Antenna.
Fig. 22. Antenna gain vs. frequency plots of the Dual Feed Microstrip Antenna.
Fig. 23. Antenna radiation efficiency plot of the Dual Feed Microstrip Antenna.
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Fig. 24. Temperature distribution in the substrate of the Dual Feed Microstrip Antenna at 1.03 GHz and at 1.09 GHz.
1.035 GHz and at 1.1 GHz respectively. A reasonable matching between the simulated and measured S parameters can be seen. Fig. 28 shows the simulated and measured VSWR of this designed antenna. In simulation, VSWR is 1.007 and 1.2 at 1.03 GHz and at 1.09 GHz respectively. At 1.035 GHz, measured
VSWR is 1.15 and at 1.1 GHz, it is 1.06. The simulated and measured antenna impedance characteristics have been shown in Fig. 29. From the simulated result, it can be seen that, impedance is 50.4 O and 48.5 O at 1.03 GHz and at 1.09 GHz respectively. At 1.035 GHz, the measured impedance is 52.6 O and at 1.1 GHz, it is 49.3 O. Figs. 30 and 31 show the Electric Field distribution in the patches at 1.03 GHz and at 1.09 GHz respectively. From these results, it can be seen that, how the electric fields are shifting from patches to patches as the port excitations are changed simultaneously. It is also understood that, E-Field distribution in the antenna is same for both the frequencies since there is no issue of phase differences between the patches. The simulated 3D Radiation Pattern of the proposed antenna at both frequencies are shown in Fig. 32 and in Fig. 33. It is observed that, at 1.03 GHz and at 1.09 GHz, the antenna is radiating in the end-fire direction that is along the azimuth plane. As, the feed excitation are changed from one combination to another, the antenna radiation beam is continuously steered from 0° to 360° in the azimuth plane. Hence, the proposed antenna can cover its whole around 360° surveillance area. The simulated 2D Polar Plots at two resonating frequencies 1.03 GHz and 1.09 GHz of the proposed antenna have been presented in Fig. 34 and in Fig. 35 respectively. Figs. 36 and 37 show
Fig. 25. Dual Feed Microstrip Antenna installed performance at 1.03 GHz.
Fig. 26. Dual Feed Microstrip Antenna installed performance at 1.09 GHz.
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Fig. 27. Simulated and measured S parameters of the proposed antenna array.
Fig. 28. Simulated and measured VSWR of the proposed antenna array.
Fig. 29. Simulated and measured impedance of the proposed antenna array.
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Fig. 30. E-Field distribution in the proposed antenna array at 1.03 GHz.
Fig. 31. E-Field distribution in the proposed antenna array at 1.09 GHz.
the simulated Cartesian Power Pattern Plots for both the frequencies. As from the simulation results, it is observed that, this proposed antenna provides the end-fire radiation, hence, during the antenna pattern measurement for 1.03 GHz and for 1.09 GHz, the antenna is mounted horizontally that is along the azimuth plane. At both the frequencies, the peak of the radiation beam is at around 0° in the azimuth plane when port 1 and 5 are simultaneously excited. After that when, port 1, 5, 2 and 6 are simultaneously excited, the radiation beam is shifted by 45° in the azimuth plane. Like this way, the beam is steered in the azimuth plane as the port excitations are changed. Hence, this proposed antenna cover the whole 360° surveillance region. The simulated antenna beamwidth is around 52° and side lobe level is 15.8 dB. As this proposed antenna provides narrow beamwidth in comparison to the dual feed antenna and the existing TCAS antenna, much high Directivity is also achieved in this antenna.
From the simulation results, it is observed that, to get the diagonal radiation pattern at both the frequencies, four antenna ports need to be excited simultaneously. But this facility was not available in the laboratory where the antenna power pattern measurement has been performed. Hence, the conventional radiation pattern measurement was not possible for this antenna and therefore, an alternative way of power pattern measurement has been executed to validate all those simulated results. First, the proposed antenna and a transmitting double-ridge horn antenna is kept in the Line of Sight (LOS) to each other. The arrangement has been done in such a way that, port 1 and 5 i.e.; patch 1 and 2 are in the LOS with the transmitting antenna. The transmitting antenna is radiating 10 dBm power. A pictorial representation of this arrangement is shown in Fig. 38. In this situation, without rotating the proposed antenna, the received power by all the ports have been measured, which is shown in Fig. 39.
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Fig. 32. 3D radiation patterns of the proposed antenna array at 1.03 GHz.
Fig. 33. 3D radiation patterns of the proposed antenna array at 1.09 GHz.
Fig. 34. Simulated polar plots of the proposed antenna array at 1.03 GHz.
Fig. 35. Simulated polar plots of the proposed antenna array at 1.09 GHz.
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Fig. 36. Simulated Cartesian power patterns of the proposed antenna array at 1.03 GHz.
Fig. 37. Simulated Cartesian power patterns of the proposed antenna array at 1.09 GHz.
Fig. 38. First arrangement during antenna power reception measurement.
These received powers are converted to the Electrical Fields using the formulas available in the literature [16] [17]. All these values have been tabulated in Table 5. From the Table, it can be seen that, the maximum electrical field is received by port 1 and 5. The total received E-Field by port 1 and 5 can be calculated by doing vector addition of E1 and E5. As, vector
E1 and E5 are in same direction, simple addition can be done. Hence, the resultant vector is given by, E1–5 = E1 + E5 = 3.81 V/m and it is maximum as compared to other received E-Fields. The resultant E-Field E1–5 is in the same direction as that of E1 and E5. The vector representation of these electrical fields is shown in Fig. 40.
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Fig. 39. Measured received power by all the ports when the proposed antenna is at 0° with the transmitting antenna.
Table 5 Measured received power and electric fields by the antenna ports for the first arrangement. Port number
Received power (in dBm)
Electric field (in V/m)
1 5 2 6 3 7 4 8
P1 = P5 = P2 = P6 = P3 = P7 = P4 = P8 =
E1 = 1.898 E5 = 1.912 E2 = 0.182 E6 = 0.192 E3 = 0.010 E7 = 0.014 E4 = 0.188 E8 = 0.193
5.44 5.38 22.96 22.65 39.88 39.17 22.78 22.65
The corresponding power of the electric field E1–5 is denoted by P1–5 and that can be calculated by using the same formulas as used before. The calculated value of P1–5 is 0.4 dBm. Hence, the total received power by the antenna port 1 and 5 is 0.4 dBm. The same measurement procedure has been repeated by keeping port 2 & 6, 3 & 7 and 4 & 8 in the LOS with the transmitting antenna. For all these cases, the total received power is around 0.4 dBm. After this measurement process, the proposed antenna is then rotated 45° clockwise with respect to LOS. In this case also, the transmitting antenna is radiating 10 dBm power. A pictorial representation of this arrangement is shown in Fig. 41. In this situation, without rotating the antenna, the received power by all the ports have been measured which is shown in Fig. 42.
Fig. 40. E-Field vector addition and its representation for the first arrangement.
As similar with the first case, these received powers are converted to the Electrical Fields using the formulas available in the literature [16] [17]. All these values have been tabulated in Table 6. From the Table, it can be seen that, the maximum electrical field is received by port 1, 5, 2 and 6. The total received E-Field by these ports can be calculated by doing vector addition of E1–5 and E2–6. E1–5 is the total electric field, received by port 1 and 5 and it is given by, E1–5 = E1 + E5 = 1.893 V/m. Similarly, we will get E2–6
Fig. 41. Second arrangement during antenna power reception measurement.
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Fig. 42. Measured received power by all the ports when the proposed antenna is at 45° with the transmitting antenna.
Table 6 Measured received power and electric fields by the antenna ports for the second arrangement. Port number
Received power (in dBm)
Electric field (in V/m)
1 5 2 6 3 7 4 8
P1 = P5 = P2 = P6 = P3 = P7 = P4 = P8 =
E1 = 0.937 E5 = 0.956 E2 = 0.999 E6 = 0.989 E3 = 0.086 E7 = 0.098 E4 = 0.092 E8 = 0.099
14.57 14.31 14.92 14.70 29.96 29.12 29.56 29.04
Fig. 43. E-Field vector addition and its representation for the second arrangement.
which is E2–6 = E2 + E6 = 1.989 V/m. The resultant E-Field vector of these two E-Fields E1–5 and E2–6 is denoted by E1–5, 2–6 and that can be calculated by applying the vector addition rule of parallelogram. The vector representation of these electrical fields is shown in Fig. 43. E1–5, 2–6 can be written as, E1–5, 2–6 = ((E1–5)2 + (E2–6)2)1/2 = 2.75 V/m and it is maximum as compared to other received E-Fields. According this vector addition rule, the direction of the resultant E-Field vector E1–5, 2–6 is in the diagonal direction with respect to the antenna structure as shown in Fig. 43. The corresponding power of the E-Field E1–5, 2–6 is denoted by P1–5, 2–6 and that can be calculated by using the same formulas as used before. The calculated value of P1–5, 2–6 is 3.23 dBm. Hence, the total received power by the antenna ports 1, 5, 2 and 6 is 3.23 dBm. P2–6, 3–7, P3–7, 4–8 and P4–8, 1–5 have been also measured by repeating the same measurement procedure with proper antenna orientation and for all these case the total received power is around 3.23 dBm in the diagonal direction. The value of the resultant power vectors P1–5, 2–6, P2–6, 3–7, P3–7, 4–8 and P4–8, 1–5 should be close to 0.4 dBm. But, as their value is 3.23 dBm, which means, 2.83 dBm power loss will be produced when, there will be simultaneously 4 ports excitation in the antenna. As a whole, by performing this power pattern testing of the proposed antenna, it can be understood that, the antenna radiates in the diagonal direction along the azimuth plane when four proper antenna ports are excited simultaneously and it somehow validates the simulation results.
Fig. 44. Simulated antenna gain vs. frequency plots of the proposed antenna array.
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Fig. 45. Antenna gain when 2 ports are excited.
Fig. 48. Temperature distribution with 4 ports excitation in the proposed antenna array.
Fig. 46. Antenna radiation efficiency plots of the proposed antenna array.
for 1.09 GHz respectively. When, four antenna ports are simultaneously excited, the radiation efficiency is 69% and 67% for 1.03 GHz and for 1.09 GHz respectively. When two ports simultaneously excited, the simulated temperature distribution in the antenna substrate for both the frequencies has been presented in Fig. 47. During the simulation, 10 W is given to the both antenna port 1 and 5 as input power. It is observed that, the maximum raised temperature in that circumstances is 3.7 °C for 1.03 GHz and for 1.09 GHz. Fig. 48 shows the simulated temperature distribution at both the frequencies, when four ports are simultaneously excited. In this case, the maximum raised temperature is 4.7 °C for 1.03 GHz and for 1.09 GHz. For this antenna also, general, the maximum temperature is well under the temperature index of FR4 Substrate. The similar CAD Model of Boeing-787 Aircraft has been also used here. The proposed antenna is placed at the top of the aircraft and then it is simulated along the aircraft model using CST Studio Suite tool and the results are shown in Fig. 49 and in Fig. 50 with proper port excitation combination. From the results, it is observed that, in this case also, the beam pattern has not been distorted due to this large electrical aircraft body. 3.3. Performance comparison between the proposed and existing TCAS antenna
Fig. 47. Temperature distribution with 2 ports excitation in the proposed antenna array.
Fig. 44 shows the simulated peak gain of the proposed antenna. Fig. 45 shows the simulated and measured peak gain of the antenna when two ports are excited. From these results, it is shown that, at 1.03 GHz the simulated and measured maximum gain is 7.46 dB and 7.3 dB respectively. Similarly, for 1.09 GHz, the simulated and measured maximum gain is 6.93 dB and 6.8 dB respectively. The simulated radiation efficiency of the antenna is shown in Fig. 46. It is understood that, when two ports are simultaneously excited, the radiation efficiency is 65% and 59% for 1.03 GHz and
Table 7 lists the comparison of the proposed antennas to the existing TCAS/ACAS antenna. From this table, it is found that, the proposed antenna provides better performance compared to the existing monopole TCAS antenna and hence, this design technique can be applied with some further development in the microstrip antenna to make it a good candidate for this application. Comparing the S parameters as seen from the Table 7, it is clear that the frequency sensitivity is better for the proposed antenna. The VSWR and impedance values of the proposed antenna are well matched like that of existing antenna. It is seen that the Gain of the proposed antenna is roughly doubled which implies the power expenditure during signal transmission is economized. From Table 7, it is observed that the Beamwidth of the proposed antenna is roughly reduced to half as compared to that of the existing TCAS antenna. Due to this narrow Beamwidth, resolving the location of the intruder aircraft in terms of bearing angle is much better and, hence, the resolution power of this proposed antenna is better than the existing one. The current TCAS antenna which is fitted in the aircraft has the Side Lobe Level of 8 dB, as shown in Table 7, while the Side Lobe
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Fig. 49. Antenna Array installed performance at 1.03 GHz.
Fig. 50. Antenna Array installed performance at 1.03 GHz.
Table 7 Performance comparison between proposed antennas and conventional TCAS antenna. Parameters
Dual feed antenna
Antenna array With 2 ports
Tx. Freq. & Rx. Freq. (In GHz) S Parameter at Tx. Freq. & Rx. Freq. (In dB) VSWR at Tx. Freq. & Rx. Freq. Impedance at Tx. Freq. & Rx. Freq. (In Ohm) Gain at Tx. Freq. & Rx. Freq. (In dB) Beamwidth at Tx. Freq. & Rx. Freq. Side Lobe Level at Tx. Freq. & Rx. Freq. (In dB) Radiation Pattern Polarization Efficiency at Tx. Freq. & Rx. Freq. Max. Temp. for 10 Watt I/P Power
Simulated Measured Simulated Measured Simulated Measured Simulated Measured Simulated Measured Simulated Measured Simulated Measured
1.04 & 1.095 1.045 & 1.1 28.8 & 15 26.4 & 14.8 1.08 & 1.5 1.2 & 1.55 51 & 47 54 & 46 6.04 & 6.34 6.08 & 6.1 79° & 77° 84° & 81° 14.3 & 13.9 10.4 & 9.5 Directional Linear 54% & 40% 1.67 °C
1.03 & 1.09 1.035 & 1.1 48.1 & 25.8 28.3 & 33.4 1.007 & 1.2 1.15 & 1.06 50.4 & 48.5 52.6 & 49.3 7.5 & 6.9 7.3 & 6.8 52° & 53° – 15.2 & 15.0 – Directional Linear 65% & 59% 3.7 °C
Existing TCAS antenna With 4 ports 1.03 & 1.09 15 1–1.5 50 7.1 & 6.7 – 53° & 54° – 14.9 & 14.2 –
69% 67% 4.7 °C
3.6 >100° 8 Directional Linear – –
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Level of the proposed antenna is around 15 dB which is almost half of the said magnitude of the existing antenna. Hence, the sensitivity of the proposed antenna is expected to be better. Due to this low level of side lobe in the proposed antenna and if this low level is considered to be the threshold value of return signal, the covering distance will likely to become more compared to the existing antenna. As the covering distance is larger, more number of aircraft which were hither to unseen in the existing system, can be detected. So, larger number of aircraft can be tracked for collision probability with respect to the own aircraft. 4. Conclusions This paper presents a designing technique for Microstrip Antenna which can be used as an aircraft tracking antenna for TCAS/ACAS application. It is shown that, the proposed microstrip antenna array provides a better performance like high gain, good directivity and narrow beamwidth as compared to that of existing monopole TCAS/ACAS antenna. The whole antenna array structure can cover the 360° surveillance region around the aircraft with proper excitation of the corresponding ports. Hence, TCAS system can measure the bearing angle of an intruder aircraft while tracking. This proposed antenna is also very much light in weight and cost effective. It has been designed using the dielectric most suited for the intended frequency of operation. The length and width of the patch has been calculated comply with the space bracket available on the fuselage of the aircraft. The above designed antenna exhibits the parameters similar to the original antenna. The proposed antenna will be fitted on the fuselage at the top of the aircraft. This fitting is aided by 8 round headed bolts so that the antenna should not fall of. The antenna has been also simulated with those 8 bolts and it has been observed that there is no additional effect in the power pattern, S parameter, VSWR and Impedance plots. The proposed antenna will not be left open at the top of the aircraft as shown in the paper. It will be encapsulated inside an aerodynamically shaped enclosure, known as ‘Radome’. For a high speed carrier a radome or an antenna of lower profile must be used to reduce the air drag. The existing TCAS Antenna is also enclosed inside a radome. Yet, the enclosure has some air-drag, however small it may be. Originally the TCAS antenna elements were having large height, however by loading the elements, the height of the elements and consequently the height of the encapsulation above the aircraft surface have been reduced so as to improve the airdrag performance [2]. The current research focus is on the designing of the antenna element. The 65 mm. height of the metallic cap will cause the height of its encapsulation to be at least of the same amount. Presently there are various antennas, i.e. Automatic Direction Finder (ADF) antenna [18], GPS antenna etc. which are fitted at the top of the aircraft like TCAS antenna. All these are encapsulated with radomes having antenna elements within each of these. Properly designed shaped-encapsulation that hides the metallic cap of 65 mm. will optimize the drag performance. Apart from the design
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considerations it also saves on the maintenance front as it is not directly exposed in the line of the airflow. Hence, this Antenna can be embedded in the body of the aircraft, so that it will offer minimum resistance to the airflow. Research on the designing of such shaped encapsulation is still going on. Therefore, with these proposed implementations in the system, this TCAS antenna would be expected to meet the requirements of the advanced avionics standards in terms of design simplicity, lightweight and high performance. Acknowledgements The authors would like to thank Space Application Centre (SAC), ISRO for proving the chance to explore their excellent measurement facilities during this research work. The authors are also very much glad to the college authority of National Institute of Technology, Rourkela, India for various financial support during this research work. References [1] Steve Henely, Rockwell Collins, The avionics hand book, 18-TCAS II, CRC Press LLC, 2001. [2] Harman WH. TCAS: a system for preventing midair collisions. Lincoln Lab J 1989;2:437–58. [3] U.S. Department of Transportation (Federal Aviation Administration), ‘‘Introduction to TCAS II”, Version 7.1, February 28, 2011 [4] Welch JD, Orlando VA. Traffic alert and collision avoidance system (TCAS): a functional overview of minimum TCAS II. USA: Lincoln Laboratory, M.I.T.; April 8, 1983. Project Report ATC-119. [5] Kuchar James K, Drumm Ann C. The traffic alert and collision avoidance system. Lincoln Lab J 2007;16. [6] Sampath KS, Rojas RG, Burnside WD. Analysis & simulation of collision voidance TCAS antennas mounted on aircraft. In: IEEE Conference, London; 1991, vol. 2, p. 948–51. [7] Harman WH, Wood ML. Triangle TCAS antenna. USA: Lincoln Laboratory, M.I. T.; 2011. Project Report ATC-380. [8] TCAS S72-1735-25, AIRNC 735. Sensor Systems Inc., Aircraft antennas since 1961. [9] Umar Khan Qasim, Bin Ihsan Mojeeb. Higher order mode excitation for high gain microstrip patch antenna. AEU Int J Electron Commun November 2014;68 (11):1073–7. [10] Shanmuganantham T, Raghavan S. Design of a compact broadband microstrip patch antenna with probe feeding for wireless applications. AEU Int J Electron Commun 2009;63(8):653–9. [11] Soon-Soo Oh, Park Wook-Ki, Jung Young-Bae, Choi Tael-II, Lee Young-Hwan. Frequency-tunable open-ring microstrip antenna with optimally-positioned varactors for radiated-power in situ measurements. AEU Int J Electron Commun September 2014;68(9):841–5. [12] Orban D, Moernaut GJK. The basics of patch antennas, orban microwave products, September 2005. [13] Wang Chien-Jen, Li Simon C, Sun Te-Liang, Lin Chun-Min. A wideband steppedimpedance open-slot antenna with end-fire directional radiation characteristics. AEU Int J Electron Commun 2013;67(3):175–81. [14] Kumar Ghosh Chandan. A compact 4-channel microstrip MIMO antenna with reduced mutual coupling. AEU Int J Electron Commun 2016;70(7):873–9. [15] Balanies CA. Antenna theory: analysis & design. 2nd ed. John Wiley & Sons Inc; 1997. [16] Fontolliet PG. Telecommunications systems, Section 3.9. PPUR Presses Polytechnics; 1996. [17] Godlewski Philippe, Tabbane Sami, Lagrange Xavier. Reseaux GSM-DCS. 4th ed. Hermes Science Publications; 1999. [18] Honeywell. Installation manual Bendix/King KR 87 automatic direction finder, May 6, 2006. Manual Number 006-00184-0006.