A novel speed sensor-less vector control of Dual Stator Induction machine with space vector based advanced 9-zone hybrid PWM for grid connected wind energy generation system

Electric Power Systems Research 163 (2018) 174–195

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Electric Power Systems Research journal homepage: www.elsevier.com/locate/epsr

A novel speed sensor-less vector control of Dual Stator Induction machine with space vector based advanced 9-zone hybrid PWM for grid connected wind energy generation system Shantanu Chatterjee a,∗ , Saibal Chatterjee b a b

Department of Electrical Engineering, National Institute of Technology, Arunachal Pradesh 791110, India Department of Electrical Engineering, North Eastern Regional Institute of Science and Technology, Arunachal Pradesh, India

a r t i c l e

i n f o

Article history: Received 2 June 2017 Received in revised form 12 September 2017 Accepted 28 February 2018 Keywords: Dual Stator Induction machine Vector control Speed sensor-less vector control Advanced bus clamping PWM Field orientation dSPACEDS-1104 DAC card

a b s t r a c t In this paper a novel speed sensor less vector control of Dual Stator Induction machine has been proposed for grid connected wind energy generation system. Dual Stator Induction machine consists of two stator winding separated by an electrical angle of 30◦ , with dissimilar pole to overcome the effect of harmonic current occurred in the stator windings. In this work the pole ratio of the two stator winding are taken as 3:1 in order to mitigate the magnetic saturation of the core and high losses which occurs at stator winding. The orientation of the machine is made in such a way that by controlling two stator currents in a perfectly decoupled way zero speed operation can be made possible. There by electrical frequency of it gets maintained at low speed, this characteristic of stator makes the Dual Stator Induction machine more suitable for speed sensor less operation. The two stator windings of the machine are fed from two independent Variable Voltage Variable Frequency inverter, which is being controlled by advanced bus clamping Pulse Width Modulation Technique. The Pulse Width Modulation proposed in this paper is a 9 zone hybrid PWM which results into lower line current ripple, reduced torque pulsation and minimized switch loss then the conventional space vector pulse width modulation, basic and advanced bus clamping PWM and the existing 3, 5 and 7 zone hybrid pulse width modulation. The simulation is being performed in MATLAB and the simulated results are being verified by experimental results. dSPACE DS-1104 Data Acquisition Card is being used to implement the overall control topology. © 2018 Elsevier B.V. All rights reserved.

1. Introduction

switches must be of high power rating. Thus the cost of system increases. By the use of multi-phase induction machine drive the power gets divided in two more of inverter legs, which results in to less voltage stress across individual switch. It eliminates the case of high power rating switches for high power drive application. More over as the switches are switched in more comparatively low voltage & low current across it, so even if the switching frequency is high the switching loss will be comparatively lower. The other advantages of multiphase induction machine are:

Three phase induction AC machine drive has been used widely in many industrial as well as renewable energy (wind/hybrid) application for past few decades due to its flexibility in speed control, easy in implementing grid tied system and simplicity in designs of control loop. Nowadays researchers have focused on inventing AC machine drive which can be used in high power & high switching frequency application. And as a result multiphase induction machine came into picture. When the three phase induction machine is used for high power and high frequency application the power semiconductor switches used in power electronics interface for the system must be able to sustain high voltage and current across it, i.e.

• • • •

∗ Corresponding author. E-mail addresses: [email protected] (S. Chatterjee), [email protected] (S. Chatterjee).

Out of different types of multiphase induction machine Dual Stator Induction machine (DSIM) is one of the most promising & widely use multiphase induction machine. DSIM can mainly categorized in two type:

https://doi.org/10.1016/j.epsr.2018.02.021 0378-7796/© 2018 Elsevier B.V. All rights reserved.

Smoothing of torque pulsation. Reduce rotor current harmonics. Smoothing DC link current Increase reliability & robustness of overall system.

S. Chatterjee, S. Chatterjee / Electric Power Systems Research 163 (2018) 174–195

• Split wound • Self Cascaded Way back in 1920 split wound based DSIM was being invented. The split wound DSIM consists of two stator winding and a squirrel cage rotor winding. The two three phase stator winding which are coupled with each other must have same pole and is applied from source of same frequency. One of main drawback of split wound DSIM is the involvement of high circulating current which arise due to low input impedance when anon-sinusoidal voltage apply at the stator [1,2]. This in turn will introduce high switching loss and there by switch of higher power rating will be required. Self-cascaded DSIM which was first introduced in 1907 is suitable to be used in low speed drive application. The reliability & robustness of split wound DSIM is comparatively higher. But the cost of construction is increased. The Dual Stator Induction machine used in the paper is having two stator winding of dissimilar pole and separated by an electrical angle of 30◦ [3]. The machine has a single rotor winding of squirrel cage configuration and made of die cast material. The 30◦ phase displacement of two stator winding enable elements of 6th harmonics torque pulsation, which in turn will reduce the rotor harmonics current which reduced the losses [4,5]. The null speed operation enables the machine to maintain a minimum frequency even at very low speed operation, which is very effective for implementation of speed sensor less control of machine. It also provides increased reliability, faster dynamic response and full scale utilization of stator winding current for production of useful torque and low circulating current. Induction machine is a very highly non-linear and higher order machine with many variables. So the control of induction machine is not a simple task. In order to control an induction machine a very robust and perfectly decoupled control is necessary. Control of induction machine can mainly be subdivided into two category [5–7] namely scalar control where the ratio of voltage and frequency i.e the flux is kept constant in order to control the speed of the machine and vector control where the frequency, phase and amplitude of the stator current is controlled to control the machine. In scalar control the torque and flux can be controlled but perfect decoupling is not possible. The change in voltage and frequency in order to control the torque or flux bring change in both the control parameters as torque and flux both are dependent on voltage and frequency of the supply. Moreover scalar control gives sluggish and poor dynamic performance. In vector control [8,9] the alignment of the flux vector is oriented in such a way that the machine behaves like a separately excited fully compensated DC machine. The control of DC machine is very simple and the control of flux and torque in order to control the speed is done in a perfectly decoupled fashion. For that reason the induction machine is made to behave like a separately excited fully compensated DC machine. Vector control of induction machine can be subdivided into three categories namely Field Oriented Control (FOC) [10,11], Direct torque control (DTC) [12] and speed sensor less control (SSLC) [13]. FOC which was first proposed by Blaschke and Hasse in 1960 can further be subdivided into two categories namely direct control and indirect control. In direct FOC the rotor flux angle ( e ) is calculated directly from the terminal voltage and current whereas in indirect FOC the rotor flux angle is calculated indirectly by firstly measuring the speed of the rotor which is then added with the synchronous speed [14–16]. The integration of the summation of rotor speed and synchronous speed gives the rotor flux angle ( e ). In Direct torque control [17] which was first patented by ABB the torque and flux of the machine is controlled by applying the suitable voltage vector states in order to increase or decrease the torque or flux. Compared to FOC control the DTC control is very simple and does not require any transformation block there by reducing the computation burden [18–20].

175

Speed sensor less vector control eliminates the use of Tacho generator which is used to measure the speed of the rotor there by reduces the cost and increases robustness of the system [21–24]. The elimination of the speed sensor also optimizes the error in control caused due to the error in speed measurement. This type of control estimates the speed from motor terminal variable [25–27]. In this paper a novel open as well as closed loop speed sensor less control of DSIM is being presented for first time. The output of the voltage and the current controller is feed to the PWM signal generator which generates the gate pulses for the inverter switches so that the inverter gives the desired output as required for the system to get synchronized to the grid. Different types of PWM techniques have been used in literature for the inverter system. Sine PWM and advanced bus clamping space vector based PWM are the most prominently used techniques. Linear control and simplicity in implementation makes Sine PWM useful as compared to other technique [28]. Other than having all the advantages of the Sine PWM, advanced bus clamping space vector PWM (ABCSVPWM) provides additional DC bus utilization factor with minimal switching [29–31]. In this paper a novel 9 zone hybrid PWM is being used for switching the rotor and grid side converters. This proposed hybrid PWM results into reduced torque pulsation, minimized switching energy loss and better harmonic performance as compared to any other conventional and hybrid PWM. The 9 zone hybrid PWM uses hybrid combination of sequences 0127 (CSVPWM), 012, 721, 1012, 2721, 0121, 7212 along with special sequence 01212 & 72121 which uses the two active state of a sector twice in a sub-cycle for equal duration of time. Equal loading of the buses, minimal switching and simplicity in implementation and high DC bus utilization factor makes it much suitable for high power application. Hybrid 3, 5 & 7 zone hybrid PWM has been proposed in literature which uses all the types of sequences mentioned above except the special bus clamping sequence 01212 & 72121 [32–35]. It has been proven by simulation and also experimentally in Section 5 of this paper that the proposed 9 Zone hybrid PWM yields best results in all prospect as compared to the 3 [35,36], 5 [37,38] & 7 [39,40] zone hybrid PWM [41–43]. The work is oriented as follows: Section 2 of the paper describes the overall system description. Modeling of the Dual Stator Induction generator is being presented in Section 3 of the paper. Section 4 discuses the proposed speed sensor-less vector control of the DSIG. In Section 5 of the paper the simulation results of the proposed speed sensor-less vector control of the DSIG used in the wind energy generation system is being presented. Advanced 9 zone hybrid PWM used for the rotor and the grid side converters is being presented in Section 6 of the paper. Section 7 of the paper describes the experimental setup of the system. Finally the paper ends with the conclusion.

2. Overall system description A six phases 1.5 KW, 2 HP, 450 V, 50 Hz squirrel cage DSIM is operated as an induction generator which is being used for a grid connected wind energy conversion system. Between the wind turbine and the DSIG a gear box of 2:1 gear ratio is being used in the proposed system. As seen in Fig. 1 two converters i.e converter I and converter II for ABC and XYZ phase are being used as rotor side converter respectively. Converter III works as the grid side converter which is connected to 3 phase grid via L filter. For synchronization of the system with the grid a synchronous reference frame base phase lock loop (PLL) is used. In order to control the speed of DSIG a novel speed sensorless vector control is implemented which eliminated the use of speed encoder. The speed is estimated by terminal variable of machine. This control topology is implemented experimentally by

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Fig. 1. Overall system diagram of the wind energy generation system.

using dSPACE DAC (DS 1104). In order to reduce the stator flux ripple, minimize the torque pulsation in the machine and reduce the switching loss of the converters, a novel advanced special sequence bus clamping PWM is used. This PWM uses the two active states (1 and 2 for sector I) twice for equal duration of time. The overall system is being simulated in MATLAB and the simulated results are also validated by the experimental results. 3. Machine modeling The DSIM is a 6 dimensional system where there two stator winding and one rotor [1–5]. Two stator winding have dissimilar no. of poles separated by angle of 30◦ . In this work the ratio of the pole of two stator winding is taken as 3:1, as this is the best combination to neglect magnetic saturation stator loss and to utilize the material of the core at its best [3–10]. The stator winding are being supplied by two independent V.V.V.F inverter. As because it a six dimensional system, the modeling of machine using actual variable become a complicated one, so the machine in this paper is being modeled in d–q reference frame. Certain assumption has been taken in order to simplify the machine modeling: • • • • • •

The stator winding are sinusoidal distributed. There is a uniform air gap between stators and rotor. Magnetic saturation of core is neglected. The two stator winding are perfectly decoupled. Core loss is neglected. Inter bar current is very low.

Fig. 2. Pictorial representation of a DSIM with stator rotor orientation.

ωsl = ωe − ωr

Fig. 2 presents a pictorial representation of a DSIM with stator rotor orientation. From Fig. 2 voltage equations of two d–q axis stator can be represented by using Eqs. (1)–(4) [10–24]. Vds1 = Rs1 Ids1 + Pϕds1 − ωe ϕqs1

(1)

Vqs1 = Rs1 Iqs1 + Pϕqs1 + ωe ϕds1

(2)

Vds2 = Rs2 Ids2 + Pϕds2 − ωe ϕqs2

(3)

Vqs2 = Rs2 Iqs2 + Pϕqs2 + ωe ϕds2

(4)

And the voltage equation of rotor in d–q axis reference frame represent by using Eqs. (5) & (6) Vdr = 0 = Rr Idr + Pϕdr − ωsl ϕqr

(5)

Vqr = 0 = Rr Iqr + Pϕqr + ωsl ϕdr

(6)

where, Vds1 ,Ids1 ,ϕds1 = d-axis stator one voltage, current and flux. Vqs1 ,Ids1 ,ϕds1 = q-axis stator one voltage, current and flux. Vds2 ,Ids2 ,ϕds2 = d-axis stator two voltage, current and flux. Vqs2 ,Iqs2 ,ϕqs2 = q-axis stator two voltage, current and flux. Vdr ,Idr ,ϕdr = d-axis rotor voltage, current and flux. Vqr ,Iqr ,ϕqr = q-axis rotor voltage, current and flux. Rs1 , Rs2 , Rr = resistance of stator one, stator two and rotor. ωe , ωsl , ωr = synchronous speed, slip speed, rotor speed. As the machine taken here is a squirrel cage induction machine where rotor bars are short circuit with each other by use of end ring. So Vdr and Vqr = 0 in Eqs. (5) & (6). The expression for stator flux linkage is given in Eqs. (7)–(10). ϕds1 = Ls1 Ids1 + Lps Ids2 + Lm Idr

(7)

ϕqs1 = Ls1 Ids2 + Lps Iqs2 + Lm Iqr

(8)

ϕds2 = Ls2 Ids2 + Lps Ids1 + Lm Idr

(9)

ϕqs2 = Ls2 Iqs2 + Lps Iqs1 + Lm Iqr

(10)

In Eqs. (7)–(10) Lps = Lsm + Lm .

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Fig. 3. Equivalent circuit representation of DSIM.

Fig. 4. Rotor bar current of the DSIG under free acceleration.

The expression for rotor flux linkage is given in Eqs. (11) & (12) ϕdr = Lr Idr + Lm Ids1 + Lm Ids2

(11)

ϕqr = Lr Iqr + Lm Iqs1 + Lm Iqs2

(12)

where, Ls1 , Ls2 , Lr = self-inductance of stator one, stator two and rotor. Lsm = common mutual leakage inductance between the two set of stator winding. Lm = mutual inductance between stator and rotor. From Eqs. (1) to (12) the equivalent circuit diagram of the DSIM can be derived as shown in Fig. 3. It can be assumed here that Ids1 = Ids2 . The expression for torque is given in Eq. (13). Te =

3P e e e e e L ((I e + Iqs2 )Idr − (Ids1 + Ids2 )Iqr ) 2 2 m qs1

(13)

Substituting the value of Idr and Iqr from Eqs. (11), (12) in to Eq. (13), Eq. (14) is obtained. Te =

3 P Lm e e e e e e ((I + Iqs2 )ϕdr − (Ids1 + Ids2 )ϕqr ) 2 2 Lr qs1

(14)

where P represents no. of poles. Fig. 4 shows the rotor bar current of the DSIG under free acceleration process. From the waveform it can be seen that the current in the rotor bar consists of two frequencies which is due to the two stator windings provided by two different frequencies. As the two stator windings of the machine are supplied from two different frequencies it results into two different magnetic field rotating at two different frequencies but as the stator have dissimilar no.

Fig. 5. Primary stator current of DSIG.

of poles the interaction of the magnetic fields cannot produce pulsating torque. Figs. 5 and 6 are the waveforms of the primary and secondary stator currents. From Fig. 5 it is quite evident that the frequencies of the secondary stator current are much higher than the primary stator current. Figs. 7 and 8 give the waveform of the two electromagnetic torques. Fig. 9 presents the waveform of the rotor speed. The rotor speed keeps rising from 0 to 0.7 s and after 0.7 the rotor speed stays constant at 710 rpm.

4. Speed sensor-less vector control of DSIG The use of speed sensor has certain disadvantage as follows • Hamper robustness of system • Make the system costly • If the speed sensor goes wrong then the results shown will be erroneous and thereby the system will loose synchronization. Therefore many of industrial application use of speed sensor are not desirable.

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Fig. 6. Secondary stator current of DSIG.

Fig. 7. Electromagnetic torque T1 of 4 poles.

Fig. 8. Electromagnetic torque T2 of 12 poles.

In this paper are a Novel scheme for speed sensor less control of DSIM is proposed both for closed & open loop control. Basically the speed is of rotor estimated from the motor terminal variable.

4.1. Open loop speed sensor less The expression of d–q axis stator flux contains Idr & Iqr which are not measurable at running condition so Idr & Iqr express in terms of stator current.

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Fig. 9. Rotor speed.

1 = Z1 Ls1

(24.b)

(15)

2 = Z2 Ls2

(24.c)

(16)

And Z, Z1 & Z2 are the leakage factors of the machine which is actually the measure of the leakage inductance of the machine and can be represented with the Eqs. (24.d)–(24.f).

Extracting the value of Idr & Iqr from Eqs. (11) & (12) Idr = Iqr =

ϕdr − Lm (Ids1 + Ids2 ) Lr ϕqr − Lm (Iqs1 + Iqs2 ) Lr

Substituting Eqs. (15), (16) in Eqs. (7)–(10) respectively Eqs. (17)–(20) are obtained. ϕds1

L2 L2 Lm = Ids1 (Ls1 − m ) + Ids2 (Lps − m ) + ϕ Lr Lr Lr dr

ϕqs1

L2 L2 Lm = Iqs1 (Ls1 − m ) + Iqs2 (Lps − m ) + ϕ Lr Lr Lr qr

(18)

ϕds2 = Ids2 (Ls2 −

2 Lm L2 Lm ) + Ids1 (Lps − m ) + ϕ Lr Lr Lr dr

ϕqs2 = Iqs2 (Ls2 −

2 Lm L2 Lm ) + Iqs1 (Lps − m ) + ϕ Lr Lr Lr qr

Vqs1 = Iqs1 (Rs1 +

2 Lm ) Ls1 Lr

(24.e)

(19)

Z2 = (1 −

2 Lm ) Ls2 Lr

(24.f)

(20)

Substituting Eqs. (15), (16) in Eqs. (5), (6) respectively Eqs. (25), (26) are obtained.

d1 d Lm ) + Ids2 + Lr dt dt dt

dt dϕqr dt

=−

1 Lm ϕ + (I + Ids2 ) + ωr ϕqr r dr r ds1

(25)

=−

1 Lm ϕqr + (I + Iqs2 ) − ωsl ϕdr r r qs1

(26)

(21)

Lm ϕ ) Lr dr

(22)

d Lm dϕdr d2 ) + Ids1 + Lr dt dt dt

−ωe (Iqs2 2 + Iqs1  +

Vqs2 = Iqs2 (Rs2 +

dϕdr

d Lm dϕqr d1 ) + Iqs2 + Lr dt dt dt

+ωe (Ids1 1 + Ids2  +

Vds2 = Ids2 (Rs2 +

(24.d)

Z1 = (1 −

dϕdr

L −ωe (Iqs1 1 + Iqs2  + m ϕqr ) Lr

2 Lm ) Lps Lr

(17)

Substituting Eqs. (17)–(20) in Eqs. (1)–(4) respectively Eqs. (21)–(24) are obtained. Vds1 = Ids1 (Rs1 +

Z = (1 −

Lm ϕ ) Lr qr

(23)

Lm dϕqr d2 d ) + Iqs1 + Lr dt dt dt

+ωe (Ids2 2 + Ids2  +

Lm ϕ ) Lr dr

(24)

where  = ZLps

(24.a)

Fig. 10. Flux distribution in DQ axis.

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Fig. 11. Proposed open loop speed sensor-less vector control of DSIM.

were  r is known as the rotor time constant and can be represented as shown in equation below r =

Lr Rr

So, the rotor flux vector ϕr rotating at a speed of ωe . From Fig. 10 the value of e can be evaluated as in Eq. (27). e = tan−1 (

4.2. Closed Loop Control

ϕqr ) ϕdr

(27) In order to get correct and accurate measurement of rotor speed closed loop speed control is necessary. The main motive of vector control is that the rotor flux (ϕr ) of rotating reference frame is coine ) and 90◦ angle with q-axis rotor cide with the d-axis rotor flux (ϕdr e ) (Fig. 12). flux (ϕqr

As it’s known that, ωe =

ωe =

de dt 1 2 ϕ 1 + ( ϕqr )

×(

ϕdr

dϕqr

− ϕqr

dt

2 ϕdr

dϕ

dr

dt

)

(28)

Substituting Eq. (25), (26) in Eq. (28) is obtained Eq. (29). Lm ϕdr (Iqs1 + Iqs2 ) − ϕqr (Ids1 + Ids2 ) ( ) 2 r ϕ2 + ϕqr

(29)

dr

Substituting the value of ωe in Eq. (29), Eq. (30) is obtained. ωr =

(ϕdr

dϕqr dt

− ϕqr

dϕ

dr

dt

)−

Lm r (ϕdr (Iqs1 2 + ϕ2 ϕdr qr

e =ϕ ϕdr r e ϕqr

dr

ωe = ωr +

The block diagram of the proposed open loop speed estimator is represented in Fig. 11. The system is called open loop system as it doesn’t have any error correction mechanism, in fact if there is any error in speed measurement then the system is unable to correct it. The proposed closed loop speed estimation system can eliminate this problem by introducing a PI controller in the control loop.

+ Iqs2 ) − ϕqr (Ids1 + Ids2 )) (30)

(where ϕr is constant)

=0

(31) (32)

By substituting Eqs. (31) & (32) in Eqs. (5) & (6), Eqs. (33) & (34) e = 0and rotor resistance R never goes to can be obtained. As Rr Idr r zero so Eq. (33) e Idr =0

ωsl = −

(33) e Rr Iqr e ϕdr

(34)

From Eq. (34) it can be seen that the slip speed is depended on e & ϕe . Placing sensor into rotating part of machine is not feasible, Iqr dr e and ϕe are not easily accessible. So rotor current & rotor flux as Iqr dr is obtained in terms of stator variable using Eqs. (11) & (12).

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Table 1 Simulation parameter table. Parameters

Values

Parameters for the wind turbine: C1 –C6

Fig. 12. Space vector diagram of FOC control based on DSIM.

Eq. (35) expresses q axis rotor current in terms q axis stator currents. e Iqr =−

e + Ie ) Lm (Iqs1 qs2

Lr

(35)

e from Eq. (35) Eq. (36) is obtained by substituting the value of Iqr in Eq. (34).

ωsl = So, r

e + Ie ) Lm (Iqs1 qs2

(36)

e r ϕdr e dϕdr

dt

e e e = Lm (Ids1 + Ids2 ) − ϕdr

(37)

e from Eq. (15) in Eq. (5), Eq. (38) can Substituting the value of Idr be obtained. e ϕdr =

e + Ie ) Lm (Ids1 ds2

(38)

d (1 + r dt )

Whenever rotor flux is having angle of e with the d axis, then d–q axis components of the rotor flux is given as  r cos e ϕdr = ϕ

&

No of blades Diameter of the blades

0.5, 116, 0.4, 5, −21, 0 respectively 3 3m

Parameters of the DSIG machine Power rating Per phase voltage Per phase stator resistance (Rs1 ) Per phase stator resistance(Rs2 ) Per phase rotor resistance referred to stator(Rr ) No. of poles Per phase leakage inductance of the stator 1 (Ls1 ) Per phase leakage inductance of the stator 2 (Ls2 ) Per phase leakage inductance of rotor (Lr ) Per phase common magnetic inductance (Lm ) Moment of inertia (J) Frictional co-efficient (B) Frequency (f) Sample time (Ts ) Rated speed Maximum speed Load connection type DC bus reference voltage

2 kW 220 V 1.325  0.435  0.816  4/12 2.2 mH 2.036 mH 2 mH 69.321 mH 0.089 kg m2 0.00722 50 Hz 1e−4 s 1500 rpm 3000 rpm Delta 400 V

makes the air gap flux density waveform sinusoidal. Fig. 15 represents the waveform the DC link voltage. It can be seen from the waveform that the magnitude of the DC link voltage varies with the change in wind speed. As the wind speed changes from 20 m/s to 25 m/s the magnitude of the DC link voltage also changes from 620 V to 780 V. The regulation of the DC link voltage is performed by the use of outer voltage controller of the converter-3 control loop. As two stator windings are present with dissimilar no. of poles in a DSIG so two electromagnetic torques are also present. Figs. 16 and 17 represents the waveform of the actual and reference electro-magnetic torque for 12 and 4 poles of the stator windings respectively. From Figs. 16 and 17 it can be observed that the actual electro-magnetic torque is faithfully tracking the reference torque and the torque is getting varied according the variation of the wind speed. From Fig. 18 it can be seen that the actual and the reference rotor speed is getting tracked in a very satisfactory manner. Fig. 19 represents the waveform of the d and q axis rotor flux vector. In vector control the rotor flux is totally aliened along the d axis rotor winding so the q axis component of the rotor flux is zero. Fig. 19 also represents the same results. Fig. 20 and 21 gives the waveform of the currents in the stator windings (ABC & XYZ phases respectively).

6. Proposed 9-zone hybrid PWM

 r sin e ϕqr = ϕ

So, sin e ϕdr = cos e ϕqr The proposed closed loop speed sensor less vector control of DSIM is shown in Fig. 13 where the estimation of rotor speed is shown in Fig. 13(b). 5. Simulation results of the speed sensor-less vector control of DSIG The parameters of the wind turbine and the DSIG is being presented in Table 1 of this paper. Fig. 14 gives the waveform of the flux density in the air gap of a DSIG. It can be seen from Fig. 14 that the waveform of the air gap flux density without dead time compensation is non sinusoidal. The introduction of dead time compensation

Conventional space vector PWM uses two active state and both the null states (0 & 7) are utilized for same amount of time in a subcycle. The vector diagram of voltages applied to a voltage source inverter (VSI) is being presented in Fig. 22. The main objectives of PWM are: (a) reduced harmonics (b) reduction of torque pulsation (c) reduced switching loss at higher switching frequency. Use of CSVPWM technique leads to higher reduction in current ripple as compared to sinusoidal PWM (SPWM). Moreover the utilization of DC bus voltage also gets increased to 7.7% as compared to SPWM and third harmonic injection PWM. The expression of dwell times can be represented as given in Eqs. (39)–(41). T1 = ma

sin(600 − ˛) sin 600

Ts

(39)

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Fig. 13. Proposed closed loop speed sensor-less vector control of DSIM. (a) Finding outωe , (b) Finding out rotor speed (ωr ).

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183

Fig. 14. Normalized flux density distribution for the DSIM in air gap.

Fig. 15. DC bus voltage variation respect to time.

Fig. 16. Electromagnetic torque for 12 poles.

T2 = ma

ma =

sin ˛ sin 600

VREF VDC

Ts

(40)

(41)

where Ts = T1 + T2 + Tz the sub is cycle duration, VREF is the reference voltage vector, T1 , T2 & TZ are the dwell time of the voltage vector states V1 , V2 , V7 & V0 respectively and ma is the modulation index. In order to further reduce the current harmonic with considerable decrease in torque pulsation a multiple division of active states

based bus clamping PWM has been presented here both in Grid and Rotor side converter control. Bus clamping PWMs are basically discontinuous PWM techniques (DPWM) where only one null state is utilized for the entire Ts duration. Bus clamping PWM techniques leads to inferior performance in terms of current ripple as compared to CSVPWM at lower modulation index but at higher modulation index (ma ≥ 0.65) the bus clamping PWM results in lower current ripple along with lower torque pulsation and reduced switching loss. The vector diagram of the current ripple over a sub-cycle for CSVPWM (0127) and different types of bus clamping PWMs

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Fig. 17. Electromagnetic torque for 4 poles.

Fig. 18. Rotor speed variation respect to time.

axis component of the stator flux ripple is being represented in Eq. (54) (Fig. 24).

2 Rrms,0127 =

T0 1 (0.5X0 )2 3 2T

+

 T1 1 (0.5X0 )2 + 0.5X0 (0.5X0 + X1 ) + (0.5X0 + X1 )2 3 T

+

 T2 1 (0.5X0 + X1 )2 − (0.5X0 + X1 )0.5X0 + (−0.5X0 )2 3 T

(42)

T0 (T1 + T2 ) 1 1 + D2 + (−0.5X0 )2 3 2T 3 T

Fig. 19. d & q-axis rotor flux waveform.

2 Rrms,012 =

+ sequences (012, 721, 0121, 7212, 1012, 2721) are shown in Fig. 23(a)–(c), (e) of this paper respectively. Fig. 23 also includes the stator flux ripple vector of a special sequence bus clamping PWM where the two active states (1 & 2 for sector 1) are divided twice in a sub-cycle and the null state (0 or 7 for sector 1) is present only once as in case of bus-clamping PWM. The expression of the q & d

 T1 4  2 4 2 T0 X + X + X0 (X0 + X1 ) + (X0 + X1 )2 27 0 T 27 0 T

4 2 (T1 + T2 ) T2 4 (X0 + X1 )2 + D 27 T 27 T

2 Rrms,721 =

 T2 4 2 T0 4  2 X + X0 (X0 + X2 ) + (X0 + X2 )2 X + 27 0 T 27 0 T

4 4 2 (T1 + T2 ) T1 + (X0 + X2 )2 + D 27 T 27 T

(43)

(44)

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185

Fig. 20. Three phase stator current for 12 poles.

2 Rrms,0121 =

+

1 2 T0 X 3 0 T

2 = Rrms,01212

 T1 1 2 X + X0 (X0 + 0.5X1 ) + (X0 + 0.5X1 )2 3 0 2T

 T2 1 + (X0 + 0.5X1 )2 − (X0 + 0.5X1 )0.5X1 + (−0.5X1 )2 3 T

+

1 T1 [(X0 )2 + (X0 )(X0 + 0.5X1 ) + (X0 + 0.5X1 )2 ] 3 2Ts

+

1 T2 [(X0 + 0.5X1 )2 + (X0 + 0.5X1 )(0.5X0 ) + (0.5X0 )2 ] 3 2Ts

+

1 T1 [(0.5X0 )2 + (0.5X0 )(−0.5X2 ) + (−0.5X2 )2 ] 3 2Ts

(45)

1 1 T1 (T1 + T2 ) + (−0.5X1 )2 + (0.5D)2 3 2T 3 T

2 Rrms,72121 =

2 Rrms,7212 =

1 2 T0 X + 3 0 T

 T2 1 2 X + X0 (X0 + 0.5X2 ) + (X0 + 0.5X2 )2 3 0 2T  T1 1 + (X0 + 0.5X2 )2 − (X0 + 0.5X2 )0.5X2 + (−0.5X2 )2 3 T

(46)

+ +

1 3

1 T1 (0.5X1 )2 3 2T

(0.5X1 )2 + 0.5X1 (0.5X1 + X0 ) + (0.5X1 + X0 )

 2 T0 T

 T1 1 (0.5X1 + X0 )2 − (0.5X1 + X0 )0.5X2 + (−0.5X2 )2 3 2T

1 T2 (0.5X2 )2 3 2T

+

 T0 1 (0.5X2 )2 + 0.5X2 (0.5X2 + X0 ) + (0.5X2 + X0 )2 3 T

+

 T1 1 (0.5X2 + X0 )2 − (0.5X2 + X0 )0.5X1 + (−0.5X1 )2 3 2T

T1 T2 T0 T1 1 1 1 + (−0.5D)2 + (0.5D)2 + D2 + (−X1 )2 3 T 3 2T T 3 T

+

1 T2 [(X0 )2 + (X0 )(X0 + 0.5X2 ) + (X0 + 0.5X2 )2 ] 3 2Ts

+

1 T1 [(X0 + 0.5X2 )2 + (X0 + 0.5X2 )(0.5X0 ) + (0.5X0 )2 ] 3 2Ts

+

1 T1 [(0.5X0 )2 + (0.5X0 )(−0.5X1 ) + (−0.5X1 )2 ] 3 2Ts

(50)

X0 = −VREF T0

(51)

X1 = [Vdc Cos(˛) − VREF ]T1

(52)

X2 = [Vdc Cos(600 − ˛) − VREF ]T2

(53)

Xd = [Vdc Sin(˛)]T1

(54)

6.1. Analysis of RMS value of the stator flux ripple depending upon the sequence of the voltage vector applied

 T1 1 2 T2 1 T0 + + D (0.5D)2 + (0.5D)D + D2 T 3 2T 3 T

2 Rrms,2721 =

T0 1 1 T2 1 T1 + T2 + (−0.5X1 )2 + (0.5Xd )2 (X0 )2 3 Ts 3 2Ts 3 Ts

(47)

1 T2 T1 1 + (−0.5D)2 + (−X2 )2 3 T 3 2T +(0.5D)2

(49)

where X0 is the q axis component of the flux ripple for the voltage vector 0 or 7. X1 & X2 are the q axis component of the flux ripple for the voltage vector 1 & 2 respectively. These quantities are presented in Eqs. (51)–(54).

1 1 T2 (T1 + T2 ) + (−0.5X2 )2 + (0.5D)2 3 2T 3 T

2 Rrms,1012 =

1 1 T2 1 T1 + T2 T0 + (−0.5X2 )2 + (0.5Xd )2 (X0 )2 3 Ts 3 2Ts 3 Ts

(48)

By substituting the expression of X0 , X1 , X2 & Xd in Eqs. (42)–(50) the Rrms,Seq will be function of amplitude of VREF i.e the modulation index (ma ), angle ˛ & sub-cycle duration Ts thereby the Rrms,Seq can be expressed as Rrms,Seq = f (VREF , ˛, Ts ). In Eq. (42) it can be observed that if angle ˛ is replaced by (˛ −60◦ ) or the T1 and T2 are interchanged or the X1 is interchanged by X2 then Eq. (42) remains unchanged thereby it can be concluded that the plot of stator flux ripple w.r.t the angle ˛ will be symmetric along ˛ equal to 30◦ for CSVPWM (0127). From Eq. (42) it can also be observed that the magnitude of the stator flux ripple is maximum at ˛ = 30◦ . From the expression ofRrms,012 (Eq. (43)), Rrms,0121 (Eq. (45)) & Rrms,01212 (Eq. (47)) it can be seen that replacement of ˛ by (˛ −60◦ ) and interchanging of active dwell times (T1 and T2 ) or interchanging of X1 & X2 leads to the RMS stator flux ripple expression of 721

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Fig. 21. Three phase stator current for 4 poles.

Fig. 22. Vector diagram of the voltage states applied to a three phase voltage source inverter.

(Eq. (44)), 7212 (Eq. (46)) & 72121 (Eq. (50)) respectively. Fig. 28 presents the graph of RMS stator flux ripple waveform vs the angle ˛ for a modulation index of 0.85 (when the VREF = 0.85VDC ). From Fig. 28 it can be observed that the magnitude of the stator flux ripple is maximum for all types of applied sequences at ˛ = 30◦ out of which sequence 01212 lead to lowest flux ripple at that value of ˛. From Fig. 25 it can be observed that the special sequence bus clamping PWM (SSBCPWM) leads to higher flux ripple from 0◦ to 10◦ and 50◦ to 60◦ of angle ˛ for a higher value of modulation index. This is due to the facts that if the modulation index increases then the magnitude of the reference voltage vector (VREF ) also increases. At ˛ equal to 0◦ and 60◦ the VREF touches the active voltage vector 1 or 2 respectively and at higher value of VREF the VERR,1 or VERR,2 almost becomes zero so the flux ripple also reduces. The value of VREF can be increased up to a maximum limit of 0.866 VDC as beyond that it will reach to over modulation so a small error persists at lower value of ˛ thereby resulting into a smaller flux ripple (Fig. 25). 6.2. Proposed 9-zone reduced torque reduced ripple reduced switching loss hybrid special sequence bus clamping PWM Bus clamping PWM of different sequences results in better performance in terms of harmonic distortion, switching loss and torque pulsation as compared to CSVPWM. The improvement becomes more effective when these bus clamping PWMs are used in combination with CSVPWM at specific subsectors of each sector. Fig. 23(a)–(c) are the different types of hybrid PWM techniques which has been used in literature in order to improve the sys-

tem performance. Fig. 23(d) is the hybrid PWM which has been proposed in this paper. Bus clamping PWM with 7212 or 0121 sequence provides reduced THD as compared to CSVPWM at higher value of the modulation index (ma ≥ 0.75). The 3-zone hybrid PWM of Fig. 23(a) uses sequence 0127 (CSVPWM) in lower modulation index (ma ≤ 0.75) and when the tip of VREF falls in region 2 or 3 i.e when the modulation index exceeds 0.75 the sequence 7212 and 0121 is applied which results into better harmonic performance as compared to CSVPWM. Five zone hybrid PWM sequence as compared to three zone hybrid PWM employs two more switching sequence namely 1012 & 2721 in addition to 0127, 0121 & 7212. From Eqs. (47) & (48) it can be proven that bus clamping PWM 1012 & 2721 outperforms sequences 0127, 0121 & 7212 in terms of RMS stator flux ripple when angle ˛ is very low close to 0◦ (Near to voltage state 1) and is very high close to 60◦ (Near to voltage state 2). And that’s the reason why in five sector hybrid when sequences 2721 & 1012 are applied in sectors 2 & 3 respectively, gives superior ripple performance as compared to three sector hybrid PWM. Fig. 23(c) represents a seven zone hybrid PWM where sequence 012 & 721 are also incorporated along with the sequences as present in five zone hybrid PWM. If fsw is the switching frequency then the sub-cycle duration for sequence 012 & 721 will be Ts = 3f1 sw while for the other sequences applied the sub-cycle duration will be Ts = 2f1 . Application of conventional bus clamping sequence sw 012 & 721 at medium modulation index results in reduced current ripple as well as reduced switching loss and lesser torque pulsation. The harmonic distortion factor vs the fundamental amplitude graph in Fig. 23 shows that seven zone hybrid PWM outperforms the three and five zone hybrid PWM in terms of harmonic distortion at higher fundamental frequency. Fig. 23(d) presents the proposed nine-zone hybrid PWM technique where sequences 01212 & 72121 are employed along with 0127 (CSVPWM), 012, 721, 0121, 7212, 1012 & 2721. From Eqs. (49) & (50) it is quite evident that double division of the two active states is the best option to reduce the RMS current ripple among all the existing bus clamping PWM techniques. More over special sequence 01212 performs best in terms of switching loss when energy saved by not switching the C phase is greater than the energy lost by switching the B phase twice in a sub-cycle. Likewise the sequence 72121 gives lowest switching loss when the energy saved by not switching the A phase is grater then the energy lost by switching the B phase twice in a sub-cycle. It can also be concluded that sequence 01212 & 72121 outperforms the other sequences in terms of switching loss when the load current through the B phase is lower than the other two phases (Figs. 26 and 28).

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Fig. 23. (a) Error vector diagram of sector 1. q & d axis stator flux ripple vector of (b) CSVPWM (0127) (c) CBCPWM-I (012) (d) CBCPWM-II (721) (e) ABCPWM-I (0121) (f) ABCPWM-II (7212) (g) ABCPWM-III (1012) (h) ABCPWM-IV (2721) (i) SSBCPWM-I (01212) (j) SSBCPWM-II (7212).

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Fig. 24. Special sequence PWMs (a) continual clamped (b) split clamped.

CSVPWM (0127) 3-Zone Hybrid PWM 5- Zone Hybrid PWM 7-Zone Hybrid PWM 9-Zone Hybrid PWM

1.4 1.3 1.2 1.1 1.0 0.9 0.8 0.7 0.6 -80

-60

-40

-20

0

20

40

60

80

Fig. 27. Variation of switching power loss w.r.t power factor angle for CSVPWM & different types of bus clamping PWMs. Fig. 25. Variation of RMS value of the stator flux vs angle (˛) at an VREF of 0.85.

The waveform of the simulated and experimental values of torque ripple is represented in Fig. 29. Table 2 is being derive from Fig. 29 where different values of torque ripple at different values of Vref (0.6, 0.8, 0.85) is being presented. From Table 2 it can be seen that that the 9 proposed hybrid 9 zone PWM performs superior in terms of torque ripple as compare to CSV PWM, and & zone hybrid PWM. For Vref of 0.8, experimentally the 9 zone PWM leads to 25.02% reduction in torque ripple as compared to CSV PWM, while in simulation the reduction in torque ripple is 26.68%. This indicates

that there is 6.2% mismatch between experimental and simulated results. This dissimilarity might be due to several idealistic assumptions taken while modeling the machine in simulation. Also the effect of dead-time and different losses which are neglected while simulating the machine contributes for this dissimilar results. But from Table 2 it can be concluded that the simulated and experimental values of torque ripple at different modulation index are very close to each other.

Fig. 26. (a) 3-zone hybrid PWM (b) 5-zone hybrid PWM (c) 7-zone hybrid PWM (d) 9-zone hybrid PWM.

0.015

1.74 5.80 14.89 5.88 25.02 35.32

0.020

CSV PWM

0.025

% age reduction in experimental value (as compare to proposed 9-zone PWM)

CSVPWM 3-Zone Hybrid PWM 5-Zone Hybrid PWM 7-Zone Hybrid PWM Proposed 9-Zone Hybrid PWM

0.030

189

7-Zone PWM

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20

30

40

Fig. 28. Variation of harmonic distortion factor w.r.t fundamental frequency for CSVPWM and different types of hybrid bus clamping PWM techniques.

5-ZONE H PWM(S) 7-ZONE H PWM(S) Proposed 9-ZONE H PWM(S)

0.07

9-Zone PWM

CSV PWM (Ex)

0.06 0.05

Proposed 9-ZONE H PWM (Ex)

0.04

1.71 8.52 11.52

10

6.07 26.68 33.62

0

0.05947 0.02825 0.01999

0.000

CSV PWM

% age reduction in simulation value (as compare to proposed 9-zone PWM)

0.005

7-Zone PWM

0.010

0.03

0.045 0.040 0.035

Proposed 9-ZONE H PWM (Ex)

0.030 0.025 0.020 0.015 0.010 0.005 0.0

0.2

0.4

0.6

0.8

1.0

Fig. 30. Simulated (S) and experimental (Ex) values of VTHD w.r.t modulation index for CSVPWM and different types of hybrid bus clamping PWMs.

Similarly, the % age reduction in total harmonic distortion in voltage is shown in Table 3, which is taken from Fig. 30. From the experimental values, the percentage reduction in VTHD as compared

0.06057 0.02999 0.02349 0.06319 0.03768 0.03091 0.05862 0.02715 0.01996

0.050

0.05964 0.02968 0.02256

0.055

0.06241 0.03703 0.03007

3-ZONE H PWM (S) 5-ZONE H PWM (S) 7-ZONE H PWM (S) Proposed 9-ZONE H PWM (S) CSV PWM (Ex)

0.060

CSV PWM

Fig. 29. Simulated (S) and experimental (Ex) values in variation of RMS value of torque ripple w.r.t reference voltage vector (VREF ) for CSVPWM and different types of hybrid bus clamping PWMs.

9-Zone PWM

1.0

7-Zone PWM

0.8

CSV PWM

0.6

Experimental results

0.4

Vref

0.2

Table 2 Comparison of simulated and experimental values of torque ripple (N-m) at different values of Vref .

0.0

Simulation results

0.00 -0.01

0.6 0.8 0.85

0.01

7-Zone PWM

0.02

CSV

5.07 22.13 35.23 27.8 54.5 66.46 6.21 22.7 37.4 28.7 56.6 57.14

7-Zone CSV

0.02543 0.01470 0.01077 0.02679 0.01888 0.01663 0.03525 0.03236 0.03212

9-Zone PWM 7-Zone PWM CSV PWM 9-Zone PWM

0.02446 0.01377 0.00972

% age reduction in simulation value (as compare to proposed 9-zone PWM) Hardware results

Fig. 31. Dimension of the stator slot. Table 4 Design objectives of the Dual Stator Induction motor. Description

Details of parameters

Motor type Motor continuous output Power Rated line to line RMS voltage Connection type Rated frequency No. of poles Rated speed Maximum speed Efficiency at rated load PF at rated load Frame size Cooling Ingress protection Insulation

3 phase, Dual Stator, squirrel cage rotor 1.126 kW (doubt2) 110/330 Star 50 Hz 2/6 696 rpm 1300 rpm 75.4% 0.8 112M Convection air cooling IP 55 Class ‘F’

to CSVPWM is 54.5% at a modulation Index of 0.8. For that same modulation index in simulation values, the percentage reduction in VTHD as compare to CSV PWM is 56.6%.

CSV PWM

0.02608 0.01782 0.01555 0.03435 0.03175 0.02268 0.6 0.8 0.85

Simulation results

7-Zone PWM

7. Experimental setup & results

Modulation index (ma )

Table 3 Comparison of simulated and experimental values of voltage harmonics (VTHD ) at different values of modulation index (ma ).

7-Zone

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% age reduction in experimental value (as compare to proposed 9-zone PWM)

190

The overall experimental setup is consisted of three steps: (a) the power circuits which consists of the Dual Stator Induction generator, inverters, DC link capacitor, filter inductor; (b) signal conditioning circuit which consists of voltage sensor, current sensors & gate driver circuits & (c) system control unit which consists of DAC card (dSPACE DS1104). The DAC card is used here to implement the overall control logic namely the special sequence bus clamping PWM pulses for the rotor & the grid side converter, the speed sensor-less vector control for the DSIG, the SRF-PLL and the current and the voltage controller. A 1.5 kW (2 HP), 415 V, 50 Hz 3-phase induction motor made by Havell’s is used here to make the Dual Stator Induction generator. A 5 HP, 220 V, 1500 rpm DC motor is used as a prime mover to drive the DSIM as a DSIG. The dimension of the stator slot of the DSIG is given in Fig. 31.where b0 = 2.76 mm; b1 = 0.3 mm; d0 = 0.9 mm; d1 = 3 mm; d2 = 1 mm; d3 = 9 mm; d4 = 2 mm; d5 = 9 mm; d6 = 2 mm; b2 = 6.2 mm (Tables 4–6). The overall circuit diagram of the experimental setup is shown in Fig. 32. In order to implement the speed sensor-less vector control topology the voltage and current of DSIG has to be sensed. LEM

S. Chatterjee, S. Chatterjee / Electric Power Systems Research 163 (2018) 174–195 Table 5 Stator design details. Air gap flux density Turns per phase Number of coils per slot Slots per pole per phase Distribution factor Pitch factor Slot opening factor Skew factor Winding factor Slot pitch Inside diameter of stator Outside diameter of stator Stator core length Effective length of stator Phase resistance at 75 degree Depth of stator core Stator leakage inductance

0.16 T 216 (2 poles) 36 6 0.966 (60◦ phase belt) 1 0.9991 1 0.965 13 mm 127 mm 209.55 mm 57.15 mm 57.81 mm 0.5  26.9 mm 14.634 mH

0.718 T 144 (6 poles) 72 4

0.3 

Table 6 Rotor design details. Length of air gap Rotor bar length Rotor Bar diameter Rotor bar cross-section area Resistance of each bar Effective resistance of each bar Resistance of each ring Rotor resistance referred to stator Slot leakage inductance per bar Rotor end winding inductance per bar

0.389 mm 0.065 m 8.98 × 10−3 m 63.4 × 10−6 m2 (63.4 mm2 ) 14.5 ␮ 89.93 ␮ 1.8 ␮/rotor slot pitch 3.28  (2 pole)

0.68  (6 pole)

0.0457 ␮H 9.6 × 10−9 H

manufacture four current sensors LA55P and four voltage sensor LV20P are being used. This voltage and current sensor is basically Hall Effect sensor as seen from the date sheet. The turn ratio of the current sensor LA55P is 1:1000 and the maximum value of secondary current is 50 A there the primary current is to be measured.

191

The circuit diagram of current sensor is shown in Fig. 38. As the current sensor and the op-amp 741 both required ±50 V supply so IC7851 and IC7915 are being used. In order to keep the output of current sensor within permissible limit of DAC card, the op-amp 741 is used. The current sensor is properly calibrated for this purpose. LV20P voltage sensor is used for sensing the machine terminal voltage, DC link capacitor voltage and grid voltage. The voltage sensor card is also calibrated according to the specification of data sheet. The PWM pluses are obtained from the DAC card DS1104 are fed through the converter through gate driver VLA517-01R. The gate driver IC VLA are manufactured by FUJI and are used to drive the N-Channel of IGBT modules of converter. This IC requires +20 V for its operation which is provided by IC LM7820. The gate driver IC provided optical isolation which ensures safe operation of converter switches (Table 7). Fig. 33(a) represents the waveform of actual and estimated rotor speed. It can be inferred that the actual rotor speed is tracking the estimated rotor speed in a quit efficient manner. It can also be seen that the rotor speed is changing from (325) rpm to (600) rpm in a duration of (5) s. Fig. 34(a) contains 4 waveforms, the first waveform represent the power of the ABC winding, second waveform represent power of XYZ winding. Third waveform represents the current along Aphase. Fourth waveform represents the current of X-phase. From this figure it can be observed that the power along ABC winding changes from (400 W) to (320 W) at time t = (2 s). By the same time power across XYZ winding increase from (726 W) to (801 W) approximately. At the same instant of time current across A-phase false down from (4.2 A) to (2.5 A) and the current across X-phase increase from (3.36 A) to (8.2 A). This happens due to the variation of the load resistance from (100 ) to (70 ). Fig. 34(b) represents the effect of magnetic saturation on the air gap flux linkage of the core. The hole magnetizing flux lies along the axis of the reference frame, so it is expected that only d-axis magnetizing inductance of machine varies with a variation of the air gap flux linkage. While the q axis magnetizing inductance does not see any variation and remain constant at an unsaturated value. As in dual stator inductance machine ABC and XYZ winding having two dissimilar

Table 7 Parameters of the hardware components. Components

Company/IC No

Specification

IGBT switches for making of RSC & GSC Gate-driver IC Dual Stator Induction generator (DSIG)

SEMIKRON (SKM50GB12T4) FUJI (VLA517-01R) Havell’s Lafeli 3 Phase Induction Motor (A local 1.7 kW Normal Induction Machine is wound as a DSIG)

1200 V, 50 A N-channel 25 mA (maximum input current) Given in Table 3

DC motor Gear box Current sensor

AEC DC Motor Mazda LEM (LEM LA-55P)

Voltage sensor

LEM (LEM LV-20P)

Bulb load Data accusation card Capacitor DC link capacitor Filter inductance DSO Power analyzer Thermocouple Single phase bridge rectifier

NA dSPACE 1104 EPCOS EPCOS NA Agilent Technologies (DSO 1014A) Tektronix (PA 4000) T Type KBPC 3510

Three phase bridge rectifier Transformers Three phase auto-transformer Single phase variac

Hirect (HD 35/12) GURU Dimmer Tech Dimmer Tech

1.5 KW (2-HP) Efficiency—74.5% 5 HP; 1500 rpm, 220 V 2:1 Turns ratio 1:1000; max value of secondary current 50 mA; nominal primary rms current 50 A. Turns ratio 2500:1000; max value of secondary current 25 mA 500 W 250 V As per the datasheet given in the website. 1000 ␮F 4700 ␮F, 450 V 5 mH & current rating is 20 A. 100 M/Hz, 2 GSa/s As per the datasheet given in the website. NA Voltage: 50–1000 V Current: 35 A As per the data sheet given in the website NA As per the data sheet given in the website As per the data sheet given in the website

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Fig. 32. Overall experimental setup.

Fig. 33. (a) Actual and estimated rotor speed (X-axis: 0.9 s/div, Y-axis: 69 rpm/div). (b) Stator ABC phase voltage waveform.

number of poles, so mutual inductance between them must be zero. When the machine is in no load condition and the ABC and XYZ winding are supplied with voltage and frequency of 110 V/50 Hz and 330 V/150 Hz respectively. The waveform of the A phase & X phase current are presented in Fig. 34(b). From Fig. 34(b) it can be observed that when rotor speed is varying with time, the current

Fig. 34. (a) Power of ABC winding with change in load resistance (X-axis: 0.66 s/div; Y-axis: 150 W/div); Power in XYZ stator winding with change in load resistance (Yaxis: 350 W/div); Current in A Phase (Y-axis: 0.34 A/div); Current in X Phase (Y-axis: 2.6 A/div), (b)Current of A & X phase when the machine is in free acceleration.

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Fig. 35. (a) Current waveform of A-phase and X-phase displaced by an angle of 30◦ (b) Waveform of  PLL .

in A phase and X phase are (15 A) and (28 A) respectively. When the rotor speed get constant then current in two phases reduces to zero. Fig. 35(a) represents current waveform of A-phase and X-phase. It’s clear from Fig. 35(a) that phase difference between A-phase and X-phase is 30◦ . Fig. 35(b) represents the waveform of the  PLL produced by the PLL block used for synchronization of the system with the grid. From Fig. 35(b) it can be concluded that estimated and measured response shows perfect tracking. Fig. 36(a) & (b) represents the no-load line current waveform of the system by application of 7 and 9 zone hybrid PWM respectively. From the figure it can be concluded that proposed 9-zone hybrid PWM leads to better harmonic performance as the THD gets reduced from 0.2032 to 0.1246. Fig. 37(b) represents the DC link voltage waveform of the system. It can be seen from the figure DC link voltage is being maintained at (600 V). Fig. 38(a) represents the waveform of the grid voltage and output voltage of converter III. From the figure it can be seen that the two voltages are perfectly in phase with each other, this is due to the use of the synchronous reference frame PLL which enables the voltage of the converter III to be properly synchronous with grid voltage. Fig. 38(b) represent the waveform of PLL output for the constant injected current reference of (0.65 A). The PLL is used in grid side converter control loop. 8. Conclusion This paper presents a novel speed sensor-less control of DSIG used in grid connected WECS, where the converters are controlled by using 9 zone hybrid PWM. The use of multi phase AC machine

in WECS reduces the magnetic saturation of the core and losses occurred in stator winding of the machine. The proposed speed sensor-less vector control of DSIG eliminates the use of speed encoder, which reduces the overall cost of the system. The use of speed sensor in conventional vector control scheme hampers the system robustness as it may give rise to problems caused due to erroneous speed measurement by the speed encoder. Speed sensorless control does not suffer from those problems as the speed is estimated from motor terminal variables. It can be seen from the simulation as well as experimental results that the proposed speed sensor-less control provides satisfactory performance as compared with conventional vector control scheme. The converters used in grid and rotor side converters are being controlled by novel 9 zone hybrid PWM. As seen in Fig. 29 the 9 zone hybrid PWM employs CSVPWM sequence (0127) along with different combinations of exiting (012, 721), advanced (0121, 7212, 1012, 2721) and advance special sequence (01212, 72121) bus clamping PWM sequences. Sequence 01212, 72121 employs two active state of a sector twice in a sub cycle for equal duration of time and the null state is applied once in a sub cycle. This special sequence BCPWM can be divided in two types namely continual and split clamp according to the application of switching sequence as seen in Fig. 27. The split clamp special sequence BCPWM provides superior harmonics performance as compared to continual clamped special sequence PWM. Continual clamp special sequence BCPWM gives better performance in terms of switching loss when the power factor is closed to unity while the split clamp special sequence BCPWM results in lower switching loss at lower power factor. The 9 zone hybrid PWM proposed in this paper provides reduced torque pulsation, reduced line current ripple and minimal switching loss. The paper presents experimental and simulated results which prove the same.

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Fig. 36. Experimental waveform of the line current at VREF = 0.85 (a) existing 7 zone hybrid PWM (THD %:-0.2032) (b) proposed 9 zone hybrid PWM (THD %: .0.1246). (2.3 A/div).

Fig. 37. (a) Phase to voltage of the converter III. (b) DC link voltage waveform (450 V/div).

Fig. 38. (a) Grid voltage and output voltage of converter III. (b) Waveform of PLL output for the constant injected current reference of (0.65 A).

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References [1] Radu Bojoi, Mario Lazzari, Francesco Profumo, Alberto Tenconi, Digital field-oriented control for dual three-phase induction motor drives, IEEE Trans. Ind. Appl. 39 (3) (2003) 752–760. [2] J.M. Guerrero, Olorunfemi Ojo, Flux level selection in vector-controlled dual stator winding induction machines, IET Electr. Power Appl. 3 (6) (2009) 562–572. [3] Peng Han, Ming Cheng, Xinchi. Wei, Ning Li, Modeling and performance analysis of a dual-stator brushless doubly fed induction machine based on spiral vector theory, IEEE Trans. Ind. Appl. 52 (2) (2016) 1380–1389. ˜ [4] Alfredo R. Munoz, Thomas A. Lipo, Dual stator winding induction machine drive, IEEE Trans. Ind. Appl. 36 (5) (2000) 1369–1379. ´ [5] Krzysztof Pienkowski, Analysis and control of dual stator winding induction motor, Arch. Electr. Eng. 61 (3) (2012) 421–438. [6] FranC¸ois BonnetFrancois, Paul-Etienne Vidal, Maria Pietrzak-David, Dual direct torque control of doubly fed induction machine, IEEE Trans. Ind. Electron. 54 (5) (2007) 2482–2490. [7] Radu Bojoi, Alberto Tenconi, Giovanni Griva, Francesco Profumo, Vector control of dual-three-phase induction-motor drives using two current sensors, IEEE Trans. Ind. Appl. 42 (5) (2006) 1284–1292. [8] R. Bojoi, E. Levi, F. Farina, Alberto Tenconi, Francesco Profumo, Dual three-phase induction motor drive with digital current control in the stationary reference frame, IEE Proc. Electr. Power Appl. 153 (1) (2006) 129–139. [9] Krishna Keshaba Mohapatra, R.S. Kanchan, M.R. Baiju, P.N. Tekwani, K. Gopakumar, Independent field-oriented control of two split-phase induction motors from a single six-phase inverter, IEEE Trans. Ind. Electron. 52 (5) (2005) 1372–1382. [10] Yifan Zhao, Thomas A. Lipo, Modeling and control of a multi-phase induction machine with structural unbalance, IEEE Trans. Energy Convers. 11 (3) (1996) 570–577. [11] Zhiqiao Wu, Olorunfemi Ojo, Jyoti Sastry, High-performance control of a dual stator winding DC power induction generator, IEEE Trans. Ind. Appl. 43 (2) (2007) 582–592. [12] Meriem Abdellatif, Mustapha Debbou, Ilhem Slama-Belkhodja, Maria Pietrzak-David, Simple low-speed sensor-less dual DTC for double fed induction machine drive, IEEE Trans. Ind. Electron. 61 (8) (2014) 3915–3922. [13] Cristian Lascu, Ion Boldea, Frede Blaabjerg, Comparative study of adaptive and inherently sensor-less observers for variable-speed induction-motor drives, IEEE Trans. Ind. Electron. 53 (1) (2006) 57–65. [14] Kouki Matsuse, Yuusuke Kouno, Hirotoshi Kawai, Shinichi Yokomizo, A speed-sensor-less vector control method of parallel-connected dual induction motor fed by a single inverter, IEEE Trans. Ind. Appl. 38 (6) (2002) 1566–1571. [15] Junyong Lu, Weiming Ma, Xiao Zhang, XinLin Long, Tan Sai, A multi segmented long-stroke dual-stator pulsed linear induction motor for electromagnetic catapult, IEEE Trans. Plasma Sci. 44 (10) (2016) 2211–2217. [16] Saptarshi Basak, Chandan Chakraborty, Dual stator winding induction machine: problems, progress, and future scope, IEEE Trans. Ind. Electron. 62 (7) (2015) 4641–4652. [17] Djafar Hadiouche, Hubert Razik, Abderrezak Rezzoug, On the modeling and design of dual-stator windings to minimize circulating harmonic currents for VSI fed AC machines, IEEE Trans. Ind. Appl. 40 (2) (2004) 506–515. [18] Wagner Fontes Godoy, Ivan Nunes da Silva, Alessandro Goedtel, Rodrigo Henrique Cunha Palácios, Tiago Drummond Lopes, Application of intelligent tools to detect and classify broken rotor bars in three-phase induction motors fed by an inverter, IET Electr. Power Appl. 10 (5) (2016) 430–439. [19] Wagner Fontes Godoy, Ivan Nunes da Silva, Alessandro Goedtel, Rodrigo Henrique Cunha Palácios, Evaluation of stator winding faults severity in inverter-fed induction motors, Appl. Soft Comput. 32 (2015) 420–431. [20] El Achkar, Rita Mbayed Maria, Georges Salloum, Sandrine Le Ballois, Eric Monmasson, Generic study of the power capability of a cascaded doubly fed induction machine, Int. J. Electr. Power Energy Syst. 86 (2017) 61–70. [21] A.S. Abdel-Khalik, M.I. Masoud, B.W. Williams, Vector controlled multiphase induction machine: harmonic injection using optimized constant gains, Electr. Power Syst. Res. 89 (2012) 116–128. [22] Jingbo Kan, Kai Zhang, Ze Wang, Indirect vector control with simplified rotor resistance adaptation for induction machines, IET Power Electron. 8 (7) (2015) 1284–1294.

195

[23] H. Amimeur, R. Abdessemed, D. Aouzellag, E. Merabet, F. Hamoudi, Modeling and analysis of dual-stator windings self-excited induction generator, J. Electr. Eng. (JEE) 8 (3) (2008) 18–24. [24] M.F. Mimouni, R. Adhifaoui, Robust speed identification for speed sensor-less vector control of current-fed double-star induction machine, ICECS 2000. The 7th IEEE International Conference on Electronics, Circuits and Systems 2 (2000) 809–813, IEEE, 2000. [25] G.D. Marques, M.F. Iacchetti, Sensor-less frequency and voltage control in the stand-alone DFIG-DC system, IEEE Trans. Ind. Electron. 64 (3) (2017) 1949–1957, http://dx.doi.org/10.1109/TIE.2016.2624262. [26] R. Cardenas, R. Pena, S. Alepuz, G. Asher, Overview of control systems for the operation of DFIGs in wind energy applications, IEEE Trans. Ind. Electron. 60 (7) (2013) 2776–2798, http://dx.doi.org/10.1109/TIE.2013.2243372. [27] B. Mahato, R. Raushan, K.C. Jana, Modulation and control of multilevel inverter for an open-end winding induction motor with constant voltage levels andharmonics, IET Power Electron. 10 (1) (2017) 71–79, http://dx.doi. org/10.1049/iet-pel.2016.0105. [28] G. Narayanan, Harish K. Krishnamurthy, Di Zhao, Rajapandian Ayyanar, Advanced bus-clamping PWM techniques based on space vector approach, IEEE Trans. Power Electron. 21 (4) (2006) 974–984. [29] A.C. Binojkumar, J.S. Siva Prasad, G. Narayanan, Experimental investigation on the effect of advanced bus-clamping pulse-width modulation on motor acoustic noise, IEEE Trans. Ind. Electron. 60 (2) (2013) 433–439. [30] Kaushik Basu, J.S. Siva Prasad, G. Narayanan, Harish K. Krishnamurthy, Rajapandian Ayyanar, Reduction of torque ripple in induction motor drives using an advanced hybrid PWM technique, IEEE Trans. Ind. Electron. 57 (6) (2010) 2085–2091. [31] Kaushik Basu, J.S. Siva Prasad, G. Narayanan, Minimization of torque ripple in PWM AC drives, IEEE Trans. Ind. Electron. 56 (2) (2009) 553–558. [32] Di Zhao, V.S.S. Pavan Kumar Hari, Gopalaratnam Narayanan, Rajapandian Ayyanar, Space-vector-based hybrid pulse-width modulation techniques for reduced harmonic distortion and switching loss, IEEE Trans. Power Electron. 25 (3) (2010) 760–774. [33] Jayanta Biswas, Meenu Nair, Vivek Gopinath, Mukti Barai, An optimized hybrid SVPWM strategy based on multiple division of active vector time (MDAVT), IEEE Trans. Power Electron. 32 (6) (2017) 4607–4618. [34] G. Narayanan, Di Zhao, Harish K. Krishnamurthy, Rajapandian Ayyanar, V.T. Ranganathan, Space vector based hybrid PWM techniques for reduced current ripple, IEEE Trans. Ind. Electron. 55 (4) (2008) 1614–1627. [35] Di Zhao, G. Narayanan, Raja Ayyanar, Switching loss characteristics of sequences involving active state division in space vector based PWM, Applied Power Electronics Conference and Exposition, 2004. APEC’04. Nineteenth Annual IEEE 1 (2004) 479–485, IEEE, 2004. [36] H. Krishnamurthy, G. Narayanan, R. Ayyanar, V.T. Ranganathan, Design of space vector-based hybrid PWM techniques for reduced current ripple, Applied Power Electronics Conference and Exposition, 2003. APEC’03. Eighteenth Annual IEEE 1 (2003) 583–588, IEEE, 2003. [37] H. Amimeur, D. Aouzellag, R. Abdessemed, K. Ghedamsi, Sliding mode control of a dual-stator induction generator for wind energy conversion systems, Int. J. Electr. Power Energy Syst. 42 (1) (2012) 60–70. [38] Samira Chekkal, Narimen Aouzellag Lahac¸ani, Djamal Aouzellag, Kaci Ghedamsi, Fuzzy logic control strategy of wind generator based on the dual-stator induction generator, Int. J. Electr. Power Energy Syst. 59 (2014) 166–175. [39] Arunava Chatterjee, Debashis Chatterjee, An improved excitation control technique of three-phase induction machine operating as dual winding generator for micro-wind domestic application, Energy Convers. Manage. 98 (2015) 98–106. [40] G. Narayanan, V.T. Ranganathan, Triangle-comparison approach and space vector approach to pulse-width modulation in inverter fed drives, J. Indian Inst. Sci. 80 (2000) 409–427. [41] D.G. Holmes, T.A. Lipo, Pulse Width Modulation for Power Converters: Principles & Practices, IEEE Press, 2003. [42] G.S. Buja, G.B. Indri, Optimal pulse width modulation for feeding AC motor, IEEE Trans. Ind. Appl. 1A-13 (1) (1977) 38–44. [43] A. Edpuganti, A.K. Rathore, Fundamental switching frequency optimal pulse-width modulation of medium-voltage nine-level inverter, IEEE Trans. Ind. Electron. 62 (7) (2015) 4096–4104, http://dx.doi.org/10.1109/TIE.2014. 2379583.