- Email: [email protected]
Design and fabrication of CSRR based tunable mechanically and electrically efficient band pass filter for K-band application
Accepted Manuscript Short communication Design and fabrication of CSRR based tunable mechanically and electrically efficient band pass filter for K-band application Anwesha Choudhury, Santanu Maity PII: DOI: Reference:
S1434-8411(16)30820-2 http://dx.doi.org/10.1016/j.aeue.2016.11.021 AEUE 51737
To appear in:
International Journal of Electronics and Communications
Received Date: Accepted Date:
25 September 2016 22 November 2016
Please cite this article as: A. Choudhury, S. Maity, Design and fabrication of CSRR based tunable mechanically and electrically efficient band pass filter for K-band application, International Journal of Electronics and Communications (2016), doi: http://dx.doi.org/10.1016/j.aeue.2016.11.021
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Author Agreement
Paper Title: “Design and fabrication of CSRR based tunable mechanically and electrically efficient band pass filter for K-band application” We are ensuring that the paper is not submitted elsewhere. The detailed of the authors are given belowAnwesha Choudhury, M.Tech
Santanu Maity, Ph.D
Research Scholar, Department of Electronics and Assistant Professor, Department of Electronics Communication Engineering, National Institute and Of
Technology,
Arunachal
Engineering,
National
Pradesh-971112, Institute Of Technology, Arunachal Pradesh-
Email: [email protected]
Thanks & Regards. Dr. Santanu Maity (Corresponding Author)
Communication
971112, Email: [email protected]
Design and fabrication of CSRR based tunable mechanically and electrically efficient band pass filter for K-band application Anwesha Choudhury1, Santanu Maity1* Department of Electronics and Communication Engineering National Institute Of Technology, Arunachal Pradesh-971112 Email: [email protected]
Abstract: The paper represents the design and fabrication of a coplanar wave guide (CPW) based Band pass filter for K – band application. In this paper for getting the desired resonant frequency with a good performance (in terms of notch level) modification of the signal line and switches are carried out. Ten distributed switches are implemented to attain the tunability. The simulated result shows that the insertion loss and return loss is improved by using the (CSRR) structure. Detailed Electro mechanical analysis of various materials and parameters are done to make the device applicable for satellite communication. The simulated and fabricated devices operating in the frequency range of 26.16-27.37 GHz has attained a Return loss of -42.76 dB and an Insertion loss of-0.9dB. It gives a tuning of 4%. The device works with a low pull-in voltage of 15.3 V. Keywords: Micro Electro Mechanical System (MEMS), Coplanar Waveguide (CPW), Band pass filter, Complementary split ring resonator (CSRR) Introduction:
Micro Electro Mechanical System (MEMS) is a technology of miniature devices
formed by combination of micromechanical and microelectronic components on a common substrate [1] .RF MEMS technology is rapidly evolving and hence new applications are continuously being reported [7-15]. RF MEMS switches, switched capacitors, tunable inductors, resonators, filters are preferred over FET and HEMT switches and PIN diodes because of their excellent RF performances [2-6] like low insertion loss and high return loss, very low intermodulation distortion, zero power consumption, miniature size and light weight .There are also some drawbacks like high switching
time and actuation voltage[16]. The design in this paper requires quite low actuation voltage and has a little bit faster switching. Higher return loss and tuning of resonant frequency can be achieved by using tuned switches [4] or by placing few switches to add to the capacitance, or by a single inductively-tuned MEMS shunt switch. Various works has been done by combining the concept of CSRR structure with MEMS [17-20]. One of important advantage of using CSRR structure is that it help in achieving tunability and makes the performance of the device better[21-23]. The dimensions of the CSRR structures can be changed to tune in to the required frequency range [24-27] and the CSRR structure is etched in the signal line of the CPW to construct the band pass filter [28-32]. Coplanar Waveguide (CPW) has many advantages; one of most important advantage is that it allows easy implementation of active and passive devices. CPW is a convenient structure to integrate CSRRs and SRRs [32]. This work describes progress toward designing and modelling of reconfigurable band pass filters based on RF micro electromechanical systems (RF-MEMS) metamaterials and usable phase shifting performance for satellite communication. The explicit use of the k –ka band for radar and satellite communication increases demands for the need of band pass filter. In this paper CPW based tunable Band pass filter is designed. CSRR structures and MEMS switches are implemented, various material and parameter variations are done to achieve tunability and higher Insertion loss and a low Return loss. Device fabrication is also done to realize the device performance. Proposed design: The proposed model of RF MEMS band pass filter is designed on coplanar waveguide (CPW). The dimensions of CPW (shown in Table 1) are calculated to ensure proper impedance matching. Various parametric analysis are carried on to choose material for the substrate, switches and dielectric layers and to deliberate the dimensions, which are discussed in the next section. Through optimization ten distributed MEMS shunt switches are deliberated [24], of which five have been positioned at the input terminal and the other 5 at the output terminal to increase the capacitance and achieve tuning of the resonant frequency. A thin dielectric layer is used to avoid shorting of the switch with the signal line. Dimensions related to switch and dielectric are shown in
Table 2. On the signal line three CSRR structures are embedded to increase the electrical length and hence the inductance.
Figure 1 Equivalent circuit model of the CSRR structure CSRRs can be effectively excited by electric fields and are more suitable for implementation in microstrip circuits than SRRs in view of the electromagnetic (EM) field distributions of microstrip lines. Introduction of CSRR is based on change of inductance and capacitance value by changing current path. In this design 1st open end represents series path where line inductance ( LLine ), gap capacitance ( C Gap ) is connected in series. Coupling capacitance ( C couple ) is the connective capacitance of 2nd ring. Effective inductance ( Leff ) and capacitance ( C eff ) is connected in parallel which shown in Figure 1. The shunt and series branch presents inductive impedance between
1/
[L (C eff
couple
]
+ Ceff ) and 1 /
[L
eff
]
C eff , which denotes equivalent negative permittivity and
capacitive impedance when the frequency is less than 1 /
[L
Line
]
CGap , which denotes equivalent
negative permeability. Figure 3 shows the schematic representation of CSRR structure. Table 2 shows dimensions of the CSRR structures. The top view of the proposed structure is shown in Figure 2. The proposed structure is miniaturized and have high return loss and low insertion loss. Table 3 shows dimensions of the RF-MEMS switch structure. Figure 4 represents the schematic representation of RF-MEMS switch (side view and 3D view).
Figure 2 Schematic representation of proposed CSRR based band pass filter structure (Top view) Table 1 CPW design parameters and measurements of RF-MEMS switch
Abbreviations Design parameters GWG
G/W/G of Ground
Measurements (µm) 30/50/30
Wgm
Width of Ground at middle, at input & at output
80
Wfs
Width of Ground in front of Switches
120
Lsig
Length of Signal Line
2000
Lio
Length of Signal Input and output
720
Wdev
Width of the device
350
Lsub
Length of the Substrate
2100
Wsub
Width of the Substrate
400
Tsub
Thickness of the Substrate
275
Tg
Thickness of the Ground AxBxCxDxE
2 30 x 60 x 640 x 50 x 20
Wge
Width of ground 1st end
40
Lse
Length of signal line 1st end
70
Lsc, Wsc
Length & width of signal central portion
520 x100
Figure 3 Schematic representation of proposed CSRR structure
Abbreviations
Table 2 Design parameters of the CSRRs structure Design Parameters Measurements(µm)
Lor
Length of outer ring of CSRR
126
Lir
Length of inner ring of CSRR
86
WCSR1
Width of 1st CSRR
80
Wring
Width of rings
10
Sr
Spacing between rings
10
Sfr, Ssr
Spacing between end of 1st ring and 2nd ring
30 & 10
WCSR2
Width of 2nd CSRR
80
Figure 4 Schematic representation of RF-MEMS switch (side view and 3D view) Table 3 Parameters and measurements of MEMS Switches and Dielectric Layer
Abbreviations Parameters
Measurements (µm)
Lb
Length of the Bridge
150
Wb
Width of the Bridge
20
Tbr
Thickness of the Bridge
1
Ssw
Spacing between switches
80
Sls
Space right after the last switch
40
La
Length of the Anchor
20
Ssa
Spacing between signal line and anchor
0.5
Ta
Thickness of the Anchor
2
Ldi
Length of the Dielectric Layer
20
Wdi
Width of the Dielectric Layer
50
Tdi
Thickness of the Dielectric Layer
0.5
g0
Gap between the MEMS Switch and Dielectric Layer at Upstate Condition
1
Optimization of electromagnetic (EM) and electromechanical performances: To observe the switching performance of the proposed tuneable band pass filter, essential electromechanical analysis is done. All electromechanical performances are observed through mathematical analysis. In this section, different parameters like material of the bridge, dielectric layer, and substrate, thickness of dielectric layer, air gap height and width of the bridges of proposed device are analyzed for their various electrical and mechanical properties. Simulation and optimization of electromagnetic behaviour for the essential k band is done by using AnsysEM HFSS and modelled mechanical parameters are simulated using MATLAB software.
I.
Comparison of material properties for the bridge: Various material properties like the
young’s modulus, poisson’s ratio ,mass density, tensile strength, spring contant that plays an important role in material (Al, Au, Cu, Pt and Ni) selection are shown in Table 4 (detas are extracted by using equation (2) and (5)). By using previous data spring constant and switching time are calculated for the beam . The spring constant ( K ) due to a uniform force applied over the beam of bridge is given in equation (1). 3
t t K = 32Ew + 8σ (1 − v) w l l
[21]
(1)
Where, E is Young’s modulus ( E = 80 GPa for gold), σ is residual stress of the beam, and v is Poisson’s coefficient ( v = 0.42 for gold), w is the width, t is the thickness and l is the length of
the beam. Schematic representation of single RF-MEMS switch is shown in Figure 5 (electrostatic and magneto static force are shown by up and down arrow). For the ideal case the equation can be modified as equation (2).
t K = 32 Ew l
3
[33]
(2)
Where residual stress ( σ ) = 0.
Figure 5 Schematic representation of single RF-MEMS switch for (mathematical representation) Table 4 Comparison of material properties for the bridge Materials
Young’s Modulus, E (GPa)
Poisson’s ratio ,υ
Strength (MPa)
Conductivity, Spring constant , k σ (S/m) (N/m)
Switching time(µs)
0.33
Mass density, ( g / cm 3 ) 2.70
Al
69
117
3.5e7
7.19
0.016
Au
79
0.42
19.3
100
4.1e7
8.23
0.02
Cu
115
0.33
21.45
220
5.9e7
11.9
0.03
Pt
168
0.39
8.902
240
1.43e7
17
0.011
Ni
200
0.31
3.10
450
9.43e6
20
0.017
Figure 6 Young’s Modulus, E (GPa) Vs Tensile strength (MPa) of Al, Au,Cu,Pt and Ni Ni is best material
with high tensile strength and Young’s Modulus (shown in Figure 6) are
considered, as higher value of Young’s Modulus and tensile strength suggest the material will deform less and will be less prone to failure. But a lower value of young’s Modulus and Poisson’s ratio is considered to reduce the pull in voltage.
Figure 7 Change of Gap height, g (micron) with pull-in voltage, Vp (V) for Al, Au,Cu,Pt and Ni for initial gap height of 3um
Low pull-in voltage ( v p ) is another important required parameter for switching optimization. Pull-in voltage for the proposed model can be expressed as equation (3).
vp =
2k g 2 ( g 0 − g ) [21] ε 0Ww
(3)
Where, g 0 is the initial height taken & g is the height of the beam above the signal line. Gap height is maintained to 2 g 0 / 3 because; the increase in the electrostatic force of the beam is greater than the increase in the restoring force. So the equation (3) is modified to equation (4) (please see the notation of Figure 5).
V p = V (2 g 0 / 3) =
8k 3 g 0 [21] 27ε 0Ww
(4)
Figure 7 shows the variation pull-in voltage with air gap height for various materials. Aluminium offers the lowest pull-in voltage and nickel offers the maximum pull-in voltage. Copper can be a good trade off, the conductivity of copper is higher than that of Aluminium, which means lesser skin depth and therefore a low loss transmission.
Figure 8 Switching time(s) Vs Spring constant (N/m) of Al, Au,Cu,Pt and Ni
Switching time related to pull-in voltage and expressed as equation (5).
t s = 3.67v p /(v s w0 ) (5)
Where w0 is angular frequency and v s = 1.4v p . Figure 8 shows the switching time variation with spring constant for different materials (Al, Au, Cu, Pt and Ni) and it is seen that Aluminium Bridge has the fastest switching among all the other considered materials. From the calculated values of switching time as shown in Table 4 nickel and aluminium can be used to achieve faster switching.
Figure 9 Young’s Modulus, E (GPa) Vs Mass density, ρ (Mg/ ) of Al,Au, Cu,Pt and Ni Figure 9 helps in selecting the most light and stiff material .This plot shows that copper has a much higher mass density and so it cannot be used for this purpose. Considering all the factors aluminium is best choice for beam material.
II.
Optimization of air gap height: By varying the air gap height only while keeping all other
parameters constant,the capacitance of the switches can be adjusted to get the required shift in the resonant frequency. The best result is obtained for an air gap height of 0.5um. Reducing the air gap height also reduces pullin voltage.
0
-5
S11 for g0 =0.5um
S Parmeter (dB)
-10
S11 for g0 =1um S11 for g0 =1.5um
-15
S11 for g0 =2um S11 for g0 =2.5um S11 for g0 =3um
-20
S21 for g0 =0.5um S21 for g0 =1um
-25
S21 for g0 =1.5um S21 for g0 =2um S21 for g0 =2.5um
-30
S21 for g0 =3um
-35 10
15
20
25
30
35
40
45
50
Frequency (GHz)
Figure 10 S Parameter variation for different gap heights without CSRR .
Figure 10 shows the simulated insertion loss and return loss for different air gap heights.The simulation is done by varying the airgap height for a constant value of dielectric thickness of 0.5um and bridge width of 20um and silicon substrate is used. From Figure 10 it can be noted that as the gap height reduces there is a left shift in the resonant frequency . To get the resonant frequency in the k – band 0.5um air gap height can be considered. As gap height is inversely proportional to up and down capacitance. When gapheight decreases then capacitance increases and as a result resonant frequency shifted to the lower value. Also lower gap has disadvantage in terms of tunability. The gap height is maintained 2 g 0 / 3 to get stable performance. If the gap height is more than 2 g 0 / 3 it is seen in Figure that the displacement is zero (shown in Figure 11). For that reason an air gap height of 1um
may be considered which gives a resonant frequency of 30.60 GHz. To reduce the resonant frequency further CSRR structures can be implemented.
Figure 11 Comparison of pull-in-voltage of MEMS switch for two different gap height (1um and 0.5um).
0
S Parameter(dB)
-5
-10
S11 with CSRR
-15
S21 without CSRR S11 with CSRR
-20
S21 without CSRR -25
-30 10
15
20
25
30
35
40
45
Frequency(GHz)
Figure 12 S Parameter variation for different gap heights with CSRR
50
It can be observed in Figure 12 that with the inclusion of the CSRR there is a left shift in the resonant frequency to 29.39 GHz from 30.60GHz.
Figure 13 Pullin voltage,Vp(V) Vs gap height,g (micron) Figure 13 depicts the pull in voltage for different beam materials . As shown in Fig 4 for an initial air gap height of 3 um the pull in voltage was about 80V for aluminium as beam material. With a mere reduction of air gap height to 1 um , the pullin voltage considerebly reduces to 15.34V.
III.
Optimization of width of the switches:-Changing the width of the switch will change the
actuation area and it is the most suitable way to tune the centre frequency while maintaining a low pull in voltage. The pull in voltage is independent from any variation of the beam width. Portion of the bridge above the signal line determines the capacitance and the portion above the gap determine the inductance. Hence increasing the width of the bridge will increase both inductance and capacitance and that will shift the resonant frequency to the left.
0
Sparameters(dB)
-5
-10
S11 for w=10um
-15
S11 for w= 30um
S11 for w=20um S11 for w=40um S11 for w=50um
-20
S21 for w=10um S21 for w=20um
-25
S21 for w=30um S21 for w=40um
-30
S21 for w=50um
-35 10
15
20
25
30
35
40
45
50
Frequency(GHz)
Figure 14 S Parameter variations for change in width of the bridge. Figure 14 shows the simulated result of the shift in resonant frequency for variation in the beam width. The simulations are done for a gap height. Variation from 10 um to 50 um is shown. Increasing the width reduces the resonant frequency. Width of the switch is not only related to electrical resonant frequency but also mechanical resonant frequency. Reduced width decreases spring constant and switching time and, which increases switching performance. Also, the value of capacitance is depends on area of bridge and, which is related to the electrical resonant frequency. The sharpest notch is obtained for width of 40um. For 40um the resonant frequency is 28.18 GHz and RL is-30.64dB
IV.
Comparison of material properties for dielectric layer:-The dielectric layer is placed
on the signal line under the switches to avoid shorting. Relative permitivity plays an important role in choosing the material for dielectric layer as it affects the hold down voltage and the capacitance of the shunt switches [25]. Dielectric charging is common phenomenon that challenges the reliability of the switches[25-28]. The switching performance and electrical resonant frequency is depends on parallel plate (signal line and bridge) capacitor, which is expressed in equation (6).
C p = ε 0Ww / ( g 0 + (t d / ε rd )) [33] Where, ε rd is the relative dielectric constant (please follow Figure 3 for other parameters).
(6)
For applied pull-in voltage created up and down capacitance can be expressed equation (7) and (8).
C up = C p (1 + Pf ) [33]
(7)
C down = (ε 0ε rd Ww) / t d
(8)
While calculating up state capacitance 25% fringing field ( Pf ). To achieve better tunability optimization of up and down capacitances are required, because the electrical resonant frequency is inversely proportional to the value of capacitance. Table 5 Comparison of relative permiability for various dielectric materials
Material SiO2 Si3N4 Al2O3 ZnO AlN HfO2 BST
Relative permitivity 3.9 7.4 9.8 10.8 9.8 20 31 800
Table 5 shows the dielectric constant or relative permitivity of various materials that can be used for dielectric layer. BST with an dielectric constant of 800, will help in achieving greater capacitance(shown in Figure 15), which in turn will reduce the resonant frequency.
Figure 15 Capacitance Vs dielectric thickness
Figure 16 Hold down voltage Vs spring constant Figure 16 shows the variation of hold down voltage with spring constant for various materials. For BST the Hold down voltage remains almost constant and is negligible for variation in spring constant.
Figure 15 suggests that ,increasing the thickness of dielctric layer, reduces the capacitance linearly. For dielectric material SiO2, Si3N4 the capacitance can be tuned by changing the dielectric thickness to some extent but for BST the capacitance is highest almost independent of the thickness of the dielectric.
Figure 17 Capacitance Vs air gap height Capacitance decreases with increasing gap height. Figure 17 shows the variation of capacitance with airgap height ,which is almost similar for all the materials.
0 -5
S parameter(dB)
-10 -15
S11 for SiO2
-20
S21for SiO2
-25
S11 for BST
-30
S21for BST
-35 -40 -45 10
15
20
25
30
35
40
45
50
Frequency(GHz)
Figure 18 S Parameter variation with frequency for dielectric material (BST and SiO2 ). Figure 18 shows the device performance for SiO2 and BST as dielectric material. As there is a very small deviation of the device performance for the different materials, S parameter variations for only two materials are shown. SiO2 has a resonant frequency of 28.18 GHz, Return loss of -40.9 dB. Insertion loss of -0.85 dB. Though BST with high dielectric permittivity of 800 reduces the resonant frequency to 27.7 GHz but deteriorates the return loss (-29dB) and insertion loss (-0.91dB). Silicon dioxide and silicon nitride have better compatibility with silicon in terms of compressive and tensile stress. Sub-stoichiometric silicon dioxide the defect gives rise to the formation of dipoles by trapping holes. This decorates the electric performance of the device. Considering all the factors it can be considered that Si3 N 4 is a good trade off between low loss and lower resonant frequency. As it helps in achieving a low resonant frequency as well as a deeper return loss and lower insertion loss.
V.
Optimization of dielectric thickness:-Dielectric thickness affects the capacitance, resonant frequency and hold down voltage. Effect on capacitance was discussed in the previously in Figure 12. Reducing the dielectric thickness increases the capacitance and so shifts the resonant frequency to the left. To prevent dielectric charging the thickness should not be greater than 0.5um and the lower value is limited by dielectric breakdown and usually taken not below 0.1um. Since in this design the actuation voltage is very small, the thickness can be reduced further.
Figure 19 S-Parameter variation for different dielectric thickness.
Figure 19 illustrates the return loss and insertion loss variation with frequency for different dielectric thickness. Decreasing the dielectric thickness left shifts the resonant frequency. But with increasing thickness the return loss reduces. The best result is got for a thickness of 0.5 um. Here the variations are done for a air gap height of 0.5um.
VI.
Comparison of material properties for substrate:-The various materials considered for the substrate and their corresponding properties are shown in Table 6. Variation of return loss and insertion loss with frequency is shown in Fig4. From the comparative analysis it is observed that the resonant frequency decreases with increasing relative permittivity, hence for GaAs substrate, resonant frequency is minimum. A better performance is also obtained for GaAs.
Table 6 Comparison of material properties for substrate.
Material
Resistivity( m)
FR4 DUROID SILICON GLASS QUARTZ POLYSTERENE NELTAC GaAs
10 e11 2 e7 6.4e2 10e10 to 10e14 7.5e17 ~e14 10e11 ~ e8
Relative permitivity 4.4 2.2 11.9 5.5 3.78 2.6 3 12.9
Resonant frequency(GHz) 31.41 39.5 27.37 33.43 32.21 29.79 39.49 26.56
Return loss( )(dB) -33.21 -30.31 -31.19 -40.32 -33.32 -31.09 -31 -41.34
Insertion loss( )(dB) -0.56 -0.43 -0.99 -0.55 -0.4 -0.36 -0.44 -0.96
Figure 20 Return loss and insertion loss Vs frequency curves for various substrate materials. Figure 20 illustrates the variation of S parameters for different substrate materials. The simulations are done for a gap height of 0.5um, dielectric thickness of 0.5um and width of the bridges are fixed at 20 um. For Gallium Arsenide a sharp notch is obtained at the resonant frequency of 26.56 GHz , Insertion loss obtained resonant frequency is -0.96dB and return loss is -41.34dB (shown in Figure 21). Hence gallium arsenide can be considered for substrate material. Gallium arsenide has a dielectric constant of 12.9. Because of its high dielectric constant dielectric loss increases this can be compensated using air in between ground and the substrate.
Figure 21:-variation of S Parameters with inclusion of air in the substrate Using air the effective dielectric constant reduces and as a result the insertion loss and return loss both reduces to -40.13 dB and -0.66 dB respectively. The proposed design has achieved a very low pullin voltage of 15.3V and has a switching time of 1.6µs. The switches have an upstate capacitance of 16.58fF and a down state capacitance of 26.51Ff. Table 7 Various mechanical properties of the proposed device
Capacitance ratio 1.59
Spring constant 7.19/14.3 N/m
Pull-in voltage 15.3 V
Hold-down voltage 2 V
Switching time 1.65/1.16 µs
Table 7 shows the different mechanical properties of the proposed device obtained after the Electromagnetic analysis of various dimensions and materials.
Figure 22 Distributed current path of the proposed device at the upstate condition. In Figure 22, the current path is shown for complete RF-MEMS switch in the upstate condition. As expected, most of the current is concentrated at the edges of the lines. The degree of symmetry of the current distribution is shown by plotting the integrated volume current of ground, signal and CSRR. The symmetric switch design (with 10 switches) has a better balanced current distribution on the signal line from port 1 to port 2 is divided in the first switch to 10th switch. One of the easiest ways to implement a digital phase shifter is by using the switched delay-line technique. The phase shift increases linearly with with subsequent delay. In this desing useable 11 o phase shift is acheaved with subsequent up and down condition of 10 switches.
Fabrication and results and discussion: High-resistivity (∼5kΩ-cm) silicon wafers used as substrate to fabricate RF-MEMS switch. Different stages of Fabrication in RF-MEMS devise is shown in Figure 23. Pyrogenic oxidation is used to grow 1-micron passivation layer of SiO2 on the wafer surface. Gold have chosen as the metal substrate to fabricate RF-MEMS switch. Although Aluminium offers the lowest pull-in voltage and fastest switching over all the other considered materials, but, Gold has a high capacitance ratio comparison to aluminium which in turn gives high isolation in the circuit. Because gold has higher stress to pulldown the bridge maximum level. It is also discussed in the previous section that beam becomes unstable at and suddenly drops down. Thick (500nm) gold layer is deposited by thermal-evaporation technique and deposited gold layer, oxide layer was patterned using photolithography technique to fabricate the CPW t-line (refer to Fig. 23 (a), (b1) & (b2)). Then 200nm thin Si3N4 is deposited by Hot Wire Chemical Vapour Deposition(HWCVD) technique and then patterned by lithography also etched by reactive ion etching technique (gas mixture: carbon tetra fluoride (CF4) and oxygen (O2)
plasma) . To fabricate the both the fixed anchors and the sacrificial layer photolithography technique with negative photoresist SU-8 2002 is used for the same realization (refer to Fig. 23 (c1) & (c2)). Here one portion of SU-8 2002 is used as anchor and another as sacrifasial layer. A layer of Gold (200 nm) is used as a metal layer for the 5-bridge pattern (refer to Fig. 23 (d1) & (d2)).
Figure 23 Optical microscopic images of fabricated RF-MEMS switch: (a) pattern is created on gold surface, (b1) etched gold CPW end view, and (b2) etched gold CPW centre view, (c1) SU8 deposited on the CPW line to form bridge, (c2) SU8 deposited on CPW to get increased height of the anchor, (d1) CSRR structure with gold bridge, (d2) five consecutive floating bridge.
Measured RF characterization result is shown in Figure 20, 21. The device attains notch level of 42.76dB for upstate and -33.45dB for downstate. An insertion loss of -0.90 dB in upstate and -1.06 dB in the downstate are obtained. The resonant frequency in the upstate is 27.37GHz and 26.16GHz in the downstate. Hence with the movement of the switch from upstate to downstate a tuning of 4 % is obtained which is portrayed in Figure 24. A Phase shift of about 11° is obtained at 27.37 GHz as shown in Figure 25.
0
S parameters(dB)
-10
-20
S11 for downstate S11 for upstate S21 for downstate
-30
S21 for upstate
-40 10
15
20
25
30
35
40
45
50
Frequency(GHz)
Figure 24 Measured RF characterization of the fabricated device in the upstate and downstate
200
upstate dow nstate
S21(Degree)
100
0
-100
-200 10
20
30
40
50
F req uency(G H z)
Figure 25 Phase shift between the upstate and downstate of the proposed device
Conclusion: The comprehensive study of k band based Band pass filter is done where tunability dispersion is achieved using RF MEMS switches and optimization technique is used to get the appreciable performance of the device in the preferred band. 4% tunability is achieved in 27.37GHz and 26.16 GHz frequency range with high level of notch -42.76 dB for upstate and -33.45 dB for down state. The MEMS based switching device is applicable for satellite communication purpose. Acknowledgement: Author would like to acknowledge INUP IITB, NIT Arunachal Pradesh, and Tezpur University for logistic support.
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S1434-8411(16)30820-2 http://dx.doi.org/10.1016/j.aeue.2016.11.021 AEUE 51737
To appear in:
International Journal of Electronics and Communications
Received Date: Accepted Date:
25 September 2016 22 November 2016
Please cite this article as: A. Choudhury, S. Maity, Design and fabrication of CSRR based tunable mechanically and electrically efficient band pass filter for K-band application, International Journal of Electronics and Communications (2016), doi: http://dx.doi.org/10.1016/j.aeue.2016.11.021
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Paper Title: “Design and fabrication of CSRR based tunable mechanically and electrically efficient band pass filter for K-band application” We are ensuring that the paper is not submitted elsewhere. The detailed of the authors are given belowAnwesha Choudhury, M.Tech
Santanu Maity, Ph.D
Research Scholar, Department of Electronics and Assistant Professor, Department of Electronics Communication Engineering, National Institute and Of
Technology,
Arunachal
Engineering,
National
Pradesh-971112, Institute Of Technology, Arunachal Pradesh-
Email: [email protected]
Thanks & Regards. Dr. Santanu Maity (Corresponding Author)
Communication
971112, Email: [email protected]
Design and fabrication of CSRR based tunable mechanically and electrically efficient band pass filter for K-band application Anwesha Choudhury1, Santanu Maity1* Department of Electronics and Communication Engineering National Institute Of Technology, Arunachal Pradesh-971112 Email: [email protected]
Abstract: The paper represents the design and fabrication of a coplanar wave guide (CPW) based Band pass filter for K – band application. In this paper for getting the desired resonant frequency with a good performance (in terms of notch level) modification of the signal line and switches are carried out. Ten distributed switches are implemented to attain the tunability. The simulated result shows that the insertion loss and return loss is improved by using the (CSRR) structure. Detailed Electro mechanical analysis of various materials and parameters are done to make the device applicable for satellite communication. The simulated and fabricated devices operating in the frequency range of 26.16-27.37 GHz has attained a Return loss of -42.76 dB and an Insertion loss of-0.9dB. It gives a tuning of 4%. The device works with a low pull-in voltage of 15.3 V. Keywords: Micro Electro Mechanical System (MEMS), Coplanar Waveguide (CPW), Band pass filter, Complementary split ring resonator (CSRR) Introduction:
Micro Electro Mechanical System (MEMS) is a technology of miniature devices
formed by combination of micromechanical and microelectronic components on a common substrate [1] .RF MEMS technology is rapidly evolving and hence new applications are continuously being reported [7-15]. RF MEMS switches, switched capacitors, tunable inductors, resonators, filters are preferred over FET and HEMT switches and PIN diodes because of their excellent RF performances [2-6] like low insertion loss and high return loss, very low intermodulation distortion, zero power consumption, miniature size and light weight .There are also some drawbacks like high switching
time and actuation voltage[16]. The design in this paper requires quite low actuation voltage and has a little bit faster switching. Higher return loss and tuning of resonant frequency can be achieved by using tuned switches [4] or by placing few switches to add to the capacitance, or by a single inductively-tuned MEMS shunt switch. Various works has been done by combining the concept of CSRR structure with MEMS [17-20]. One of important advantage of using CSRR structure is that it help in achieving tunability and makes the performance of the device better[21-23]. The dimensions of the CSRR structures can be changed to tune in to the required frequency range [24-27] and the CSRR structure is etched in the signal line of the CPW to construct the band pass filter [28-32]. Coplanar Waveguide (CPW) has many advantages; one of most important advantage is that it allows easy implementation of active and passive devices. CPW is a convenient structure to integrate CSRRs and SRRs [32]. This work describes progress toward designing and modelling of reconfigurable band pass filters based on RF micro electromechanical systems (RF-MEMS) metamaterials and usable phase shifting performance for satellite communication. The explicit use of the k –ka band for radar and satellite communication increases demands for the need of band pass filter. In this paper CPW based tunable Band pass filter is designed. CSRR structures and MEMS switches are implemented, various material and parameter variations are done to achieve tunability and higher Insertion loss and a low Return loss. Device fabrication is also done to realize the device performance. Proposed design: The proposed model of RF MEMS band pass filter is designed on coplanar waveguide (CPW). The dimensions of CPW (shown in Table 1) are calculated to ensure proper impedance matching. Various parametric analysis are carried on to choose material for the substrate, switches and dielectric layers and to deliberate the dimensions, which are discussed in the next section. Through optimization ten distributed MEMS shunt switches are deliberated [24], of which five have been positioned at the input terminal and the other 5 at the output terminal to increase the capacitance and achieve tuning of the resonant frequency. A thin dielectric layer is used to avoid shorting of the switch with the signal line. Dimensions related to switch and dielectric are shown in
Table 2. On the signal line three CSRR structures are embedded to increase the electrical length and hence the inductance.
Figure 1 Equivalent circuit model of the CSRR structure CSRRs can be effectively excited by electric fields and are more suitable for implementation in microstrip circuits than SRRs in view of the electromagnetic (EM) field distributions of microstrip lines. Introduction of CSRR is based on change of inductance and capacitance value by changing current path. In this design 1st open end represents series path where line inductance ( LLine ), gap capacitance ( C Gap ) is connected in series. Coupling capacitance ( C couple ) is the connective capacitance of 2nd ring. Effective inductance ( Leff ) and capacitance ( C eff ) is connected in parallel which shown in Figure 1. The shunt and series branch presents inductive impedance between
1/
[L (C eff
couple
]
+ Ceff ) and 1 /
[L
eff
]
C eff , which denotes equivalent negative permittivity and
capacitive impedance when the frequency is less than 1 /
[L
Line
]
CGap , which denotes equivalent
negative permeability. Figure 3 shows the schematic representation of CSRR structure. Table 2 shows dimensions of the CSRR structures. The top view of the proposed structure is shown in Figure 2. The proposed structure is miniaturized and have high return loss and low insertion loss. Table 3 shows dimensions of the RF-MEMS switch structure. Figure 4 represents the schematic representation of RF-MEMS switch (side view and 3D view).
Figure 2 Schematic representation of proposed CSRR based band pass filter structure (Top view) Table 1 CPW design parameters and measurements of RF-MEMS switch
Abbreviations Design parameters GWG
G/W/G of Ground
Measurements (µm) 30/50/30
Wgm
Width of Ground at middle, at input & at output
80
Wfs
Width of Ground in front of Switches
120
Lsig
Length of Signal Line
2000
Lio
Length of Signal Input and output
720
Wdev
Width of the device
350
Lsub
Length of the Substrate
2100
Wsub
Width of the Substrate
400
Tsub
Thickness of the Substrate
275
Tg
Thickness of the Ground AxBxCxDxE
2 30 x 60 x 640 x 50 x 20
Wge
Width of ground 1st end
40
Lse
Length of signal line 1st end
70
Lsc, Wsc
Length & width of signal central portion
520 x100
Figure 3 Schematic representation of proposed CSRR structure
Abbreviations
Table 2 Design parameters of the CSRRs structure Design Parameters Measurements(µm)
Lor
Length of outer ring of CSRR
126
Lir
Length of inner ring of CSRR
86
WCSR1
Width of 1st CSRR
80
Wring
Width of rings
10
Sr
Spacing between rings
10
Sfr, Ssr
Spacing between end of 1st ring and 2nd ring
30 & 10
WCSR2
Width of 2nd CSRR
80
Figure 4 Schematic representation of RF-MEMS switch (side view and 3D view) Table 3 Parameters and measurements of MEMS Switches and Dielectric Layer
Abbreviations Parameters
Measurements (µm)
Lb
Length of the Bridge
150
Wb
Width of the Bridge
20
Tbr
Thickness of the Bridge
1
Ssw
Spacing between switches
80
Sls
Space right after the last switch
40
La
Length of the Anchor
20
Ssa
Spacing between signal line and anchor
0.5
Ta
Thickness of the Anchor
2
Ldi
Length of the Dielectric Layer
20
Wdi
Width of the Dielectric Layer
50
Tdi
Thickness of the Dielectric Layer
0.5
g0
Gap between the MEMS Switch and Dielectric Layer at Upstate Condition
1
Optimization of electromagnetic (EM) and electromechanical performances: To observe the switching performance of the proposed tuneable band pass filter, essential electromechanical analysis is done. All electromechanical performances are observed through mathematical analysis. In this section, different parameters like material of the bridge, dielectric layer, and substrate, thickness of dielectric layer, air gap height and width of the bridges of proposed device are analyzed for their various electrical and mechanical properties. Simulation and optimization of electromagnetic behaviour for the essential k band is done by using AnsysEM HFSS and modelled mechanical parameters are simulated using MATLAB software.
I.
Comparison of material properties for the bridge: Various material properties like the
young’s modulus, poisson’s ratio ,mass density, tensile strength, spring contant that plays an important role in material (Al, Au, Cu, Pt and Ni) selection are shown in Table 4 (detas are extracted by using equation (2) and (5)). By using previous data spring constant and switching time are calculated for the beam . The spring constant ( K ) due to a uniform force applied over the beam of bridge is given in equation (1). 3
t t K = 32Ew + 8σ (1 − v) w l l
[21]
(1)
Where, E is Young’s modulus ( E = 80 GPa for gold), σ is residual stress of the beam, and v is Poisson’s coefficient ( v = 0.42 for gold), w is the width, t is the thickness and l is the length of
the beam. Schematic representation of single RF-MEMS switch is shown in Figure 5 (electrostatic and magneto static force are shown by up and down arrow). For the ideal case the equation can be modified as equation (2).
t K = 32 Ew l
3
[33]
(2)
Where residual stress ( σ ) = 0.
Figure 5 Schematic representation of single RF-MEMS switch for (mathematical representation) Table 4 Comparison of material properties for the bridge Materials
Young’s Modulus, E (GPa)
Poisson’s ratio ,υ
Strength (MPa)
Conductivity, Spring constant , k σ (S/m) (N/m)
Switching time(µs)
0.33
Mass density, ( g / cm 3 ) 2.70
Al
69
117
3.5e7
7.19
0.016
Au
79
0.42
19.3
100
4.1e7
8.23
0.02
Cu
115
0.33
21.45
220
5.9e7
11.9
0.03
Pt
168
0.39
8.902
240
1.43e7
17
0.011
Ni
200
0.31
3.10
450
9.43e6
20
0.017
Figure 6 Young’s Modulus, E (GPa) Vs Tensile strength (MPa) of Al, Au,Cu,Pt and Ni Ni is best material
with high tensile strength and Young’s Modulus (shown in Figure 6) are
considered, as higher value of Young’s Modulus and tensile strength suggest the material will deform less and will be less prone to failure. But a lower value of young’s Modulus and Poisson’s ratio is considered to reduce the pull in voltage.
Figure 7 Change of Gap height, g (micron) with pull-in voltage, Vp (V) for Al, Au,Cu,Pt and Ni for initial gap height of 3um
Low pull-in voltage ( v p ) is another important required parameter for switching optimization. Pull-in voltage for the proposed model can be expressed as equation (3).
vp =
2k g 2 ( g 0 − g ) [21] ε 0Ww
(3)
Where, g 0 is the initial height taken & g is the height of the beam above the signal line. Gap height is maintained to 2 g 0 / 3 because; the increase in the electrostatic force of the beam is greater than the increase in the restoring force. So the equation (3) is modified to equation (4) (please see the notation of Figure 5).
V p = V (2 g 0 / 3) =
8k 3 g 0 [21] 27ε 0Ww
(4)
Figure 7 shows the variation pull-in voltage with air gap height for various materials. Aluminium offers the lowest pull-in voltage and nickel offers the maximum pull-in voltage. Copper can be a good trade off, the conductivity of copper is higher than that of Aluminium, which means lesser skin depth and therefore a low loss transmission.
Figure 8 Switching time(s) Vs Spring constant (N/m) of Al, Au,Cu,Pt and Ni
Switching time related to pull-in voltage and expressed as equation (5).
t s = 3.67v p /(v s w0 ) (5)
Where w0 is angular frequency and v s = 1.4v p . Figure 8 shows the switching time variation with spring constant for different materials (Al, Au, Cu, Pt and Ni) and it is seen that Aluminium Bridge has the fastest switching among all the other considered materials. From the calculated values of switching time as shown in Table 4 nickel and aluminium can be used to achieve faster switching.
Figure 9 Young’s Modulus, E (GPa) Vs Mass density, ρ (Mg/ ) of Al,Au, Cu,Pt and Ni Figure 9 helps in selecting the most light and stiff material .This plot shows that copper has a much higher mass density and so it cannot be used for this purpose. Considering all the factors aluminium is best choice for beam material.
II.
Optimization of air gap height: By varying the air gap height only while keeping all other
parameters constant,the capacitance of the switches can be adjusted to get the required shift in the resonant frequency. The best result is obtained for an air gap height of 0.5um. Reducing the air gap height also reduces pullin voltage.
0
-5
S11 for g0 =0.5um
S Parmeter (dB)
-10
S11 for g0 =1um S11 for g0 =1.5um
-15
S11 for g0 =2um S11 for g0 =2.5um S11 for g0 =3um
-20
S21 for g0 =0.5um S21 for g0 =1um
-25
S21 for g0 =1.5um S21 for g0 =2um S21 for g0 =2.5um
-30
S21 for g0 =3um
-35 10
15
20
25
30
35
40
45
50
Frequency (GHz)
Figure 10 S Parameter variation for different gap heights without CSRR .
Figure 10 shows the simulated insertion loss and return loss for different air gap heights.The simulation is done by varying the airgap height for a constant value of dielectric thickness of 0.5um and bridge width of 20um and silicon substrate is used. From Figure 10 it can be noted that as the gap height reduces there is a left shift in the resonant frequency . To get the resonant frequency in the k – band 0.5um air gap height can be considered. As gap height is inversely proportional to up and down capacitance. When gapheight decreases then capacitance increases and as a result resonant frequency shifted to the lower value. Also lower gap has disadvantage in terms of tunability. The gap height is maintained 2 g 0 / 3 to get stable performance. If the gap height is more than 2 g 0 / 3 it is seen in Figure that the displacement is zero (shown in Figure 11). For that reason an air gap height of 1um
may be considered which gives a resonant frequency of 30.60 GHz. To reduce the resonant frequency further CSRR structures can be implemented.
Figure 11 Comparison of pull-in-voltage of MEMS switch for two different gap height (1um and 0.5um).
0
S Parameter(dB)
-5
-10
S11 with CSRR
-15
S21 without CSRR S11 with CSRR
-20
S21 without CSRR -25
-30 10
15
20
25
30
35
40
45
Frequency(GHz)
Figure 12 S Parameter variation for different gap heights with CSRR
50
It can be observed in Figure 12 that with the inclusion of the CSRR there is a left shift in the resonant frequency to 29.39 GHz from 30.60GHz.
Figure 13 Pullin voltage,Vp(V) Vs gap height,g (micron) Figure 13 depicts the pull in voltage for different beam materials . As shown in Fig 4 for an initial air gap height of 3 um the pull in voltage was about 80V for aluminium as beam material. With a mere reduction of air gap height to 1 um , the pullin voltage considerebly reduces to 15.34V.
III.
Optimization of width of the switches:-Changing the width of the switch will change the
actuation area and it is the most suitable way to tune the centre frequency while maintaining a low pull in voltage. The pull in voltage is independent from any variation of the beam width. Portion of the bridge above the signal line determines the capacitance and the portion above the gap determine the inductance. Hence increasing the width of the bridge will increase both inductance and capacitance and that will shift the resonant frequency to the left.
0
Sparameters(dB)
-5
-10
S11 for w=10um
-15
S11 for w= 30um
S11 for w=20um S11 for w=40um S11 for w=50um
-20
S21 for w=10um S21 for w=20um
-25
S21 for w=30um S21 for w=40um
-30
S21 for w=50um
-35 10
15
20
25
30
35
40
45
50
Frequency(GHz)
Figure 14 S Parameter variations for change in width of the bridge. Figure 14 shows the simulated result of the shift in resonant frequency for variation in the beam width. The simulations are done for a gap height. Variation from 10 um to 50 um is shown. Increasing the width reduces the resonant frequency. Width of the switch is not only related to electrical resonant frequency but also mechanical resonant frequency. Reduced width decreases spring constant and switching time and, which increases switching performance. Also, the value of capacitance is depends on area of bridge and, which is related to the electrical resonant frequency. The sharpest notch is obtained for width of 40um. For 40um the resonant frequency is 28.18 GHz and RL is-30.64dB
IV.
Comparison of material properties for dielectric layer:-The dielectric layer is placed
on the signal line under the switches to avoid shorting. Relative permitivity plays an important role in choosing the material for dielectric layer as it affects the hold down voltage and the capacitance of the shunt switches [25]. Dielectric charging is common phenomenon that challenges the reliability of the switches[25-28]. The switching performance and electrical resonant frequency is depends on parallel plate (signal line and bridge) capacitor, which is expressed in equation (6).
C p = ε 0Ww / ( g 0 + (t d / ε rd )) [33] Where, ε rd is the relative dielectric constant (please follow Figure 3 for other parameters).
(6)
For applied pull-in voltage created up and down capacitance can be expressed equation (7) and (8).
C up = C p (1 + Pf ) [33]
(7)
C down = (ε 0ε rd Ww) / t d
(8)
While calculating up state capacitance 25% fringing field ( Pf ). To achieve better tunability optimization of up and down capacitances are required, because the electrical resonant frequency is inversely proportional to the value of capacitance. Table 5 Comparison of relative permiability for various dielectric materials
Material SiO2 Si3N4 Al2O3 ZnO AlN HfO2 BST
Relative permitivity 3.9 7.4 9.8 10.8 9.8 20 31 800
Table 5 shows the dielectric constant or relative permitivity of various materials that can be used for dielectric layer. BST with an dielectric constant of 800, will help in achieving greater capacitance(shown in Figure 15), which in turn will reduce the resonant frequency.
Figure 15 Capacitance Vs dielectric thickness
Figure 16 Hold down voltage Vs spring constant Figure 16 shows the variation of hold down voltage with spring constant for various materials. For BST the Hold down voltage remains almost constant and is negligible for variation in spring constant.
Figure 15 suggests that ,increasing the thickness of dielctric layer, reduces the capacitance linearly. For dielectric material SiO2, Si3N4 the capacitance can be tuned by changing the dielectric thickness to some extent but for BST the capacitance is highest almost independent of the thickness of the dielectric.
Figure 17 Capacitance Vs air gap height Capacitance decreases with increasing gap height. Figure 17 shows the variation of capacitance with airgap height ,which is almost similar for all the materials.
0 -5
S parameter(dB)
-10 -15
S11 for SiO2
-20
S21for SiO2
-25
S11 for BST
-30
S21for BST
-35 -40 -45 10
15
20
25
30
35
40
45
50
Frequency(GHz)
Figure 18 S Parameter variation with frequency for dielectric material (BST and SiO2 ). Figure 18 shows the device performance for SiO2 and BST as dielectric material. As there is a very small deviation of the device performance for the different materials, S parameter variations for only two materials are shown. SiO2 has a resonant frequency of 28.18 GHz, Return loss of -40.9 dB. Insertion loss of -0.85 dB. Though BST with high dielectric permittivity of 800 reduces the resonant frequency to 27.7 GHz but deteriorates the return loss (-29dB) and insertion loss (-0.91dB). Silicon dioxide and silicon nitride have better compatibility with silicon in terms of compressive and tensile stress. Sub-stoichiometric silicon dioxide the defect gives rise to the formation of dipoles by trapping holes. This decorates the electric performance of the device. Considering all the factors it can be considered that Si3 N 4 is a good trade off between low loss and lower resonant frequency. As it helps in achieving a low resonant frequency as well as a deeper return loss and lower insertion loss.
V.
Optimization of dielectric thickness:-Dielectric thickness affects the capacitance, resonant frequency and hold down voltage. Effect on capacitance was discussed in the previously in Figure 12. Reducing the dielectric thickness increases the capacitance and so shifts the resonant frequency to the left. To prevent dielectric charging the thickness should not be greater than 0.5um and the lower value is limited by dielectric breakdown and usually taken not below 0.1um. Since in this design the actuation voltage is very small, the thickness can be reduced further.
Figure 19 S-Parameter variation for different dielectric thickness.
Figure 19 illustrates the return loss and insertion loss variation with frequency for different dielectric thickness. Decreasing the dielectric thickness left shifts the resonant frequency. But with increasing thickness the return loss reduces. The best result is got for a thickness of 0.5 um. Here the variations are done for a air gap height of 0.5um.
VI.
Comparison of material properties for substrate:-The various materials considered for the substrate and their corresponding properties are shown in Table 6. Variation of return loss and insertion loss with frequency is shown in Fig4. From the comparative analysis it is observed that the resonant frequency decreases with increasing relative permittivity, hence for GaAs substrate, resonant frequency is minimum. A better performance is also obtained for GaAs.
Table 6 Comparison of material properties for substrate.
Material
Resistivity( m)
FR4 DUROID SILICON GLASS QUARTZ POLYSTERENE NELTAC GaAs
10 e11 2 e7 6.4e2 10e10 to 10e14 7.5e17 ~e14 10e11 ~ e8
Relative permitivity 4.4 2.2 11.9 5.5 3.78 2.6 3 12.9
Resonant frequency(GHz) 31.41 39.5 27.37 33.43 32.21 29.79 39.49 26.56
Return loss( )(dB) -33.21 -30.31 -31.19 -40.32 -33.32 -31.09 -31 -41.34
Insertion loss( )(dB) -0.56 -0.43 -0.99 -0.55 -0.4 -0.36 -0.44 -0.96
Figure 20 Return loss and insertion loss Vs frequency curves for various substrate materials. Figure 20 illustrates the variation of S parameters for different substrate materials. The simulations are done for a gap height of 0.5um, dielectric thickness of 0.5um and width of the bridges are fixed at 20 um. For Gallium Arsenide a sharp notch is obtained at the resonant frequency of 26.56 GHz , Insertion loss obtained resonant frequency is -0.96dB and return loss is -41.34dB (shown in Figure 21). Hence gallium arsenide can be considered for substrate material. Gallium arsenide has a dielectric constant of 12.9. Because of its high dielectric constant dielectric loss increases this can be compensated using air in between ground and the substrate.
Figure 21:-variation of S Parameters with inclusion of air in the substrate Using air the effective dielectric constant reduces and as a result the insertion loss and return loss both reduces to -40.13 dB and -0.66 dB respectively. The proposed design has achieved a very low pullin voltage of 15.3V and has a switching time of 1.6µs. The switches have an upstate capacitance of 16.58fF and a down state capacitance of 26.51Ff. Table 7 Various mechanical properties of the proposed device
Capacitance ratio 1.59
Spring constant 7.19/14.3 N/m
Pull-in voltage 15.3 V
Hold-down voltage 2 V
Switching time 1.65/1.16 µs
Table 7 shows the different mechanical properties of the proposed device obtained after the Electromagnetic analysis of various dimensions and materials.
Figure 22 Distributed current path of the proposed device at the upstate condition. In Figure 22, the current path is shown for complete RF-MEMS switch in the upstate condition. As expected, most of the current is concentrated at the edges of the lines. The degree of symmetry of the current distribution is shown by plotting the integrated volume current of ground, signal and CSRR. The symmetric switch design (with 10 switches) has a better balanced current distribution on the signal line from port 1 to port 2 is divided in the first switch to 10th switch. One of the easiest ways to implement a digital phase shifter is by using the switched delay-line technique. The phase shift increases linearly with with subsequent delay. In this desing useable 11 o phase shift is acheaved with subsequent up and down condition of 10 switches.
Fabrication and results and discussion: High-resistivity (∼5kΩ-cm) silicon wafers used as substrate to fabricate RF-MEMS switch. Different stages of Fabrication in RF-MEMS devise is shown in Figure 23. Pyrogenic oxidation is used to grow 1-micron passivation layer of SiO2 on the wafer surface. Gold have chosen as the metal substrate to fabricate RF-MEMS switch. Although Aluminium offers the lowest pull-in voltage and fastest switching over all the other considered materials, but, Gold has a high capacitance ratio comparison to aluminium which in turn gives high isolation in the circuit. Because gold has higher stress to pulldown the bridge maximum level. It is also discussed in the previous section that beam becomes unstable at and suddenly drops down. Thick (500nm) gold layer is deposited by thermal-evaporation technique and deposited gold layer, oxide layer was patterned using photolithography technique to fabricate the CPW t-line (refer to Fig. 23 (a), (b1) & (b2)). Then 200nm thin Si3N4 is deposited by Hot Wire Chemical Vapour Deposition(HWCVD) technique and then patterned by lithography also etched by reactive ion etching technique (gas mixture: carbon tetra fluoride (CF4) and oxygen (O2)
plasma) . To fabricate the both the fixed anchors and the sacrificial layer photolithography technique with negative photoresist SU-8 2002 is used for the same realization (refer to Fig. 23 (c1) & (c2)). Here one portion of SU-8 2002 is used as anchor and another as sacrifasial layer. A layer of Gold (200 nm) is used as a metal layer for the 5-bridge pattern (refer to Fig. 23 (d1) & (d2)).
Figure 23 Optical microscopic images of fabricated RF-MEMS switch: (a) pattern is created on gold surface, (b1) etched gold CPW end view, and (b2) etched gold CPW centre view, (c1) SU8 deposited on the CPW line to form bridge, (c2) SU8 deposited on CPW to get increased height of the anchor, (d1) CSRR structure with gold bridge, (d2) five consecutive floating bridge.
Measured RF characterization result is shown in Figure 20, 21. The device attains notch level of 42.76dB for upstate and -33.45dB for downstate. An insertion loss of -0.90 dB in upstate and -1.06 dB in the downstate are obtained. The resonant frequency in the upstate is 27.37GHz and 26.16GHz in the downstate. Hence with the movement of the switch from upstate to downstate a tuning of 4 % is obtained which is portrayed in Figure 24. A Phase shift of about 11° is obtained at 27.37 GHz as shown in Figure 25.
0
S parameters(dB)
-10
-20
S11 for downstate S11 for upstate S21 for downstate
-30
S21 for upstate
-40 10
15
20
25
30
35
40
45
50
Frequency(GHz)
Figure 24 Measured RF characterization of the fabricated device in the upstate and downstate
200
upstate dow nstate
S21(Degree)
100
0
-100
-200 10
20
30
40
50
F req uency(G H z)
Figure 25 Phase shift between the upstate and downstate of the proposed device
Conclusion: The comprehensive study of k band based Band pass filter is done where tunability dispersion is achieved using RF MEMS switches and optimization technique is used to get the appreciable performance of the device in the preferred band. 4% tunability is achieved in 27.37GHz and 26.16 GHz frequency range with high level of notch -42.76 dB for upstate and -33.45 dB for down state. The MEMS based switching device is applicable for satellite communication purpose. Acknowledgement: Author would like to acknowledge INUP IITB, NIT Arunachal Pradesh, and Tezpur University for logistic support.
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