Design of fractal-based CMOS bandpass filter for WirelessHD system

Design of fractal-based CMOS bandpass filter for WirelessHD system

Microelectronics Journal 42 (2011) 1252–1256 Contents lists available at SciVerse ScienceDirect Microelectronics Journal journal homepage: www.elsev...

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Microelectronics Journal 42 (2011) 1252–1256

Contents lists available at SciVerse ScienceDirect

Microelectronics Journal journal homepage: www.elsevier.com/locate/mejo

Design of fractal-based CMOS bandpass filter for WirelessHD system Wei-Yu Chen a,n, Shoou-Jinn Chang a, Min-Hang Weng b, Cheng-Yuan Hung c a

Institute of Microelectronics and Department of Electrical Engineering, Advanced Optoelectronic Technology Center, Center for Micro/Nano Science and Technology, National Cheng Kung University, Taiwan Medical devices and opto-electronics equipment department Metal Industries Research & Development Center, Taiwan c Department of Electronics Engineering and Computer Sciences Tung-Fang Institute of Technology, Taiwan b

a r t i c l e i n f o

a b s t r a c t

Article history: Received 12 February 2011 Received in revised form 3 August 2011 Accepted 9 August 2011 Available online 9 September 2011

We proposed a fractal-based dual-mode bandpass filter (BPF) using a standard CMOS process for application of 60 GHz WirelessHD system. We first investigated the effect of coupling feedlines of I/O ports set at different layer of M3 and M4 layer on the transmission loss of the resonator, and verified the nature coupling of fractal-based dual-mode filter. Experimental result shows that the designed filter with a fractional-bandwidth (FBW) of 23%, an insertion loss about 7 dB and return loss larger than 10 dB. Additionally, two transmission zeros are appeared at the passband edges, thus much improve the selectivity of the proposed CMOS BPF. The result indicates that fractal-based structure is feasible and can meet the requirement in the mm-wave application. & 2011 Elsevier Ltd. All rights reserved.

Keywords: Fractal Dual-mode mm-wave CMOS WirelessHD

1. Introduction The wireless transmission with high capacity and high data rates always plays a desired goal to meet in communication system. Currently, the blooming WirelessHD system servers to organize an industry-led standardization effort to define a nextgeneration wireless digital interface specification for consumer electronics and PC products. Specifically, the WirelessHD specification is based on the unlicensed 60 GHz band with the 7 GHz of continuous bandwidth and a theoretical data rates as high as 25 Gbps. It enables wireless connectivity for uncompressed streaming high definition audio, video and data between source devices and high-definition displays [1]. Moreover, great efforts have been made to integrate the front-end components on the low-resistivity silicon substrate using stand CMOS process recently [2–5]. The CMOS technology offers the potential for the front-end component to realize the RF system-on-chip (SoC). As is well known, the bandpass filter (BPF) is the key component in the front-end circuit of wireless communication system; however, there are few investigations concerning about the WirelessHD application and implemented using standard CMOS process at the same time. Besides, the dual-mode filter is attractive since its advantages of narrow bandwidth, two symmetric transmission zeros, and high-selectivity characteristics in filter frequency response [6,7]. For the consideration of large substrate loss of the low-resistivity silicon up to mm-wave frequencies, the resonator with high quality

factor shall be used. Among the various types of resonator, dualmode resonator is found to have high quality factor. Moreover, the description of geometry of Sierpinski based fractal, designed guideline and the discussions of high quality and miniaturization have been investigated [8,9]. In Refs. [8], it is the first study of the Sierpinski square resonator. The property of resonant behavior and the external quality factor were well analyzed and discussed. In Ref. [9], the dual-mode bandpass filter was realized and implemented on the ultra-thin liquid crystal polymer substrate using four Sierpinski squares with third order. However, it is a challenge to design an mm-wave bandpass filter using standard CMOS process when considering the large substrate loss of the low-resistivity silicon up to mm-wave frequencies. In this paper, we developed a fractal-based dual-mode bandpass filter using a standard CMOS process for the WirelessHD system, as shown in Fig. 1. First, the design of input/output (I/O) ports is analyzed. We investigated the effect of coupling feedlines of I/O ports set at different layer of M3 and M4 layer on the transmission loss of the resonator. Second, we discussed that nature coupling of dual-mode filter based on the Sierpinski structure. Finally, the designed BPF was fabricated and measured to experimentally verify the simulated results of our design.

2. Fractal-based dual-mode CMOS BPF 2.1. Design of the feedlines for the fractal-based BPF

n

Corresponding author. E-mail address: [email protected] (W.-Y. Chen).

0026-2692/$ - see front matter & 2011 Elsevier Ltd. All rights reserved. doi:10.1016/j.mejo.2011.08.004

In this design, the Sierpinski based dual-mode resonator is adopted for the reasons of the high quality factor and

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Fig. 1. (a) Circuit layout and (b) cross section of layer configuration in standard CMOS process. W1 ¼ 15 mm, W2 ¼ 10 mm, L1 ¼ 480 mm, L2 ¼440 mm M1 ¼0.665 mm, M2¼ 0.64 mm, M3 ¼0.64 mm, M4¼ 0.925 mm, IMD1 ¼1.665 mm, IMD2 ¼1.64 mm, IMD3 ¼1.64 mm. (Via: Via hole, M1: Metal-1, M2: Metal-2, M3: Metal-3, M4: Metal-4, IMD1: Metal-1 and thin oxide-1, IMD2: Metal-2 and thin oxide-2, IMD3: Metal-3 and thin oxide-3.)

miniaturization. In our previous study, the resonant frequencies of the Sierpinski square are lower and the external quality factors are higher as the higher iteration is performed. Up to the fourth order Sierpinski square, the external quality factors will achieve the saturation value about 210. Therefore, in this study, the fourth order Sierpinski square are chosen for the design of dual-mode filter [8]. In order to meet the requirement of the WirelessHD system on the silicon substrate, we chose the fourth order of Sierpinski square geometry as the main resonator structure, as shown in Fig. 1. Moreover, since the fractal-based geometry is mainly generated according to the iterative method. When the iterative order is larger than seven, the minimum line-width is smaller than 0.25 mm. Namely, the linewidth in the structure will limit the iterative order of the Sierpinski square when using the stand 0.35 mm CMOS process. The I/O ports with the coupling feedlines are designed to provide the energy transfer to the resonator structure. Since the center frequency of the Sierpinski dual-mode filter is designed at 60 GHz, the side length of 480 mm is derived using Eq. (1) [9] and the slightly discrepancy is well optimized by the EM simulation [10] L1 ¼

150  R pffiffiffiffi fD  er

ðUnit:mmÞ:

ð1Þ

where the resonant frequency ratio, R, is defined as fN/f0, fN is the resonant frequency of the nth iteration order resonator and f0 is the resonant frequency of the patch resonator. fD is the designed frequency (fD ¼60 GHz, R¼0.38). To provide the higher coupling energy in the proposed dualmode filter, we adopted a novel scheme. Namely, the enhanced feedlines are set at M3 layer and the I/O ports are set at M4 layer for the consideration of on-wafer measurement and the lower series resistance due to the thickness of the metal layer. The I/O ports set at M4 layer is connected to M3 layer using the via holes, as shown in Fig. 1(b). To further know the reason why we designed the enhanced feedlines set at M3 layer, the transmission loss of the resonator using the proposed new scheme and conventional scheme of feedlines are simulated and compared. Layout of the resonator structure with the feedlines of I/O port located at M3 layer and M4 layer is shown in Fig. 2(a) and (b), respectively. Note that, the feedlines shown in Fig. 2(b) is set aside the resonator with a gap of 5 mm and a length of 480 mm. The transmission loss of the resonator structure with two different coupling schemes is shown in Fig. 2(c). It is found the magnitude of S21 is about  2.9 dB for the novel scheme using

Fig. 2. Layout of the resonator structure with the feedlines of I/O port located at (a) M3 layer, (b) M4 layer and (c) transmission loss of the resonator structure with two different coupling schemes.

the feedlines set at M3 layer, on the other hand, the magnitude of S21 is about 41.3 dB for the conventional scheme using the feedlines set at M4 layer. It indicates that the feedlines set at M3 layer provides a strong coupling strength than that set at M4 layer, since the feedlines set at M3 layer provides a higher capacitive factor. 2.2. Coupling characteristics To form the filter response, the I/O ports are located orthogonally to the fractal based dual-mode resonator and the perturbation element shall be added in the corner of the resonator [6]. Moreover, it is known for that a BPF based on a dual-mode resonator can achieve elliptic response or the response with two real-axis transmission zeros [7]. The appearance and location of the transmission zeros depends on the cutting or filling of perturbation element. To identify the nature of the coupling between degenerate modes of the proposed fourth order Sierpinski square resonator, we use square cells with an area of 225 mm2 for the basic cell of perturbation element for the cutting or filling, as shown in Fig. 3. A full-wave electromagnetic (EM) simulator [10] is used to characterize the coupling between the degenerate modes. Fig. 4 shows the current distribution of the fourth order Sierpinski square resonator without and with filled or cut perturbation elements. The current distributions on four corners are identical at the resonance frequency, as shown in Fig. 4(a). Therefore, the degenerate resonance modes in the fourth order Sierpinski square resonator do not couple with each other. When seven basic cells are filled at the corner B (or D) on the symmetry axis, as shown in Fig. 4(b), this type of the perturbation would result in an increase in the inductance per unit length within the resonator and consequently an inductive effect exists on the dual-mode Sierpinski square resonator. Additionally, the BPF shows an elliptic response, as confirmed by our simulations shown in Fig. 5. On the other hand, when 13 basic cells are cut at

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Fig. 3. Conception of the cell division.

Fig. 4. Simulated current distribution pattern for the fourth order Sierpinski square resonator at the resonant frequency for a single mode. (a) Without perturbation element, (b) with 7 cells of perturbation element filled at the inner corner and (c) with 13 cells of perturbation element cut at the inner corner.

corner B (and C) on the symmetry axis. At this time, the BPF shows a response with two real-axis transmission zeros, as shown in Fig. 5. Therefore, to achieve a better passband selectivity, the cells filled at the inner corner are used as the perturbation elements for this design, as shown in Fig. 1(a). To further investigate the perturbation’s size on the coupling property, the simulated split resonance frequencies of these degenerate modes of the Sierpinski dual-mode resonator and the corresponding coupling coefficient as functions of perturbation’s size are shown in Fig. 6. It is found that the split between the mode frequencies decreases when the size of the perturbation element filled or cut at the inner corner is less than 18 cells (4050 mm2) and 16 cells (3600 mm2), respectively, while it increases when the size further is larger than 18 cells (4050 mm2) filled or 16 cells (3600 mm2) cut at the inner corner. The corresponding coupling coefficient (K) can be calculated as (1) Fig. 5. Responses of fourth order Sierpinski square filter with elliptic response (Fig. 4(b)) and a response with two real-axis transmission zeros (Fig. 4(c)).

K¼ the corner B (or D), as shown in Fig. 4(c), this kind of the perturbation would increase the capacitance per unit length of the resonator due to the maximum current distribution on the

f22 f12 f22 f12

ð2Þ

where f2 and f1 are the resonant frequency of the mode I and mode II with respect to the degenerate modes, respectively.

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Fig. 6. Simulated two resonant frequencies (mode I and mode II) and coupling coefficient as functions of different size of the perturbation element. Fig. 7. The simulated and measured responses of the proposed CMOS BPF (inserted picture is the die photo).

To meet an optimized filter responses and a better coupling of two generated modes, the perturbation element is filled by 25 cells and the coupling coefficient is about 0.126.

3. Implement and measurement results After careful layout, the proposed dual-mode CMOS BPF, utilizing the fourth order Sierpinski resonator discussed above, was implemented using standard 0.35 mm CMOS process (D35) having a thickness about 4 mm. It is assumed that the dielectric constants of each poly layers within 0.35 mm process are the same, thus the equivalent dielectric constant can be obtained as 3.9 according to the formulation [11] !1 ! N N X X tn ere ¼ tn ð3Þ

signal line, which significantly degrades the inductive quality of the circuit. It can be improved using the advanced standard CMOS process with the thicker thickness i.e. the 0.18 mm or 0.13 mm process. Moreover, using the cells filled at the inner corner as the perturbation element, two transmission zeros appeared at the passband edges are clearly observed, as discussed in Fig. 4(b), thus much improving the selectivity of the proposed CMOS BPF.

4. Conclusions

where tn is the height of the nth substrate layer and ern is the dielectric constant of the nth substrate layer. The physical size of the fabricated CMOS BPF is compact and only 480 mm  480 mm, i.e., approximately 0.192lg by 0.192lg, where lg is the guided wavelength at the center frequency of 60 GHz. The fabricated CMOS filter was measured on the probe station by an HP8510C Network Analyzer after the standard on wafer calibration of Short-Open-Load-Thru (SOLT). Fig. 7 shows the simulated and measured results where the inserted picture is the die photo in measurement. To meet the requirement of measurement, one of the I/O ports is bended. The fabricated CMOS BPF has a measured center frequency of 60 GHz, a fractional-bandwidth (FBW) of 23%, an insertion loss less than 7 dB and a return loss larger than 10 dB. It is known that the circuit quality factor, Q0, is defined as

We have designed and implemented a fractal-based CMOS BPF for the WirelessHD system. Sierpinski based fractal is used as the main dual-mode resonator. Using a new scheme of the feedlines, the coupling strength of the I/O ports can be enhanced. The nature coupling of the dual-mode filter is also verified and discussed. The physical size of the fabricated CMOS BPF is compact and only 480 mm  480 mm, i.e., approximately 0.192lg by 0.192lg, where lg is the guided wavelength at the center frequency of 60 GHz. The measured results of the fabricated CMOS BPF at 60 GHz have a fractional-bandwidth (FBW) of 23%, a return loss larger than 10 dB and an insertion loss about 7 dB, which is due to the close proximity of the ground plane to the signal line, significantly degrading the inductive quality of the circuit. Moreover, two transmission zeros appeared at the passband edges are clearly observed, thus much improving the selectivity of the proposed CMOS BPF. The proposed filter provides a feasible result to apply the fractal-based structure in the mm-wave application.

1 1 1 l0 ðac þ ad Þ pffiffiffiffiffiffi ¼ þ ¼ Q0 Qc Qd p ere

References

e

n ¼ 1 rn

n¼1

ð4Þ

where ac and ad are the attenuation coefficient corresponding to the conductor and dielectric losses, respectively, and Qc and Qd, are the quality factors corresponding to conductor and dielectric losses, respectively. In our previous study, the external quality factors of the fourth order Sierpinski square resonator realized on the low loss dielectric substrate achieve the saturation value about 210. Since the low-resistivity silicon has a very large substrate loss up to mm-wave frequencies, the external quality factors of the proposed filter using fourth order Sierpinski square resonator is 4. However, we still verified that the measured passband perform of our design using standard 0.35 mm CMOS process is in the acceptable range. The measured insertion loss about 7 dB is due to the close proximity of the ground plane to the

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