A new microstrip coupling system for realization of a differential dual-band bandpass filter

Accepted Manuscript Regular paper A new microstrip coupling system for realization of a differential dual-band bandpass filter Gholamreza Karimi, Mohsen Amirian, Ali Lalbakhsh, Mahnaz Ranjbar PII: DOI: Reference:

S1434-8411(18)32100-9 https://doi.org/10.1016/j.aeue.2018.11.004 AEUE 52570

To appear in:

International Journal of Electronics and Communications

Received Date: Revised Date: Accepted Date:

4 August 2018 5 October 2018 1 November 2018

Please cite this article as: G. Karimi, M. Amirian, A. Lalbakhsh, M. Ranjbar, A new microstrip coupling system for realization of a differential dual-band bandpass filter, International Journal of Electronics and Communications (2018), doi: https://doi.org/10.1016/j.aeue.2018.11.004

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Int. J. Electron. Commun. (AEÜ)

A new microstrip coupling system for realization of a differential dual-band bandpass filter Gholamreza Karimi a,, Mohsen Amiriana, Ali Lalbakhshb, and Mahnaz Ranjbar c a

Department of Electrical Engineering, Faculty of Engineering, Razi University, Kermanshah, Iran Department of Electrical Engineering, Kermanshah Branch, Islamic Azad University, Kermanshah, Iran c Young Researchers and Elite Club, Kermanshah Branch, Islamic Azad University, Kermanshah, Iran b

A RT I C L E I N F O

A BS T RA C T

Article history: Received Received in revised form Accepted Available online

In this paper, a new Differential Dual-Band Wide-Bandpass Filter (DDWBF) using a new coupling system is presented. The adjustable coupling system composed of Two Coupled UShaped Structures (TCUSs) which create and control two initial passbands. Four TCUSs are then placed in four ports and connected through a defected fractal-based low impedance transmission line. Third order of Sierpinski fractal is applied on the transmission line to improve the out of band rejection of the differential mode along with common mode suppression. An analytical analysis is also presented to calculate the transmission zeros and poles of the DDWBF which verifies the authenticity of the design procedure. A prototype of the filter has been fabricated and tested which shows a very good agreement with the predicted results. According to the measured results, two wide passbands with fractional bandwidths of 45% and 13.5% at central frequencies of 4.32 and 8.61 GHz, respectively, have been achieved. The differential mode return loss is higher than 15 dB in both bands and the common mode suppression level is better than 14.85 dB.

Keywords: Differential structure Dual band U-shape Coupled line

Sierpinski fractal

———  Corresponding author. E-mail addresses: [email protected] (G.Karimi), [email protected] (M.Amirian), [email protected] (A.Lalbakhsh), [email protected] (M.Ranjbar).

1. Introduction Differential microstrip filters are important components in the modern wireless communication systems because of their high immunity to the environmental noise. There is a broad range of applications for differential structures, such as wideband bandpass filters [1-4], power dividers [5], antennas [6, 7], and dual/multi-band and band-stop filters [8-14]. Embedded defected ground structure (DGS) [11, 12], slot line resonators [13], coupled feed lines [15] stepped-impedance band-stop filters [16, 17], and split ring resonators (SRRs) [18], are different methods for achieving tunable wideband and dual-band band-pass filters (DBPF). A class of differential BPF with a specialized Asymmetric Parallel-Coupled Line (APCL) was utilized to improve the controllability of this type of filter [8]. The APCL was responsible to create and control a notch band in the wideband response of the DBPF. In [9] asymmetrical couple lines were proposed in the structure of DDBPF to provide a balanced dual band response. A controllable DDBPF using stepped-impedance resonators were proposed in [10], where the size of the filter was large. DGS units were used to suppress the CM noise within two Differential Mode (DM) passbands [11, 12]; however, the CM between the two passbands is undesirable. Slot-line resonators were designed and used to independently control two passbands with an acceptable CM suppression level [13]. In [14] a compact band-stop filter for CM noise suppression in WiMAX and TDLTE devices was designed. A complicated structure was obtained using DGS layers which were separated by the FR4 dielectric substrate. In [15] to realize an adjustable passband, the coupled feed lines combined with T-shaped open stubs were employed. A stepped-impedance band-stop filter with extended upper passbands was introduced in [16]. Band-rejection characteristics were achieved in [17] by introducing folded resonators in the configuration of dual-band bandstop filter proposed. In [18] a systematical analysis for the couplings of the modified Complementary SRR (CSRRs) were introduce, where interdigital capacitors and CSRRs generated two passbands separately. In this paper, a high-performance differential dual bandpass filter is presented. The filter is composed of a new coupling system, explained in Section II, which provides two independently controllable passbands. This coupling system comprises of Two Coupled U-Shaped Structures (TCUS) which control the two passbands through unequal coupling paths. Sierpinski fractal geometry is also applied to the I-shaped transmission line to increase the out-of-band rejection in the DM and improve CM response. Final simulated and measured results are discussed in Section IV, where a performance comparison between the most recent DBPF is presented, verifying the competitiveness of the proposed filter. 2. Two coupled U-shaped structure 2.1. U-shaped and single band coupling structure

Fig. 1(a) illustrates a basic U-shaped structure to resonate at a specific frequency. The input admittance of the U-shaped resonator is calculated by tan (1) Yin  j 2 ZU where θ=βL is the electrical length of each of the U-shaped arms and ZU denotes the characteristic impedance of the resonator. According to Eq. (1), it is seen that the U-shaped resonator resonants at frequencies with θ=kπ. In order to have a wideband response, a strong coupling needs to be realized to create the initial wide passband [19, 20]. Fig. 1(b) shows the configuration of the proposed single band coupling system composed of TCUSs. The Insertion loss of the TCUS is yielded using the calculated transmission matrix by Eq. (2) (Ze and Zo are even and odd modes characteristic impedances calculated using [21]) and the imaginary of input admittance is calculated by Eq. (3) (Z0=50 Ω is the characteristic impedance of two ports).  Ze  Zo cos   Tcoupled   Z e  Z o  j 4 sin   Z Z e o 

1 Z e  Z o   Z e  Z o  cos 2    2 2( Z e  Z o ) sin   Ze  Zo  cos   Ze  Zo  2

j

2

    2 2 2   16Z 0 (cos 2   1)  Z e  Z o   Z e  Z o  cos 2  Im(Yin )  4 2 2  Z e  Z o     2 2   16Z 0 ( Z e  Z o ) sin  cos   2 csc 2  Z  Z   Z e  Z o cos    o  e   

(2)

(3)

Resonant frequencies can be calculated through Im(Yin)=0 to form the single passband. So, the electrical length and the resonant frequencies of TCUS are 

  cos 1    

f r (GHz) 

Z 0  Z e  Z o  2 2 Z 0  Z e  Z o  2

300 2L  eff

2

   

(4)

(5)

where ɛeff denotes the effective dielectric constant of the microstrip line and L is the length of TCUS in millimeters. According to Eq. (4), for every value of Ze and Zo, two values of θ (θ2=π-θ1) are yielded, leading to two resonant frequencies. For example, for Ze=222Ω and Zo=73Ω yielded for Fig. 1(b), θ1= 0.32π and θ2=0.68π which lead to fr1= 4.1, and fr2=6.5 GHz, respectively. Theoretical results are plotted in Fig. 1(c) along with the simulated results by ADS Momentum. As it can be seen from the insertion loss under weak coupling condition, TCUS introduces two controllable transmission poles to form the single passband.

2.2. Dual band coupling structure One of the significant characteristics of the proposed coupling system (TCUS) is passing the input signals through two independent coupling paths which can be used to create two independent passbands. To do so, U-shaped resonators’ arms of TCUS is bent to create two unequal coupling paths, resulting in a dual independently controllable passband. This structure is named unequal TCUS and depicted in Fig. 2. To verify the ability of the proposed coupling system in providing two independent passbands, the transmission matrix of unequal TCUS has been theoretically extracted (see Eq. (6a) and (6b)) and the input admittance of the structure is calculated by Eq. (7a)-(7e). According to the resonance condition, the imaginary part of Yin is equal to zero (BC-AD=0) and plotted in Fig. 3(a) for different values of L1. This analysis shows that L1 controls the second passband, by tuning the transmission poles (f3 and f4 in Fig. 3(a)), while it has almost no effects on the first band. The insertion loss of unequal TCUS under weak coupling condition is also simulated and shown in Fig. 3(b), verifying the authenticity of the calculations.

a

b

c

Fig. 1. (a) U-shaped resonator, (b) Two coupled U-shaped structure (W1=W2=g1=0.1 mm, L=9 mm, g=1.3 mm), (c) Insertion loss.

Fig. 2. The proposed dual band coupling system

a

b

c

d

Fig. 3. Tunability of the dual passband (a) Calculated susceptance of the unequal TCUS with different L1, when L2=9.2 mm. (b) Simulated result of S21 with weak coupling with different L1. (c) Calculated susceptance of the Unequal TCUS with different L2, when L1=10.1 mm. (d) Simulated result of S21 under weak coupling condition with different L2. Z Z2 A  2Z e  Z o ( sin 2 1  sin 2 1 cos 2  2 2 2 Z Z2 2 2 )  sin 2 2  Zsin 222cos cos 2  1 1 1  1 2 2

(7b) 2 2  cos 1 Z sin 2  sin 2  sin 21 sin 2 2   1 Z 2 cos 2   1  2 Z 2 cos 2  2 2 2 cos  2 cos  2 2 1  2 2 2 2 2 2 2 2 Z  j 2 B  200(1  Z cos 1  cos  2   2Z cos 1 cos2  2 2  2 2 2 2 2 sin 1 Z cos  2  1  sin  2 Z cos 1  1 sin 1 Z cos  2  1 (7c) sin  2 Z co  Z2 TU   2 sin 1 sin  2  sin 21 sin 2 2  2)  2 sin 21 2 sin 2 2 2 2 2 2 4 2 2  Z cos   1  Z  Z 1  Z c os   cos   Z cos  cos    2 1 2 1 C  100 2 Z  Z  Z 1  Z 2 sin 2  sin 2  2 2  j e o (7d) e o  1 Z 2 2   2 2 2 2 2 2  2 sin 1 Z cos  2  1  sin  2 Z cos 1  1 sin 1 Z cos  2  1  sin  2 Z  2 2 2 2 2 4 2 2 (7e)

 



 







 



























 

D  Z e  Z o  1  Z cos  2  Z cos 1  Z cos 1 cos  2



(6a) Likewise, the same analysis can be used to show the adjustability of the first passband by changing L2. As it is depicted in Fig. 3(c) and 3(d) increasing L2 cause the first band to shift to lower frequencies. Therefore, the independent tunability of both passbands is verified only by changing the length of L1and L2. Z

Ze  Zo Ze  Zo

Yin 

A  jB C  jD

(6b)

&

Im(Yin ) 

BC  AD C 2  D2

(7a)

Fig. 4. Initial layout of differential dual-band band-pass filter.



 



3. Differential dual wideband bandpass filter 3.1. Even and odd-modes equivalent networks Fig. 4 shows a configuration of a DDWBF. This is composed of four unequal TCUSs connected through an I-shaped transmission line to form a differential bandpass filter. Modal analysis is used to explain the filtering mechanism of the proposed structure. In the odd mode and under differential excitation, I-shaped resonator turns into a short-circuited Tshaped resonator as shown in Fig. 5(a). In the even mode when a common excitation is applied, the I-shaped transmission line plays the role of an open-circuited T-shaped resonator (Fig. 5(b)). The transmission matrix of the T-shaped resonator for both cases (short and open circuited) is calculated as

 Z1 sin 21 cot  2 cos 21  2Z 2 Tshort    sin 21 cos 2 1 cot 1 )  j( Z  Z2 1 

a

2  Z1 sin 2 1 cot  2 ) Z2  Z  cos 21  1 sin 21 cot  2  2Z 2 

j ( Z1 sin 21 cot  2 

b

c

(8)

 Z1 2 sin 21 tan 2  1  2 cos 1  2Z 2 Topen   2 sin 21 cos 1 tan1  j(  )  Z1 Z2 

2  Z1 sin 2 1 tan 2 )  Z2  Z   1  2 cos 2 1  1 sin 21 tan 2  2Z 2 

 j ( Z1 sin 21 

where Z1, Z2 are impedances and θ1, θ2 are electrical lengths of the T-shaped resonator. The resonant frequencies of the T-shaped resonator can be calculated using transmission matrix by setting Im(Yin)=0 for each mode. Thus, odd mode resonances, extracted from Eq. (8), occur at θodd1=π/2, θodd2=π/5, and θodd3=4π/5. Fig. 5(c) shows the calculated Im(Yin) and the insertion loss of odd mode under weak coupling condition to verify the accuracy of the analysis. Likewise, even mode resonant frequency is extracted from Eq. (9) and occurs at θeven=π/2, as shown in Fig. 5(d). The number of resonance frequencies forming the bandwidth is independent of Z1 and Z2, while the odd mode resonance locations are controlled by Z, where Z=Z1/Z2. The following figures show the variation of Im(Yin) for various Z. As seen from Fig. 6(a) increasing Z causes the odd mode resonances to become closer to each other’s (θ1 has shifted from 22.5° to 45° and θ3 shifted from 157.5° to 135° and θ2 remained unchanged). As shown in Fig. 6(b) changing Z has no effects on the even mode resonance frequency. In this paper, to achieve a dual passband response with the central frequencies of 4.32 and 8.61 GHz, Z1and Z2 were set to 49.5 and 55 respectively (i.e., Z=0.9). In order to visualize all transmission poles contributing to the dual-band response, the differential mode (DM) insertion loss of the filter in Fig. 4 is shown in Fig. 7. f1, f2, and fT1 are transmission poles in the first operating band, where f1 and f2 are generated and controlled by TCUS and fT1 is controlled by the Tshaped resonator. Likewise, f2, f4, and fT1 are transmission poles in the second operating band, where, f2, and f4 are independently controlled by TCUS and fT1 is controlled by the T-shaped resonator. TZ1 and TZ2 are transmission zeroes created by TCUSs. 3.2. Out of band improvement using sierpinski fractal

d

Fig. 5. T-shaped resonator. (a) Equivalent circuit in the odd mode. (b) Equivalent circuit in the even mode. (c) Calculated susceptance and simulation results in the odd mode mention in the revised (θ=θ2=2θ1). (d) Calculated susceptance and simulation results in the even mode (θ=θ2=2θ1).

(9)

Fig. 7. EM simulation of the initial differential dual band band-pass filter, shown in Fig. 4, under weak coupling condition (Z=144 Ω, Ze=222 Ω, Zo=73 Ω)

As it is shown in Fig. 7, a controllable dual passband response has been realized; however, the skirt performance is suboptimal. More precisely, the upper transition band of the second passband is undesirably wide. To tackle this issue, the deep transmission zero, located at 10 GHz in Fig. 7, needs to be shifted to a lower

a

frequency. This can be done by taking advantage of the spacefilling property of fractal geometries [22]. Several microwave filters have been modified and improved by different fractal geometries such as Koch, Minkowski, Hilbert, Peano, and Sierpinski [23-25]. Hence, Sierpinski fractal is applied on the Ishaped transmission line. In fractal geometry given any integer N, a segments of length L is the sum of N straight segment of lengths R=L/N, each of which can be obtained from the original segment by similarly of ratio R, with an appropriate focal point [26] (Fig. 8(a)). Fig. 8(b) shows the generation procedure of Sierpinski fractal up to 3rd iteration order as well as the effects of fractal geometry on differential and common modes of the insertion loss of the I-shaped transmission line.

As it can be seen in Fig. 8(c), the application of the fractal causes the transmission zero at 12.6 GHz to shift to lower frequencies as the iteration order increases. This causes the out of band transmission zero in Fig. 9(b) to relocate to 10.2 GHz in the 3th iteration, improving the upper sharpness of the filter. In terms of CM response improvement, Sierpinski fractal causes the deep transmission zero at 6.2 GHz in Fig. 8(d) shift to 5.4 GHz. This attenuates the undesired peak in the CM responses in Fig. 9(c). The fractal has also caused the 3 dB stopband bandwidth in CM

a b

c

b

a

d c

Fig. 8. Sierpinski fractal resonator. (a) Schematic illustration of Sierpinski fractal for iterations 0 to 3. (b) Generation procedure of Sierpinski fractal applied on the I-shaped transmission line (g1=0.25mm, g2=0.75mm, g3=2.25mm). (c) Differential magnitude mode response of the transmission lines for different iterations. (d) Common mode magnitude response of the transmission lines for different iterations.

b

Fig. 9. Proposed filter after the application of Sierpinski fractal; (a) layout L3=2.25mm, L4=3.9mm), (b) Fig. 6. (L1=9.6mm, The variation L2=5mm, of Im(Yin) for various Z. (a) Odd mode. (b) differential mode response for different iteration orders, (c) common Even mode. mode response for different iteration orders.

a

b

c

Fig. 10. (a) Equivalent circuits of the second order Sierpinski fractal resonator. (b) Equivalent circuits for DM excitation. (c) Equivalent circuits for CM excitation.

(W in Fig. 8(d)) increase from 5 GHz to 6.5 GHz, contributing to a further 10 dB attenuation in the CM, as depicted in Fig. 9(c). The layout of the final filter after applying Sierpinski fractal along with its CM and DM responses for different iteration orders are depicted in Fig. 9(a). The expected modification created by the fractal has been reflected in Fig. 9(b) and 9(c), where improved DM roll-off and CM suppression have been achieved. It should be noted that the application of Sierpinski fractal has not disturbed the adjustability of the passbands by TCUS. Lumped equivalent circuits of the second order Sierpinski fractal resonator are depicted in Fig. 10 for both differential and common-mode excitations. 4. Simulation and Result The filter has been fabricated on a substrate (Rogers_Ro 4003) with a relative dielectric constant, thickness and loss tangent of

3.38, 0.508 mm and 0.0022, respectively. Fig. 11(a) shows a photograph of the implemented filter. All simulations have been carried out by ADS Momentum (EM-simulator) and the fabricated prototype was measured by HP Agilent Keysight 8720B. As it can be seen from the measured and simulated results in Fig. 11(b), two flat passbands are realized, where the first band is from 3.35 to 5.30 GHz with a 3 dB FBW of 45% and the second band extends from 8.03 to 9.2 GHz with the 3 dB FBW of 13.5%. The differential mode bandwidths up to -2 dB level of the first and second bands are from 3.5 to 5.13 GHz (FBW of 35.5%) and 8.2 to 9.1 GHz (FBW of 10.4%), respectively. The return losses of the DM in the first and second passbands are better than 12 and 15 dB, respectively. The upper stopband is up to 14.1 GHz with an attenuation level better than 30 dB. The filter has four narrow transition bands with cut-off magnitudes of -3 and 20 dB. The first and second transition bands are from 2.85 to

Table. 1. Comparison with other dual-band filters. Ref. [8] [9] [10] [18] [25] This work

Differential structure

Electrical size (λg* λg)

Physical size (mm* mm)

Operating frequency (GHz)

DM bandwidth up to -2dB level (GHz)

Fractional bandwidth 3 dB (%)

Return loss

Insertion loss

Yes Yes Yes No No Yes

0.6*0.5 0.5*0.2 0.45*0.38 0.1*0.09 0.6*0.6 0.31*0.35

18.8*16.8 35.2*23.4 30*25 16.5*15 22.85*22.85 14*14.65

2.5-7.45 2.4-3.57 2.45-5.85 1.8-3.2 2.39-3.45 4.3-8.6

4.8 0.2-0.18 0.25-0.09 1.2-0.55 0.1-0.1 1.63-0.9

89 7.5-5 6.5-3 92.7-18.8 5.7-3.3 45-14

10.5 18-17 15 21-20 23 12-15

0.4-0.65 0.87-1.9 1.06-2.04 0.5-0.4 0.9-0.49 0.6-1.8

a

b

c

d

Fig. 11. (a) Photograph of the fabricated differential dual-band bandpass filter. Simulated and measured S-parameters for DDWBF in (b) D-mode (c) C-mode (d) Differential-to-common and common-to differential mode conversion.

3.35 GHz (0.5 GHz width) and 5.3 to 5.7 GHz (0.4 GHz width), respectively. The third and fourth transition bands extend from 7.8 to 8.03 GHz (0.23 GHz width) and 9.2 to 9.6 GHz (0.4 GHz width), respectively. The Measured roll-offs for the first and second passbands are 34, 42 and 77, 28, respectively. The insertion loss, including SMA loss, throughout the two passbands in the DM is better than 0.6 and 1.8 dB, respectively. Fig. 10(c) shows the results of S-parameter in common mode. Return loss over the two passbands for CM is better than 16 dB. Differentialto-common mode conversion plot is shown in Fig. 11(d). A performance comparison between the presented filter and some of the most recent published dual-band filters are tabulated in Table 1.

6.

5. Conclusion

11.

In this paper, an efficient differential wideband bandpass filter was proposed. The filter is composed of a new coupling system named Unequal Two Coupled U-shaped Structures (TCUS), which provides two independently controllable passbands. Four Unequal TCUSs along with a fractal-based transmission line form the differential dual passband filter. The bandwidths of two passbands are 1.95 and 1.17 GHz, respectively. The return loss of the proposed filter is better than 15 dB in the differential mode and the common-mode suppression level is higher than 16 dB, thanks to the application of fractal geometry. The filter has a high rejection level (30 dB) in the upper stopband up to 14.1 GHz and the insertion loss in the first and second bands is better than 0.6 and 1.8 dB, showing a stable passband response. Transmission matrix and modal analysis have been used to theoretically justify the filtering mechanism of the proposed structure. The filter is compact with the effective size of 14.0 mm×14.6 mm (0.31λg×0.35λg). Considering all the outstanding features of the presented dual-band filter; it is a good candidate to be used in microwave applications.

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