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Integrated design of filtering antenna with high selectivity and improved performance for L-band applications
Accepted Manuscript Regular paper Integrated design of filtering antenna with high selectivity and improved performance for L-band applications Bhagirath Sahu, Soni Singh, Manoj Kumar Meshram, Surya Pal Singh PII: DOI: Reference:
S1434-8411(18)31050-1 https://doi.org/10.1016/j.aeue.2018.10.015 AEUE 52542
To appear in:
International Journal of Electronics and Communications
Received Date: Revised Date: Accepted Date:
26 April 2018 11 October 2018 13 October 2018
Please cite this article as: B. Sahu, S. Singh, M.K. Meshram, S.P. Singh, Integrated design of filtering antenna with high selectivity and improved performance for L-band applications, International Journal of Electronics and Communications (2018), doi: https://doi.org/10.1016/j.aeue.2018.10.015
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Integrated design of filtering antenna with high selectivity and improved performance for L-band applications Bhagirath Sahu, Soni Singh, Manoj Kumar Meshram, *Surya Pal Singh Author 1: Bhagirath Sahu Affiliation: Indian Institute of Technology (Banaras Hindu University), Varanasi, India Degree: M.Tech. E-mail address: [email protected] Author 2: Soni Singh Affiliation: Indian Institute of Technology (Banaras Hindu University), Varanasi, India Degree: Ph.D. E-mail address: [email protected] Author 3: Manoj Kumar Meshram Affiliation: Indian Institute of Technology (Banaras Hindu University), Varanasi, India Degree: Ph.D. E-mail address: [email protected] Author 4: Surya Pal Singh Affiliation: Indian Institute of Technology (Banaras Hindu University), Varanasi, India Degree: Ph.D. E-mail address: [email protected] *Corresponding
author details:
Prof. Surya Pal Singh Institute Professor Department of Electronics Engineering Indian Institute of Technology (Banaras Hindu University), Varanasi Varanasi-221005, India E-mail address: [email protected] Mobile No.: +91–9415495333
Integrated design of filtering antenna with high selectivity and improved performance for L-band applications ABSTRACT An integrated design of compact filtering antenna having high selectivity and suppressed unwanted harmonics is proposed in this paper. The proposed filtering antenna is obtained by integrating a modified elliptic-shaped monopole antenna with a modified interdigital bandpass filter (IBPF). At first, a modified elliptic-shaped monopole antenna is designed and investigated through numerical simulation and experimentally. Further, the modified IBPF reported in the literature is integrated with the proposed monopole antenna to achieve good passband selectivity and suppressed unwanted harmonics for the proposed filtering antenna. The proposed filtering antenna is analysed through numerical simulation software. The simulated -10 dB reflection coefficient bandwidth of the proposed filtering antenna covers the frequency range 1.01 – 1.96 GHz with improved band-edge selectivity and unwanted harmonic suppression up to 8 GHz (= 5.4 f0, where f0 is the centre frequency of passband). It has nearly stable omnidirectional radiation patterns over the whole frequency band. The proposed filtering antenna was fabricated and experimentally tested. The experimental results are nearly in agreement with corresponding numerical simulation results. The proposed filtering antenna can be a suitable candidate for various L-band wireless communication applications. Keywords: Integrated design; filtering antenna; suppressed unwanted harmonics; high selectivity; L-band 1.
Introduction
The wireless communication system has been experiencing a revolutionary growth in the last few decades. This growth has been caused due to invention of many wireless products and
services including global positioning system (GPS), satellite communications, mobile communications, and amateur radio operations. In these systems, the size of both antennas and filters designed using first principles is comparable to the wavelength of signal. The components designed in this way would be of large physical size for the desired applications and hence need was felt to miniaturize the conventionally designed antennas. Keeping this aspect in view, various types of antennas [1–9] have been investigated. Among these antennas, planar monopole antennas [4–9] of different shapes viz rectangular, circular, and elliptical have been of particular research interest because of their compact size, simple structure, wide frequency impedance bandwidth, easy fabrication and omnidirectional radiation patterns. Also, these antennas have some other drawbacks, such as poor band-edge selectivity and presence of unwanted harmonics. Due to these drawbacks, chances of unwanted frequency components being received by the antennas can not be ruled out. This would affect the performance of systems using these antennas. Hence, the antenna is usually followed by a suitable bandpass filter (BPF) to improve its performance over the frequency band of operation and to suppress unwanted harmonics. To improve the signal-to-noise ratio (S/N) performance of a receiver used for aforesaid applications, it is imperative to place low noise amplifier (LNA), which includes BPF in its circuit, in close proximity with the antenna by reducing the length of transmission line connecting the antenna to LNA. That provided the researchers a purpose to integrate filter and antenna into a single compact component, known as filtering antenna that performs filtering and radiation functions simultaneously. Various types of filtering antennas are available in the literature [10–17]. In [11], integrated design of antenna and filter is reported which share same ground plane to achieve compact co-design of filter-antenna sub-system. The cascaded approach used to design antenna-filter systems is reported in [12–14]. In [15,16], a resonator of the filter structure is replaced by radiating element to obtain compact
filtering antenna having both filtering and radiation characteristics. A compact planar antenna integrated with lowpass filter (LPF) for suppression of upper harmonic band is reported in [17]. The reported filtering antennas required improvement in terms of size and/or filtering and radiation performance. In this paper, a compact L-band filtering antenna having sharp cut-off performance and suppressed unwanted harmonics is reported. The filtering antenna is obtained through integration of a modified elliptic-shaped monopole antenna with a modified interdigital bandpass filter (IBPF) reported in the literature. The BPF is responsible for obtaining improved cut-off performance in the desired frequency band with suppression of unwanted harmonics. In the initial phase, a modified elliptic-shaped monopole antenna was designed and analysed through numerical simulation and measurement. Further, the modified IBPF proposed by us and reported in the literature earlier [18] is integrated with the proposed antenna without any extra matching circuit to achieve compact integrated system with good impedance matching within the desired passband and highly suppressed out-of-band unwanted harmonics. The proposed integrated system will henceforth be referred to as filtering antenna. The proposed filtering antenna was studied through numerical simulation and measurement. The numerical simulation study is carried out using Ansys high-frequency structure simulator (HFSS) numerical simulation software based on finite element method (FEM) [19]. Roger RT/duroid 6010 substrate (εr = 10.2, tanδ = 0.0023, thickness = 1.27 mm) was used for design, simulation and experimental studies of the proposed monopole antenna and the proposed filtering antenna. To the best of the authors’ knowledge, the proposed filtering antenna is more compact in terms of electrical size with good cut-off performance and highly suppressed unwanted harmonics up to 8 GHz compared to filtering antennas reported in the literature.
2.
Proposed modified elliptic-shaped monopole antenna (MPA)
2.1.
Design and investigation of the proposed antenna
The schematic of the proposed modified elliptic-shaped MPA is shown in Fig. 1(a). The proposed MPA is evolved from a simple circular-shaped MPA. The design stages involved in the evolution of the proposed antenna starting from a simple circular-shaped MPA is shown in Fig. 2. Fig. 2 shows the geometries of five basic shapes of MPA along with the proposed modified elliptic-shaped MPA. Circular-shaped MPAs having partial rectangular ground plane and modified partial ground plane have been designated as ‘Type-I’ and ‘Type-II’ antennas respectively, whereas elliptic-shaped MPAs having partial rectangular ground plane and modified partial ground plane have been designated as ‘Type-III’ and ‘Type-IV’ antennas respectively. ‘Type-V’ antenna consists of a modified elliptic-shaped patch which is a combination of two modified elliptic-shaped patches and partial rectangular ground plane. Roger RT/duroid 6010 substrate material (εr = 10.2, tanδ = 0.0023, thickness = 1.27mm) is used for design of all MPAs along with the proposed antenna. The proposed antenna consists of a modified elliptic-shaped patch and modified ground plane fed through a 50 Ω microstrip line with tapered transition. The top side of the proposed antenna consists of a combination of two modified elliptic-shaped patches connected to 50 Ω microstrip feedline through a tapered transition. The modified elliptic-shaped patch is responsible for increasing the electrical length of the proposed antenna which shifts its resonant frequency to lower side without increasing its physical size, thereby making the antenna more compact. The bottom side of the proposed antenna contains modified ground plane having curved shape and a rectangular notch in the central region of its upper side. The rectangular notch helps in efficiently tuning the mutual coupling between the radiating element and the ground over a wide frequency range. The curved shape in the upper region of the ground plane is introduced for smooth transition from one resonant mode to another across wide bandwidth which ensures good
impedance matching over wide frequency range. The values of optimized geometrical parameters of the proposed antenna are given in Table 1. The overall size of the proposed antenna is 78.8 mm × 47 mm × 1.27 mm. The basic shapes of MPA along with the proposed antenna have been designed and numerically simulated in order to compare their reflection coefficient-frequency characteristics. For a fair comparison, physical size of all the antennas including the proposed antenna is kept same. Fig. 3 shows the simulated variations of reflection coefficient with frequency for six different geometries of antenna, which include ‘Type-I’, ‘Type-II’, ‘Type-III’, ‘Type-IV’, ‘Type-V’, and the proposed antennas. To clearly show the differences in characteristics of different types of MPAs, reflection coefficient versus frequency plots of these antennas are shown in Fig. 3 upto 4 GHz. From Fig. 3, it can easily be observed that the proposed MPA provides the widest -10 dB reflection coefficient bandwidth of 88.9% in the frequency range 1.0 – 2.6 GHz along with good impedance matching over the desired frequency range. Further, the proposed MPA is numerically simulated for its radiation patterns, realized gain-frequency characteristic, and total efficiency-frequency characteristic in addition to its input characteristic. Furthermore, the prototype of the proposed antenna was fabricated using printed circuit board (PCB) Technology for experimental study in order to validate the simulation results for antenna’s input characteristics, radiation patterns and realized gainfrequency characteristics. Fig. 1(b) shows the top and bottom views of the fabricated prototype of the proposed modified elliptic-shaped MPA. A sub-miniature version A (SMA) connector was used for feeding microwave power to the antenna through microstrip line as shown in Fig. 1(b).
Fig. 1. (a) Geometry of the proposed modified elliptic-shaped monopole antenna. (b) Top and bottom views of the fabricated prototype of the proposed modified elliptic-shaped monopole antenna.
Table 1. Optimized geometrical parameter values of the proposed antenna (all dimensions are in millimeter). la = 78.8 wa = 47 lf = 46 wf = 1.1 lc = 3.9 wc = 3.7 lt = 2.7 wt = 4 lg1 = 40.5 lg2 = 40.6 a1 = 32.6 a2 = 7 a3 = 32.5 b1 = 6.2 b2 = 17.4 b3 = 5.5
Fig. 2. Geometries of different monopole antennas: circular-shaped with partial rectangular ground plane (Type-I), circular-shaped with modified partial ground plane (Type-II), ellipticshaped with partial rectangular ground plane (Type-III), elliptic-shaped with modified partial ground plane (Type-IV), modified elliptic-shaped with partial rectangular ground plane (Type-V), and modified elliptic-shaped with modified partial ground plane (proposed antenna).
Fig. 3. Numerically simulated variations of reflection coefficient with frequency for different geometries (Type-I, Type-II, Type-III, Type-IV, Type-V and Proposed) of the monopole antennas. 2.2.
Results and discussion
2.2.1
Reflection coefficient-frequency characteristics
Fig. 4 shows numerically simulated and measured variations of reflection coefficient of the proposed modified elliptic-shaped monopole antenna with frequency. From Fig. 4, it can be observed that simulated -10 dB reflection coefficient bandwidth of the proposed antenna covers the frequency range 1.0 – 2.6 GHz. It can also be observed from Fig. 4 that around simulated frequency of 5.7 GHz, unwanted harmonics are present. Because of the presence of unwanted harmonics, signal interference occurs in a communication system. Also, it affects the S/N performance of the receiver circuitry of the communication system. Hence, in order to achieve better S/N performance of receiver circuitry in a wireless communication system, out-of-band harmonics should be suppressed. The magnitudes of reflection coefficient of the proposed antenna were measured using Anritsu vector network analyser (VNA) Master (Model: MS2038C). The measured -10 dB reflection coefficient bandwidth of the proposed antenna covers the frequency range 1.01 – 2.50 GHz. It can be seen from Fig. 4 that the simulated and measured results are in agreement with each other excepting minor variations over the frequency range of interest due to fabrication tolerances and measurement errors.
Fig. 4. Numerically simulated and measured variations of reflection coefficient of the proposed antenna with frequency.
Fig. 5. Numerically simulated and measured radiation patterns of the proposed antenna for three different frequencies of 1.1, 1.8, 2.5 GHz.
2.2.2
Radiation patterns and peak realized gain-frequency characteristics
The simulated and measured co- and cross-polarized radiation pattern plots of the proposed antenna in E- and H-planes for three different frequencies (1.1, 1.8, 2.5 GHz) are presented in Fig. 5. From Fig. 5, it can be observed that the proposed antenna exhibits good omnidirectional co-polar radiation patterns with low cross-polarization levels at the frequencies of interest. The cross-polarization levels in E-plane are found to be less as compared to the corresponding levels in H-plane at a given frequency. Fig. 6 shows the simulated and measured realized gain values of the proposed antenna with frequency. The simulated (measured) realized gain values vary in the range 2.2 – 3.5 dB (1.9 – 3.3 dB) over the operating frequency range 1.0 – 2.6 GHz (1.01 – 2.50 GHz).
Fig. 6. Numerically simulated and measured gain values of the proposed antenna as functions of frequency. 2.2.3
Simulated total efficiency-frequency characteristic
The total efficiency of an antenna includes the losses at the input terminals as well as the losses within the structure of the antenna. Fig. 7 shows the simulated variation of total efficiency of the proposed antenna with frequency. From Fig. 7, it can be observed that values of total efficiency vary in the range 86 – 93 % over the desired passband 1.0 – 2.6 GHz. In the undesired frequency range, the total efficiency initially decreases but starts increasing from 3.3 GHz and peaks at 5.7 GHz, where its value reaches to ~ 94 %.
Thereafter, it shows decreasing trend upto 7.2 GHz and finally more or less increasing trend in the frequency range 7.2 – 8.0 GHz.
Fig. 7. Simulated total efficiency of the proposed antenna as a function of frequency. Table 2. Comparison of the proposed monopole antenna with the antennas reported in the literature. Reference
Lower cut-off frequency, fl (GHz)
Physical size (mm × mm)
Electrical size (λl × λl)
[5]
2.69
50 × 42
0.448 × 0.376
[6]
1.02
110 × 124
0.374 × 0.42
[7]
1.05
80 × 74
0.28 × 0.259
[8]
2.8
55.9 × 33
0.52 × 0.308
Proposed antenna
1.01
78.8 × 47
0.265 × 0.16
Realized gain (dBi)
Total efficiency
~0 at 2.69 GHz, and 0.5 at 4 GHz 0.5 at 1.02 GHz, and 1.5 at 5 GHz
N. A.
N. A. 2.32 at 2.9 GHz, and 1.21 at 3.4 GHz 2.2 – 3.5
96 % at 1.02 GHz, Minimum 76 % N. A. Minimum 85 % 86 – 93 %
λl is the free space wavelength at the lower cut-off frequency (fl). N.A. stands for ‘not available’ 2.2.4
Performance comparison of the proposed antenna with those reported in literature
The dimension and performance based comparison of the proposed monopole antenna with those reported in the literature is provided in Table 2. It is apparent from Table 2 that overall electrical size of the proposed antenna (in terms of wavelength, λl) is smaller than the antennas reported in the literature. The proposed antenna provides almost stable radiation
patterns with reasonably good realized gain values over its operating frequency band. However, the proposed antenna needs improvement in terms of cut-off performance in desired frequency band and out-of-band (unwanted) harmonics. 3.
The modified interdigital bandpass filter (IBPF)
3.1
Design of modified IBPF
In order to achieve sharp cut-off performance and wide stopband with reasonably suppressed out-of-band unwanted harmonics for the proposed MPA, the compact modified IBPF reported in [18] by us is used. Certain design principles were considered while designing the modified IBPF for its integration with the proposed MPA in order to obtain an L-band filtering antenna for applications in L-band wireless communication. Firstly, the dielectric substrate employed for the BPF should be same as that used for the antenna. Secondly, the characteristic impedance of the microstrip line feeds at the input and output ports of the filter must be equal to 50 Ω i.e. the microstrip line width at the input and output ports of the filter should meet 50 Ω impedance criterion. Thirdly, the -10 dB reflection coefficient bandwidth of the filter should cover L-band frequency range in order to finally obtain the L-band filtering antenna after its integration with the proposed MPA. The dielectric substrate, RT/duroid 6010, having dielectric constant of 10.2 and thickness of 1.27 mm was used for design of the modified IBPF. Fig. 8(a) shows the perspective view of the BPF having spurlines and defected ground structures (DGSs). Fig. 8(b) and 8(c) show the top and bottom views of the BPF. The band notched characteristic of spurline, and harmonic suppression along with miniaturization characteristic of DGS is responsible for compactness along with wide stopband characteristics of the reported BPF. For the sake of brevity, detailed design procedure, which is available in [18] is not explained here. The geometrical parameter values of the BPF for L-band operation are given in Table 3. The size of the compact BPF is 20.2 mm × 9.0 mm × 1.27 mm.
3.2
Results and discussion
The simulated variations of the magnitude of S-parameters of the BPF with frequency are shown in Fig. 9. From Fig. 9, it can be observed that -10 dB reflection coefficient bandwidth of the BPF is 1.05 GHz (= 71.6 %) and covers the frequency range 0.94 – 1.99 GHz. The 3dB passband frequency range of the BPF is 0.89 – 2.05 GHz. As far as stopband performance is concerned, the upper stopband attenuation level is better than 17 dB up to 8 GHz. The maximum value of simulated insertion loss at a mid-band frequency of the filter passband is found to be 0.35 dB.
Fig. 8. Geometry of the BPF (a) Perspective view, (b) Top view, and (c) Bottom view. Table 3. Geometrical parameter values of the BPF (all dimensions are in millimetre) l1 = 19.3 w1 = 1.8 l2 = 17.3 w2 = 1
l3 = 17.2 w3 = 2.4 g12 = 0.2
g23 = 0.3 d1 = 0.4
d2 = 0.8
l4 = 1.6
l5 = 2.8
l6 = 2.7
w4 = 0.9
l7 = 0.8
l8 = 2.6
l9 = 0.3
g2 = 0.4
w5 = 0.4 l10 = 3.7
l11 = 0.9
l12 = 2.5 l13 = 1.3
g1 = 0.7
w6 = 0.2 l14 = 8.4
4.
Proposed filtering antenna
4.1
Design and investigation of the proposed filtering antenna
In order to obtain an integrated system providing appropriate filtering and radiation characteristics for L-band applications, the L-band compact modified IBPF is integrated with modified elliptic-shaped MPA. The L-band integrated system is called as L-band filtering
antenna. The L-band integrated system/component is designed due to various advantages and applications of L-band frequency range (1 – 2 GHz). The GPS signals are used in L-band of frequency spectrum because L-band wave can penetrate clouds, fog, rain, storm and vegetation, and it requires less expensive hardware and smaller size components/system. Lband is also used for satellite communication, mobile communication, and aircraft surveillance.
Fig. 9. Numerically simulated variations of the magnitude of S-parameters of the BPF versus frequency [18]. For design of the integrated system, certain design principles were considered during optimization of its geometrical parameters. Firstly, the individual components should be designed on the same dielectric substrate i.e. RT/duroid 6010 substrate (εr = 10.2, tanδ = 0.0023, and thickness = 1.27 mm) in the present case. Secondly, the characteristic impedance of the microstrip line feeds at the input and output ports of the filter must be equal to 50 Ω in order to meet the 50 Ω impedance criterion at the input port of the antenna. Thirdly, the -10 dB reflection coefficient bandwidth of the filter should cover L-band frequency range i.e. 1 – 2 GHz in order to finally obtain the L-band filtering antenna after its integration with the proposed MPA. The geometrical parameters of the connecting transmission lines of the antenna and the filter are optimised through numerical simulation in order to match their input impedance with the common reference impedance of 50 Ω. Fig. 10 (a) shows geometry
of the proposed filtering antenna obtained through integration of modified elliptic-shaped monopole antenna with the modified IBPF reported in [18]. The geometrical parameters of the integrated system are optimized to achieve desired performance in L-band. The optimized geometrical parameter values of the proposed integrated design of filtering antenna are given in Table 4. The prototype of the proposed filtering antenna was fabricated on RT/duroid 6010 substrate (εr = 10.2, tanδ = 0.0023, and thickness = 1.27 mm) using PCB technology for experimental study. Fig. 10 (b) shows top and bottom views of the fabricated prototype of the proposed integrated filtering antenna. The overall physical size of the proposed filtering antenna is 91.8 mm × 47 mm × 1.27 mm.
Fig. 10. (a) Geometry of the proposed integrated design of filtering antenna. (b) Top and bottom views of the fabricated prototype of the proposed filtering antenna. Fig. 11 shows the simulated variations of reflection coefficient with frequency for the proposed modified elliptic-shaped monopole antenna with and without integration of the BPF. From Fig. 11, it can be observed that the proposed integrated design (also called the proposed filtering antenna) provides sharp band-edge selectivity at lower and upper edges of
frequency band (1.01 – 1.96 GHz) along with good suppression of unwanted harmonics up to 8 GHz.
Table 4. Optimized geometrical parameter values of the proposed integrated design of filtering antenna (all dimensions are in millimetre) L = 91.8
W = 47
lf 1 = 3.5
lc = 3.9
wc = 3.7
lt = 2.7
b2 = 17.4
b3 = 5.5
g12 = 0.2
g23 = 0.3 d1 = 0.4
w4 = 0.9
l7 = 0.8
l12 = 2.5
l13 = 1.3
wf 1 = 1.7 lf 2 = 43 wt = 4
a1 = 32.6
l1 = 19.3 w1 = 1.8
l2 = 17.3
d2 = 0.8
l8 = 2.6 w6 = 0.2
wf 2 = 1.4 lg3 = 51 a2 = 7
lg4 = 51.1
a3 = 32.5
b1 = 6.2
w2 = 1
l3 = 17.2
w3 = 2.4
l4 = 1.6
l5 = 2.8
l6 = 2.7
g1 = 0.7
l9 = 0.3
g2 = 0.4
w5 = 0.4
l10 = 3.7
l11 = 0.9
l14 = 8.4
–
–
–
–
Fig. 11. Numerically simulated variations of reflection coefficient of the proposed modified elliptic-shaped monopole antenna with and without integration of the BPF versus frequency. 4.2.
Equivalent circuit model of the proposed filtering antenna
The equivalent circuit model of the proposed filtering antenna can be helpful in the computer aided design (CAD) of the RF front-end of a transceiver [20]. Fig. 12 shows the equivalent circuit model of the proposed filtering antenna which is based on a heuristic modification of the Foster's network synthesis technique [21, 22]. The resistive elements are introduced to model metal, dielectric and radiation losses.
Fig. 12. Equivalent circuit model of the proposed filtering antenna.
The input impedance of the antenna is modelled by means of the equivalent network depicted in Fig. 12. In order to describe the antenna frequency behaviour, following series expansion is adopted to describe the antenna frequency behaviour [22]
1 Zin ( ) j L0 jC0
where Qn n RnCn and n LnCn
1/2
N max
Rn n 1 jQ n 1 n n
(1)
. In Equation (1), Nmax is the number of modes
needed to properly describe the frequency behaviour of the antenna input impedance, ω is the operating radian frequency, ωn is the radian frequency of the n-th resonant mode, C0 is the quasi-static input capacitance, L0 is an inductance that takes into account higher order modes as well as probe feed effects, while Cn, Ln, Rn, and Qn are the capacitance, inductance, resistance, and quality factor respectively, describing the damped resonance processes that take place in the antenna structure. From the analysis of the circuit, C0 can be found as C0 lim 0
Im{YinEM ( )}
Im{YinEM ( )} Im{YinEM ( )} lim 0
(2)
Starting from C0, L0 can be computed as L0 1/ {[0 ]2 C0 } where ω0 is the lowest frequency for which Im{ZinEM (0 )} 0 . The values of Rn and Qn can be estimated using Equations (3) and (4) respectively. Rn Re{ZinEM (n )},
(3)
and
Qn
Im{ZinEM (n )} Im{ZinEM (n )} n Im{ZinEM (n )} n 2 Rn 4 Rn
(4)
Now, parameter values of the circuit model can be evaluated using Equations (1) - (4). As shown in Fig. 14, approximately seven resonant modes are necessary to represent the antenna frequency behaviour in L-band. The equivalent circuit parameter values are given in Table 5.
Table 5. Parameter values of the equivalent circuit model for the proposed filtering antenna. C0 = 5.25 pF
L0 = 0.02 μH
R1 = 46.353 Ω
L1 = 1.431 nH
C1 = 19.5 pF
R2 = 41.646 Ω
L2 = 1.64 nH
C2 = 16.46 pF
R3 = 47.347 Ω
L3 = 1.383 nH
C3 = 15.09 pF
R4 = 50.278 Ω
L4 = 1.457 nH
C4 = 11.33 pF
R5 = 76.614 Ω
L5 = 0.641 nH
C5 = 19.23 pF
R6 = 35.893 Ω
L6 = 1.105 nH
C6 = 7.01 pF
R7 = 77.435 Ω
L7 = 0.468 nH
C7 = 15.76 pF
4.3.
Surface current distribution
For better understanding of integration effectiveness of the BPF with the proposed modified elliptic-shaped antenna for suppression of unwanted harmonics, the current distributions on the surface of the proposed antenna with and without the BPF are studied. Fig. 13 depicts the current distributions on the surface of the proposed antenna with and without the BPF at a frequency of 5.7 GHz (a simulated frequency for unwanted harmonic without BPF). It can be noticed from Fig. 13(a) that higher level of current reaches the modified elliptic-shaped radiating patch from the input port in the absence of BPF. Therefore, in order to suppress the harmonics, the BPF was used to block the passage of current from input port to radiating modified elliptic-shaped antenna. It can be clearly observed from Fig. 13 (b) that in presence of the BPF, the current is more concentrated near the BPF, and negligible current reaches the modified elliptic-shaped antenna.
Fig. 13. Simulated surface current distributions of the proposed antenna at a frequency of 5.7 GHz (an undesired response frequency without BPF) (a) without BPF, and (b) with BPF.
Fig. 14. Numerically simulated and measured variations of reflection coefficient of the proposed filtering antenna versus frequency. 4.4.
Results and discussion
4.4.1. Reflection coefficient-frequency characteristics The numerically simulated and measured variations of the reflection coefficient of the proposed filtering antenna with frequency are shown in Fig. 14. The values of reflection coefficient of the proposed filtering antenna were measured at different frequencies using Anritsu vector network analyser (VNA) Master (Model: MS2038C). The simulated (measured) -10 dB reflection coefficient bandwidth of the proposed filtering antenna covers the frequency range 1.01 – 1.96 GHz (1.02 – 1.92 GHz). The simulated and measured results are in agreement with each other excepting minor variations occurring over the frequency band of interest due to fabrication tolerances and measurement errors. 4.4.2. Radiation patterns The simulated and measured co- and cross-polarized radiation pattern plots of the proposed filtering antenna in E- and H-planes for two different frequencies (1.1, 1.8 GHz) which are
within operating frequency band are presented in Fig. 15. From Fig. 15, it can be observed that the proposed filtering antenna has good omni-directional co-polar radiation patterns with low cross-polarization levels at the frequencies considered in the study. The proposed filtering antenna exhibits similar radiation patterns with tolerable cross-polarization levels when compared with the proposed antenna without BPF presented in Fig. 5 for same frequency. It shows that the radiation characteristics of the proposed filtering antenna are more or less maintained with good filtering characteristics over the frequency band of interest.
Fig. 15. Numerically simulated and measured radiation patterns of the proposed filtering antenna for two different frequencies. 4.4.3. Peak realized gain-frequency characteristics Fig. 16 shows the variations of simulated and measured realized gain values of the proposed antenna with and without integration of the BPF with frequency. From Fig. 16, it can be observed that the simulated and measured realized gain-frequency characteristics of the
proposed filtering antenna are nearly in agreement with each other. The simulated (measured) realized gain values of the filtering antenna vary in the range 1.99 – 2.96 dB (1.82 – 2.71 dB) over the operating frequency band 1.01 – 1.96 GHz (1.02 – 1.92 GHz). It can also be seen from Fig. 16 that drastic reduction in gain values occurs at lower and upper edges of the frequency band. Though the simulated realized gain values vary in the range 2.1 – 3.5 dB over the passband of isolated antenna without BPF similar to those for the proposed filtering antenna, change in its gain value beyond the upper edge of frequency band is not large. The proposed filtering antenna provides simulated lowest realized gain value of -18 dB at 2.1 GHz beyond the upper passband edge. In case of the isolated antenna without BPF, simulated realized gain value at 2.1 GHz is 3.1 dB, which is 21.1 dB higher than that for the proposed filtering antenna. Further, simulated realized gain value of -18 dB is obtained at 0.85 GHz below lower passband edge of the proposed filtering antenna. The simulated realized gain value for the isolated antenna without BPF at 0.85 GHz is -4 dB, which is 14 dB higher than that for the proposed filtering antenna. This shows the performance superiority of the proposed filtering antenna over the isolated antenna without BPF in terms of realized gain parameter.
Fig. 16. Numerically simulated and measured variations of realized gain values of the proposed antenna with and without BPF versus frequency.
4.4.4. Simulated total efficiency-frequency characteristic Fig. 17 shows the variations of simulated total efficiency of the proposed antenna with and without integration of the BPF with frequency. It can be observed from Fig. 17 that the simulated total efficiency of the proposed filtering antenna vary in the range 85 – 89 % within its operating frequency band but drastic reduction in total efficiency occurs on either side of passband (= 1.01 – 1.96 GHz). The simulated total efficiency values vary in the range 86 – 93 % over the passband of isolated antenna without BPF similar to those for the proposed filtering antenna. It can be stated after comparing the total efficiency-frequency characteristics of the proposed antenna with and without integration of the BPF that significant reduction in the total efficiency value occurs above the desired passband in case of the proposed filtering antenna i.e. the proposed monopole antenna integrated with the proposed modified IBPF as compared with the isolated antenna without integration of BPF. This shows the performance superiority of the proposed filtering antenna over the isolated antenna without BPF in terms of total efficiency parameter.
Fig. 17. Numerically simulated variations of total efficiency of the proposed antenna with and without integration of the proposed BPF versus frequency. 4.4.5. Performance comparison of the proposed filtering antenna with those reported in literature Finally, performance and dimension based comparison of the proposed filtering antenna with those reported in the literature is given in Table 6. It is apparent from Table 6 that overall
electrical size of the proposed filtering antenna (in terms of wavelength, λl) is smaller as compared with the filtering antennas reported in the literature. The proposed filtering antenna also provides good harmonic suppression with improved selectivity. Table 6. Performance comparison of the proposed filtering antenna with filtering antennas reported in the literature.
Reference
[11] [14] [15] [16] [17]
Measured -10 dB reflection coefficient bandwidth (FBW) 4.06 – 4.26 GHz (4.7%) 2.32 – 2.55 GHz (9.4%) 2.26 – 2.66 (16.3%) 1.99 – 2.07 GHz (3.5%) 2.5 – 3.3 GHz (27.5 %)
Measured harmonic suppression Not available Upto 3.5 GHz Upto 3.5 GHz Not available Upto 14 GHz
Physical Size (mm × mm) 20 × 40 70 × 75 30 × 45 85 × 80 53.3 × 34.2
Electrical Size (λl × λl) 0.27 × 0.54 0.54 × 0.58 0.226 × 0.339 0.56 × 0.53 0.44 × 0.285
Realized Total gain efficiency (dBi)
1.4 – 4.3
N. A.
3.3
N. A.
2.41
N. A.
5
N. A.
2.5
N. A.
Proposed 1.99 – 85 – 89 1.02 – 1.92 0.31 × 2.96 % filtering Upto 8 GHz 91.8 × 47 GHz (61.2%) 0.16 antenna λl is the free space wavelength at the lower cut-off frequency (fl). FBW stands for ‘fractional bandwidth’. N.A. stands for ‘not available’.
5.
Conclusion
A compact filtering antenna having sharp cut-off performance and highly suppressed unwanted harmonics has been described. The proposed filtering antenna has been obtained through integration of the modified elliptic-shaped monopole antenna with the modified IBPF reported in the literature. The proposed filtering antenna is compact in size with good bandedge selectivity and suppressed out-of-band harmonics up to 8 GHz. The proposed filtering antenna and the antenna without optimized modified IBPF have been investigated through numerical simulation and measurements and the results have been found to be nearly in agreement with each other. It is finally concluded that the proposed compact filtering antenna
can be a good candidate for various L-band wireless communication applications, such as satellite communication, global positioning system, and aircraft surveillance system.
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Sun GH, Wong SW, Zhu L, Chu QX. A compact printed filtering antenna with good suppression of upper harmonic band. IEEE Antenn Wireless Propag Lett 2016;15:134952.
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Sahu B, Tripathi P, Singh S, Meshram MK, Singh SP. Investigation on compact modified wideband interdigital bandpass filter with wide stopband using spurlines and defected ground structures. Microw Opt Technol Lett 2016;58(11):2634-9.
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Bhagirath Sahu received the B.E. degree in Electronics and Communication Engineering from R.G.P.V., Bhopal, India, in 2010, and the M.Tech degree in Electronics Engineering (Microwave Engineering) from IIT (BHU), Varanasi, India, in 2013. He is currently working towards the Ph.D. degree in Electronics Engineering (Microwave Engineering) from IIT (BHU), Varanasi, India. He is author or co-author of about twelve Journal papers and twenty international/national conference papers. His research interests include microstrip filters, filtering antennas, dielectric resonator antennas, antennas for wireless/biomedical applications and metamaterials. Mr. Sahu is a student member of the Institute of Electrical and Electronics Engineers (IEEE) Microwave Theory and Techniques Society (MTT-S). Soni Singh was born in Deoria, UP, India in 1989. She received the B.Sc. degree in Electronics from DDU Gorakhpur University, India in 2009 and M. Sc. degree in Electronics from Banasthali University, Rajasthan, India in 2011. She completed her Ph.D. work in 2016 from the Department of Electronics Engineering, Indian Institute of Technology (Banaras Hindu University), Varanasi, India. She is author or co-author of twelve journal papers and twenty international/national conference papers. Her major research activity includes dielectric loaded metal horn antenna, medical applications of microwaves, and antenna technology. Manoj Kumar Meshram received the Ph.D. degree from the Department of Electronics Engineering, IT, BHU (now IIT (BHU)), Varanasi, India, in the year 2001. In August 2001, he joined as Assistant Professor in the Department of Electronics and Communication Engineering at DIT, Dehradun, India. In July 2002, he joined as a Lecturer in the Department of Electronics Engineering, IT, BHU, Varanasi, India, where presently, he is working as Associate Professor. His research interests include microwave and millimeter wave technology, meta-material, reconfigurable antennas, frequency scanning antennas for radar system and microwave imaging. He has published about hundred papers in International Journals. Dr. Meshram is a Senior Member of the IEEE, member of Indian Science Congress Association, India, Life Member of Indian Society of Technical Education (ISTE), India, and a Life Member of Institution of Electronics and Telecommunication Engineers (IETE), India. S. P. Singh (corresponding author) received the Ph.D. degree in Electronics Engineering (Microwave Engineering) from BHU, Varanasi, India, in 1989. He had been employed at Department of Electronics Engineering, IT, BHU (now IIT (BHU)), Varanasi since 1980 and retired as Professor on January 31, 2016. Presently, he is re-employed as Institute Professor in the same institute. He has been Coordinator of the UGC funded CAS program during 2005–2011, and Head of the Department during 2010–2013. He has supervised/co-supervised fourteen Ph.D. theses and sixty five M.Tech. Dissertation projects. He has authored/coauthored over two hundred research papers in international/national journals and conference proceedings. His areas of current research and publications include bio-electromagnetics, microwave antennas and microwave measurements. Prof. Singh is a Life Senior Member of IEEE, USA and a Life Fellow of IETE, India.
S1434-8411(18)31050-1 https://doi.org/10.1016/j.aeue.2018.10.015 AEUE 52542
To appear in:
International Journal of Electronics and Communications
Received Date: Revised Date: Accepted Date:
26 April 2018 11 October 2018 13 October 2018
Please cite this article as: B. Sahu, S. Singh, M.K. Meshram, S.P. Singh, Integrated design of filtering antenna with high selectivity and improved performance for L-band applications, International Journal of Electronics and Communications (2018), doi: https://doi.org/10.1016/j.aeue.2018.10.015
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Integrated design of filtering antenna with high selectivity and improved performance for L-band applications Bhagirath Sahu, Soni Singh, Manoj Kumar Meshram, *Surya Pal Singh Author 1: Bhagirath Sahu Affiliation: Indian Institute of Technology (Banaras Hindu University), Varanasi, India Degree: M.Tech. E-mail address: [email protected] Author 2: Soni Singh Affiliation: Indian Institute of Technology (Banaras Hindu University), Varanasi, India Degree: Ph.D. E-mail address: [email protected] Author 3: Manoj Kumar Meshram Affiliation: Indian Institute of Technology (Banaras Hindu University), Varanasi, India Degree: Ph.D. E-mail address: [email protected] Author 4: Surya Pal Singh Affiliation: Indian Institute of Technology (Banaras Hindu University), Varanasi, India Degree: Ph.D. E-mail address: [email protected] *Corresponding
author details:
Prof. Surya Pal Singh Institute Professor Department of Electronics Engineering Indian Institute of Technology (Banaras Hindu University), Varanasi Varanasi-221005, India E-mail address: [email protected] Mobile No.: +91–9415495333
Integrated design of filtering antenna with high selectivity and improved performance for L-band applications ABSTRACT An integrated design of compact filtering antenna having high selectivity and suppressed unwanted harmonics is proposed in this paper. The proposed filtering antenna is obtained by integrating a modified elliptic-shaped monopole antenna with a modified interdigital bandpass filter (IBPF). At first, a modified elliptic-shaped monopole antenna is designed and investigated through numerical simulation and experimentally. Further, the modified IBPF reported in the literature is integrated with the proposed monopole antenna to achieve good passband selectivity and suppressed unwanted harmonics for the proposed filtering antenna. The proposed filtering antenna is analysed through numerical simulation software. The simulated -10 dB reflection coefficient bandwidth of the proposed filtering antenna covers the frequency range 1.01 – 1.96 GHz with improved band-edge selectivity and unwanted harmonic suppression up to 8 GHz (= 5.4 f0, where f0 is the centre frequency of passband). It has nearly stable omnidirectional radiation patterns over the whole frequency band. The proposed filtering antenna was fabricated and experimentally tested. The experimental results are nearly in agreement with corresponding numerical simulation results. The proposed filtering antenna can be a suitable candidate for various L-band wireless communication applications. Keywords: Integrated design; filtering antenna; suppressed unwanted harmonics; high selectivity; L-band 1.
Introduction
The wireless communication system has been experiencing a revolutionary growth in the last few decades. This growth has been caused due to invention of many wireless products and
services including global positioning system (GPS), satellite communications, mobile communications, and amateur radio operations. In these systems, the size of both antennas and filters designed using first principles is comparable to the wavelength of signal. The components designed in this way would be of large physical size for the desired applications and hence need was felt to miniaturize the conventionally designed antennas. Keeping this aspect in view, various types of antennas [1–9] have been investigated. Among these antennas, planar monopole antennas [4–9] of different shapes viz rectangular, circular, and elliptical have been of particular research interest because of their compact size, simple structure, wide frequency impedance bandwidth, easy fabrication and omnidirectional radiation patterns. Also, these antennas have some other drawbacks, such as poor band-edge selectivity and presence of unwanted harmonics. Due to these drawbacks, chances of unwanted frequency components being received by the antennas can not be ruled out. This would affect the performance of systems using these antennas. Hence, the antenna is usually followed by a suitable bandpass filter (BPF) to improve its performance over the frequency band of operation and to suppress unwanted harmonics. To improve the signal-to-noise ratio (S/N) performance of a receiver used for aforesaid applications, it is imperative to place low noise amplifier (LNA), which includes BPF in its circuit, in close proximity with the antenna by reducing the length of transmission line connecting the antenna to LNA. That provided the researchers a purpose to integrate filter and antenna into a single compact component, known as filtering antenna that performs filtering and radiation functions simultaneously. Various types of filtering antennas are available in the literature [10–17]. In [11], integrated design of antenna and filter is reported which share same ground plane to achieve compact co-design of filter-antenna sub-system. The cascaded approach used to design antenna-filter systems is reported in [12–14]. In [15,16], a resonator of the filter structure is replaced by radiating element to obtain compact
filtering antenna having both filtering and radiation characteristics. A compact planar antenna integrated with lowpass filter (LPF) for suppression of upper harmonic band is reported in [17]. The reported filtering antennas required improvement in terms of size and/or filtering and radiation performance. In this paper, a compact L-band filtering antenna having sharp cut-off performance and suppressed unwanted harmonics is reported. The filtering antenna is obtained through integration of a modified elliptic-shaped monopole antenna with a modified interdigital bandpass filter (IBPF) reported in the literature. The BPF is responsible for obtaining improved cut-off performance in the desired frequency band with suppression of unwanted harmonics. In the initial phase, a modified elliptic-shaped monopole antenna was designed and analysed through numerical simulation and measurement. Further, the modified IBPF proposed by us and reported in the literature earlier [18] is integrated with the proposed antenna without any extra matching circuit to achieve compact integrated system with good impedance matching within the desired passband and highly suppressed out-of-band unwanted harmonics. The proposed integrated system will henceforth be referred to as filtering antenna. The proposed filtering antenna was studied through numerical simulation and measurement. The numerical simulation study is carried out using Ansys high-frequency structure simulator (HFSS) numerical simulation software based on finite element method (FEM) [19]. Roger RT/duroid 6010 substrate (εr = 10.2, tanδ = 0.0023, thickness = 1.27 mm) was used for design, simulation and experimental studies of the proposed monopole antenna and the proposed filtering antenna. To the best of the authors’ knowledge, the proposed filtering antenna is more compact in terms of electrical size with good cut-off performance and highly suppressed unwanted harmonics up to 8 GHz compared to filtering antennas reported in the literature.
2.
Proposed modified elliptic-shaped monopole antenna (MPA)
2.1.
Design and investigation of the proposed antenna
The schematic of the proposed modified elliptic-shaped MPA is shown in Fig. 1(a). The proposed MPA is evolved from a simple circular-shaped MPA. The design stages involved in the evolution of the proposed antenna starting from a simple circular-shaped MPA is shown in Fig. 2. Fig. 2 shows the geometries of five basic shapes of MPA along with the proposed modified elliptic-shaped MPA. Circular-shaped MPAs having partial rectangular ground plane and modified partial ground plane have been designated as ‘Type-I’ and ‘Type-II’ antennas respectively, whereas elliptic-shaped MPAs having partial rectangular ground plane and modified partial ground plane have been designated as ‘Type-III’ and ‘Type-IV’ antennas respectively. ‘Type-V’ antenna consists of a modified elliptic-shaped patch which is a combination of two modified elliptic-shaped patches and partial rectangular ground plane. Roger RT/duroid 6010 substrate material (εr = 10.2, tanδ = 0.0023, thickness = 1.27mm) is used for design of all MPAs along with the proposed antenna. The proposed antenna consists of a modified elliptic-shaped patch and modified ground plane fed through a 50 Ω microstrip line with tapered transition. The top side of the proposed antenna consists of a combination of two modified elliptic-shaped patches connected to 50 Ω microstrip feedline through a tapered transition. The modified elliptic-shaped patch is responsible for increasing the electrical length of the proposed antenna which shifts its resonant frequency to lower side without increasing its physical size, thereby making the antenna more compact. The bottom side of the proposed antenna contains modified ground plane having curved shape and a rectangular notch in the central region of its upper side. The rectangular notch helps in efficiently tuning the mutual coupling between the radiating element and the ground over a wide frequency range. The curved shape in the upper region of the ground plane is introduced for smooth transition from one resonant mode to another across wide bandwidth which ensures good
impedance matching over wide frequency range. The values of optimized geometrical parameters of the proposed antenna are given in Table 1. The overall size of the proposed antenna is 78.8 mm × 47 mm × 1.27 mm. The basic shapes of MPA along with the proposed antenna have been designed and numerically simulated in order to compare their reflection coefficient-frequency characteristics. For a fair comparison, physical size of all the antennas including the proposed antenna is kept same. Fig. 3 shows the simulated variations of reflection coefficient with frequency for six different geometries of antenna, which include ‘Type-I’, ‘Type-II’, ‘Type-III’, ‘Type-IV’, ‘Type-V’, and the proposed antennas. To clearly show the differences in characteristics of different types of MPAs, reflection coefficient versus frequency plots of these antennas are shown in Fig. 3 upto 4 GHz. From Fig. 3, it can easily be observed that the proposed MPA provides the widest -10 dB reflection coefficient bandwidth of 88.9% in the frequency range 1.0 – 2.6 GHz along with good impedance matching over the desired frequency range. Further, the proposed MPA is numerically simulated for its radiation patterns, realized gain-frequency characteristic, and total efficiency-frequency characteristic in addition to its input characteristic. Furthermore, the prototype of the proposed antenna was fabricated using printed circuit board (PCB) Technology for experimental study in order to validate the simulation results for antenna’s input characteristics, radiation patterns and realized gainfrequency characteristics. Fig. 1(b) shows the top and bottom views of the fabricated prototype of the proposed modified elliptic-shaped MPA. A sub-miniature version A (SMA) connector was used for feeding microwave power to the antenna through microstrip line as shown in Fig. 1(b).
Fig. 1. (a) Geometry of the proposed modified elliptic-shaped monopole antenna. (b) Top and bottom views of the fabricated prototype of the proposed modified elliptic-shaped monopole antenna.
Table 1. Optimized geometrical parameter values of the proposed antenna (all dimensions are in millimeter). la = 78.8 wa = 47 lf = 46 wf = 1.1 lc = 3.9 wc = 3.7 lt = 2.7 wt = 4 lg1 = 40.5 lg2 = 40.6 a1 = 32.6 a2 = 7 a3 = 32.5 b1 = 6.2 b2 = 17.4 b3 = 5.5
Fig. 2. Geometries of different monopole antennas: circular-shaped with partial rectangular ground plane (Type-I), circular-shaped with modified partial ground plane (Type-II), ellipticshaped with partial rectangular ground plane (Type-III), elliptic-shaped with modified partial ground plane (Type-IV), modified elliptic-shaped with partial rectangular ground plane (Type-V), and modified elliptic-shaped with modified partial ground plane (proposed antenna).
Fig. 3. Numerically simulated variations of reflection coefficient with frequency for different geometries (Type-I, Type-II, Type-III, Type-IV, Type-V and Proposed) of the monopole antennas. 2.2.
Results and discussion
2.2.1
Reflection coefficient-frequency characteristics
Fig. 4 shows numerically simulated and measured variations of reflection coefficient of the proposed modified elliptic-shaped monopole antenna with frequency. From Fig. 4, it can be observed that simulated -10 dB reflection coefficient bandwidth of the proposed antenna covers the frequency range 1.0 – 2.6 GHz. It can also be observed from Fig. 4 that around simulated frequency of 5.7 GHz, unwanted harmonics are present. Because of the presence of unwanted harmonics, signal interference occurs in a communication system. Also, it affects the S/N performance of the receiver circuitry of the communication system. Hence, in order to achieve better S/N performance of receiver circuitry in a wireless communication system, out-of-band harmonics should be suppressed. The magnitudes of reflection coefficient of the proposed antenna were measured using Anritsu vector network analyser (VNA) Master (Model: MS2038C). The measured -10 dB reflection coefficient bandwidth of the proposed antenna covers the frequency range 1.01 – 2.50 GHz. It can be seen from Fig. 4 that the simulated and measured results are in agreement with each other excepting minor variations over the frequency range of interest due to fabrication tolerances and measurement errors.
Fig. 4. Numerically simulated and measured variations of reflection coefficient of the proposed antenna with frequency.
Fig. 5. Numerically simulated and measured radiation patterns of the proposed antenna for three different frequencies of 1.1, 1.8, 2.5 GHz.
2.2.2
Radiation patterns and peak realized gain-frequency characteristics
The simulated and measured co- and cross-polarized radiation pattern plots of the proposed antenna in E- and H-planes for three different frequencies (1.1, 1.8, 2.5 GHz) are presented in Fig. 5. From Fig. 5, it can be observed that the proposed antenna exhibits good omnidirectional co-polar radiation patterns with low cross-polarization levels at the frequencies of interest. The cross-polarization levels in E-plane are found to be less as compared to the corresponding levels in H-plane at a given frequency. Fig. 6 shows the simulated and measured realized gain values of the proposed antenna with frequency. The simulated (measured) realized gain values vary in the range 2.2 – 3.5 dB (1.9 – 3.3 dB) over the operating frequency range 1.0 – 2.6 GHz (1.01 – 2.50 GHz).
Fig. 6. Numerically simulated and measured gain values of the proposed antenna as functions of frequency. 2.2.3
Simulated total efficiency-frequency characteristic
The total efficiency of an antenna includes the losses at the input terminals as well as the losses within the structure of the antenna. Fig. 7 shows the simulated variation of total efficiency of the proposed antenna with frequency. From Fig. 7, it can be observed that values of total efficiency vary in the range 86 – 93 % over the desired passband 1.0 – 2.6 GHz. In the undesired frequency range, the total efficiency initially decreases but starts increasing from 3.3 GHz and peaks at 5.7 GHz, where its value reaches to ~ 94 %.
Thereafter, it shows decreasing trend upto 7.2 GHz and finally more or less increasing trend in the frequency range 7.2 – 8.0 GHz.
Fig. 7. Simulated total efficiency of the proposed antenna as a function of frequency. Table 2. Comparison of the proposed monopole antenna with the antennas reported in the literature. Reference
Lower cut-off frequency, fl (GHz)
Physical size (mm × mm)
Electrical size (λl × λl)
[5]
2.69
50 × 42
0.448 × 0.376
[6]
1.02
110 × 124
0.374 × 0.42
[7]
1.05
80 × 74
0.28 × 0.259
[8]
2.8
55.9 × 33
0.52 × 0.308
Proposed antenna
1.01
78.8 × 47
0.265 × 0.16
Realized gain (dBi)
Total efficiency
~0 at 2.69 GHz, and 0.5 at 4 GHz 0.5 at 1.02 GHz, and 1.5 at 5 GHz
N. A.
N. A. 2.32 at 2.9 GHz, and 1.21 at 3.4 GHz 2.2 – 3.5
96 % at 1.02 GHz, Minimum 76 % N. A. Minimum 85 % 86 – 93 %
λl is the free space wavelength at the lower cut-off frequency (fl). N.A. stands for ‘not available’ 2.2.4
Performance comparison of the proposed antenna with those reported in literature
The dimension and performance based comparison of the proposed monopole antenna with those reported in the literature is provided in Table 2. It is apparent from Table 2 that overall electrical size of the proposed antenna (in terms of wavelength, λl) is smaller than the antennas reported in the literature. The proposed antenna provides almost stable radiation
patterns with reasonably good realized gain values over its operating frequency band. However, the proposed antenna needs improvement in terms of cut-off performance in desired frequency band and out-of-band (unwanted) harmonics. 3.
The modified interdigital bandpass filter (IBPF)
3.1
Design of modified IBPF
In order to achieve sharp cut-off performance and wide stopband with reasonably suppressed out-of-band unwanted harmonics for the proposed MPA, the compact modified IBPF reported in [18] by us is used. Certain design principles were considered while designing the modified IBPF for its integration with the proposed MPA in order to obtain an L-band filtering antenna for applications in L-band wireless communication. Firstly, the dielectric substrate employed for the BPF should be same as that used for the antenna. Secondly, the characteristic impedance of the microstrip line feeds at the input and output ports of the filter must be equal to 50 Ω i.e. the microstrip line width at the input and output ports of the filter should meet 50 Ω impedance criterion. Thirdly, the -10 dB reflection coefficient bandwidth of the filter should cover L-band frequency range in order to finally obtain the L-band filtering antenna after its integration with the proposed MPA. The dielectric substrate, RT/duroid 6010, having dielectric constant of 10.2 and thickness of 1.27 mm was used for design of the modified IBPF. Fig. 8(a) shows the perspective view of the BPF having spurlines and defected ground structures (DGSs). Fig. 8(b) and 8(c) show the top and bottom views of the BPF. The band notched characteristic of spurline, and harmonic suppression along with miniaturization characteristic of DGS is responsible for compactness along with wide stopband characteristics of the reported BPF. For the sake of brevity, detailed design procedure, which is available in [18] is not explained here. The geometrical parameter values of the BPF for L-band operation are given in Table 3. The size of the compact BPF is 20.2 mm × 9.0 mm × 1.27 mm.
3.2
Results and discussion
The simulated variations of the magnitude of S-parameters of the BPF with frequency are shown in Fig. 9. From Fig. 9, it can be observed that -10 dB reflection coefficient bandwidth of the BPF is 1.05 GHz (= 71.6 %) and covers the frequency range 0.94 – 1.99 GHz. The 3dB passband frequency range of the BPF is 0.89 – 2.05 GHz. As far as stopband performance is concerned, the upper stopband attenuation level is better than 17 dB up to 8 GHz. The maximum value of simulated insertion loss at a mid-band frequency of the filter passband is found to be 0.35 dB.
Fig. 8. Geometry of the BPF (a) Perspective view, (b) Top view, and (c) Bottom view. Table 3. Geometrical parameter values of the BPF (all dimensions are in millimetre) l1 = 19.3 w1 = 1.8 l2 = 17.3 w2 = 1
l3 = 17.2 w3 = 2.4 g12 = 0.2
g23 = 0.3 d1 = 0.4
d2 = 0.8
l4 = 1.6
l5 = 2.8
l6 = 2.7
w4 = 0.9
l7 = 0.8
l8 = 2.6
l9 = 0.3
g2 = 0.4
w5 = 0.4 l10 = 3.7
l11 = 0.9
l12 = 2.5 l13 = 1.3
g1 = 0.7
w6 = 0.2 l14 = 8.4
4.
Proposed filtering antenna
4.1
Design and investigation of the proposed filtering antenna
In order to obtain an integrated system providing appropriate filtering and radiation characteristics for L-band applications, the L-band compact modified IBPF is integrated with modified elliptic-shaped MPA. The L-band integrated system is called as L-band filtering
antenna. The L-band integrated system/component is designed due to various advantages and applications of L-band frequency range (1 – 2 GHz). The GPS signals are used in L-band of frequency spectrum because L-band wave can penetrate clouds, fog, rain, storm and vegetation, and it requires less expensive hardware and smaller size components/system. Lband is also used for satellite communication, mobile communication, and aircraft surveillance.
Fig. 9. Numerically simulated variations of the magnitude of S-parameters of the BPF versus frequency [18]. For design of the integrated system, certain design principles were considered during optimization of its geometrical parameters. Firstly, the individual components should be designed on the same dielectric substrate i.e. RT/duroid 6010 substrate (εr = 10.2, tanδ = 0.0023, and thickness = 1.27 mm) in the present case. Secondly, the characteristic impedance of the microstrip line feeds at the input and output ports of the filter must be equal to 50 Ω in order to meet the 50 Ω impedance criterion at the input port of the antenna. Thirdly, the -10 dB reflection coefficient bandwidth of the filter should cover L-band frequency range i.e. 1 – 2 GHz in order to finally obtain the L-band filtering antenna after its integration with the proposed MPA. The geometrical parameters of the connecting transmission lines of the antenna and the filter are optimised through numerical simulation in order to match their input impedance with the common reference impedance of 50 Ω. Fig. 10 (a) shows geometry
of the proposed filtering antenna obtained through integration of modified elliptic-shaped monopole antenna with the modified IBPF reported in [18]. The geometrical parameters of the integrated system are optimized to achieve desired performance in L-band. The optimized geometrical parameter values of the proposed integrated design of filtering antenna are given in Table 4. The prototype of the proposed filtering antenna was fabricated on RT/duroid 6010 substrate (εr = 10.2, tanδ = 0.0023, and thickness = 1.27 mm) using PCB technology for experimental study. Fig. 10 (b) shows top and bottom views of the fabricated prototype of the proposed integrated filtering antenna. The overall physical size of the proposed filtering antenna is 91.8 mm × 47 mm × 1.27 mm.
Fig. 10. (a) Geometry of the proposed integrated design of filtering antenna. (b) Top and bottom views of the fabricated prototype of the proposed filtering antenna. Fig. 11 shows the simulated variations of reflection coefficient with frequency for the proposed modified elliptic-shaped monopole antenna with and without integration of the BPF. From Fig. 11, it can be observed that the proposed integrated design (also called the proposed filtering antenna) provides sharp band-edge selectivity at lower and upper edges of
frequency band (1.01 – 1.96 GHz) along with good suppression of unwanted harmonics up to 8 GHz.
Table 4. Optimized geometrical parameter values of the proposed integrated design of filtering antenna (all dimensions are in millimetre) L = 91.8
W = 47
lf 1 = 3.5
lc = 3.9
wc = 3.7
lt = 2.7
b2 = 17.4
b3 = 5.5
g12 = 0.2
g23 = 0.3 d1 = 0.4
w4 = 0.9
l7 = 0.8
l12 = 2.5
l13 = 1.3
wf 1 = 1.7 lf 2 = 43 wt = 4
a1 = 32.6
l1 = 19.3 w1 = 1.8
l2 = 17.3
d2 = 0.8
l8 = 2.6 w6 = 0.2
wf 2 = 1.4 lg3 = 51 a2 = 7
lg4 = 51.1
a3 = 32.5
b1 = 6.2
w2 = 1
l3 = 17.2
w3 = 2.4
l4 = 1.6
l5 = 2.8
l6 = 2.7
g1 = 0.7
l9 = 0.3
g2 = 0.4
w5 = 0.4
l10 = 3.7
l11 = 0.9
l14 = 8.4
–
–
–
–
Fig. 11. Numerically simulated variations of reflection coefficient of the proposed modified elliptic-shaped monopole antenna with and without integration of the BPF versus frequency. 4.2.
Equivalent circuit model of the proposed filtering antenna
The equivalent circuit model of the proposed filtering antenna can be helpful in the computer aided design (CAD) of the RF front-end of a transceiver [20]. Fig. 12 shows the equivalent circuit model of the proposed filtering antenna which is based on a heuristic modification of the Foster's network synthesis technique [21, 22]. The resistive elements are introduced to model metal, dielectric and radiation losses.
Fig. 12. Equivalent circuit model of the proposed filtering antenna.
The input impedance of the antenna is modelled by means of the equivalent network depicted in Fig. 12. In order to describe the antenna frequency behaviour, following series expansion is adopted to describe the antenna frequency behaviour [22]
1 Zin ( ) j L0 jC0
where Qn n RnCn and n LnCn
1/2
N max
Rn n 1 jQ n 1 n n
(1)
. In Equation (1), Nmax is the number of modes
needed to properly describe the frequency behaviour of the antenna input impedance, ω is the operating radian frequency, ωn is the radian frequency of the n-th resonant mode, C0 is the quasi-static input capacitance, L0 is an inductance that takes into account higher order modes as well as probe feed effects, while Cn, Ln, Rn, and Qn are the capacitance, inductance, resistance, and quality factor respectively, describing the damped resonance processes that take place in the antenna structure. From the analysis of the circuit, C0 can be found as C0 lim 0
Im{YinEM ( )}
Im{YinEM ( )} Im{YinEM ( )} lim 0
(2)
Starting from C0, L0 can be computed as L0 1/ {[0 ]2 C0 } where ω0 is the lowest frequency for which Im{ZinEM (0 )} 0 . The values of Rn and Qn can be estimated using Equations (3) and (4) respectively. Rn Re{ZinEM (n )},
(3)
and
Qn
Im{ZinEM (n )} Im{ZinEM (n )} n Im{ZinEM (n )} n 2 Rn 4 Rn
(4)
Now, parameter values of the circuit model can be evaluated using Equations (1) - (4). As shown in Fig. 14, approximately seven resonant modes are necessary to represent the antenna frequency behaviour in L-band. The equivalent circuit parameter values are given in Table 5.
Table 5. Parameter values of the equivalent circuit model for the proposed filtering antenna. C0 = 5.25 pF
L0 = 0.02 μH
R1 = 46.353 Ω
L1 = 1.431 nH
C1 = 19.5 pF
R2 = 41.646 Ω
L2 = 1.64 nH
C2 = 16.46 pF
R3 = 47.347 Ω
L3 = 1.383 nH
C3 = 15.09 pF
R4 = 50.278 Ω
L4 = 1.457 nH
C4 = 11.33 pF
R5 = 76.614 Ω
L5 = 0.641 nH
C5 = 19.23 pF
R6 = 35.893 Ω
L6 = 1.105 nH
C6 = 7.01 pF
R7 = 77.435 Ω
L7 = 0.468 nH
C7 = 15.76 pF
4.3.
Surface current distribution
For better understanding of integration effectiveness of the BPF with the proposed modified elliptic-shaped antenna for suppression of unwanted harmonics, the current distributions on the surface of the proposed antenna with and without the BPF are studied. Fig. 13 depicts the current distributions on the surface of the proposed antenna with and without the BPF at a frequency of 5.7 GHz (a simulated frequency for unwanted harmonic without BPF). It can be noticed from Fig. 13(a) that higher level of current reaches the modified elliptic-shaped radiating patch from the input port in the absence of BPF. Therefore, in order to suppress the harmonics, the BPF was used to block the passage of current from input port to radiating modified elliptic-shaped antenna. It can be clearly observed from Fig. 13 (b) that in presence of the BPF, the current is more concentrated near the BPF, and negligible current reaches the modified elliptic-shaped antenna.
Fig. 13. Simulated surface current distributions of the proposed antenna at a frequency of 5.7 GHz (an undesired response frequency without BPF) (a) without BPF, and (b) with BPF.
Fig. 14. Numerically simulated and measured variations of reflection coefficient of the proposed filtering antenna versus frequency. 4.4.
Results and discussion
4.4.1. Reflection coefficient-frequency characteristics The numerically simulated and measured variations of the reflection coefficient of the proposed filtering antenna with frequency are shown in Fig. 14. The values of reflection coefficient of the proposed filtering antenna were measured at different frequencies using Anritsu vector network analyser (VNA) Master (Model: MS2038C). The simulated (measured) -10 dB reflection coefficient bandwidth of the proposed filtering antenna covers the frequency range 1.01 – 1.96 GHz (1.02 – 1.92 GHz). The simulated and measured results are in agreement with each other excepting minor variations occurring over the frequency band of interest due to fabrication tolerances and measurement errors. 4.4.2. Radiation patterns The simulated and measured co- and cross-polarized radiation pattern plots of the proposed filtering antenna in E- and H-planes for two different frequencies (1.1, 1.8 GHz) which are
within operating frequency band are presented in Fig. 15. From Fig. 15, it can be observed that the proposed filtering antenna has good omni-directional co-polar radiation patterns with low cross-polarization levels at the frequencies considered in the study. The proposed filtering antenna exhibits similar radiation patterns with tolerable cross-polarization levels when compared with the proposed antenna without BPF presented in Fig. 5 for same frequency. It shows that the radiation characteristics of the proposed filtering antenna are more or less maintained with good filtering characteristics over the frequency band of interest.
Fig. 15. Numerically simulated and measured radiation patterns of the proposed filtering antenna for two different frequencies. 4.4.3. Peak realized gain-frequency characteristics Fig. 16 shows the variations of simulated and measured realized gain values of the proposed antenna with and without integration of the BPF with frequency. From Fig. 16, it can be observed that the simulated and measured realized gain-frequency characteristics of the
proposed filtering antenna are nearly in agreement with each other. The simulated (measured) realized gain values of the filtering antenna vary in the range 1.99 – 2.96 dB (1.82 – 2.71 dB) over the operating frequency band 1.01 – 1.96 GHz (1.02 – 1.92 GHz). It can also be seen from Fig. 16 that drastic reduction in gain values occurs at lower and upper edges of the frequency band. Though the simulated realized gain values vary in the range 2.1 – 3.5 dB over the passband of isolated antenna without BPF similar to those for the proposed filtering antenna, change in its gain value beyond the upper edge of frequency band is not large. The proposed filtering antenna provides simulated lowest realized gain value of -18 dB at 2.1 GHz beyond the upper passband edge. In case of the isolated antenna without BPF, simulated realized gain value at 2.1 GHz is 3.1 dB, which is 21.1 dB higher than that for the proposed filtering antenna. Further, simulated realized gain value of -18 dB is obtained at 0.85 GHz below lower passband edge of the proposed filtering antenna. The simulated realized gain value for the isolated antenna without BPF at 0.85 GHz is -4 dB, which is 14 dB higher than that for the proposed filtering antenna. This shows the performance superiority of the proposed filtering antenna over the isolated antenna without BPF in terms of realized gain parameter.
Fig. 16. Numerically simulated and measured variations of realized gain values of the proposed antenna with and without BPF versus frequency.
4.4.4. Simulated total efficiency-frequency characteristic Fig. 17 shows the variations of simulated total efficiency of the proposed antenna with and without integration of the BPF with frequency. It can be observed from Fig. 17 that the simulated total efficiency of the proposed filtering antenna vary in the range 85 – 89 % within its operating frequency band but drastic reduction in total efficiency occurs on either side of passband (= 1.01 – 1.96 GHz). The simulated total efficiency values vary in the range 86 – 93 % over the passband of isolated antenna without BPF similar to those for the proposed filtering antenna. It can be stated after comparing the total efficiency-frequency characteristics of the proposed antenna with and without integration of the BPF that significant reduction in the total efficiency value occurs above the desired passband in case of the proposed filtering antenna i.e. the proposed monopole antenna integrated with the proposed modified IBPF as compared with the isolated antenna without integration of BPF. This shows the performance superiority of the proposed filtering antenna over the isolated antenna without BPF in terms of total efficiency parameter.
Fig. 17. Numerically simulated variations of total efficiency of the proposed antenna with and without integration of the proposed BPF versus frequency. 4.4.5. Performance comparison of the proposed filtering antenna with those reported in literature Finally, performance and dimension based comparison of the proposed filtering antenna with those reported in the literature is given in Table 6. It is apparent from Table 6 that overall
electrical size of the proposed filtering antenna (in terms of wavelength, λl) is smaller as compared with the filtering antennas reported in the literature. The proposed filtering antenna also provides good harmonic suppression with improved selectivity. Table 6. Performance comparison of the proposed filtering antenna with filtering antennas reported in the literature.
Reference
[11] [14] [15] [16] [17]
Measured -10 dB reflection coefficient bandwidth (FBW) 4.06 – 4.26 GHz (4.7%) 2.32 – 2.55 GHz (9.4%) 2.26 – 2.66 (16.3%) 1.99 – 2.07 GHz (3.5%) 2.5 – 3.3 GHz (27.5 %)
Measured harmonic suppression Not available Upto 3.5 GHz Upto 3.5 GHz Not available Upto 14 GHz
Physical Size (mm × mm) 20 × 40 70 × 75 30 × 45 85 × 80 53.3 × 34.2
Electrical Size (λl × λl) 0.27 × 0.54 0.54 × 0.58 0.226 × 0.339 0.56 × 0.53 0.44 × 0.285
Realized Total gain efficiency (dBi)
1.4 – 4.3
N. A.
3.3
N. A.
2.41
N. A.
5
N. A.
2.5
N. A.
Proposed 1.99 – 85 – 89 1.02 – 1.92 0.31 × 2.96 % filtering Upto 8 GHz 91.8 × 47 GHz (61.2%) 0.16 antenna λl is the free space wavelength at the lower cut-off frequency (fl). FBW stands for ‘fractional bandwidth’. N.A. stands for ‘not available’.
5.
Conclusion
A compact filtering antenna having sharp cut-off performance and highly suppressed unwanted harmonics has been described. The proposed filtering antenna has been obtained through integration of the modified elliptic-shaped monopole antenna with the modified IBPF reported in the literature. The proposed filtering antenna is compact in size with good bandedge selectivity and suppressed out-of-band harmonics up to 8 GHz. The proposed filtering antenna and the antenna without optimized modified IBPF have been investigated through numerical simulation and measurements and the results have been found to be nearly in agreement with each other. It is finally concluded that the proposed compact filtering antenna
can be a good candidate for various L-band wireless communication applications, such as satellite communication, global positioning system, and aircraft surveillance system.
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Bhagirath Sahu received the B.E. degree in Electronics and Communication Engineering from R.G.P.V., Bhopal, India, in 2010, and the M.Tech degree in Electronics Engineering (Microwave Engineering) from IIT (BHU), Varanasi, India, in 2013. He is currently working towards the Ph.D. degree in Electronics Engineering (Microwave Engineering) from IIT (BHU), Varanasi, India. He is author or co-author of about twelve Journal papers and twenty international/national conference papers. His research interests include microstrip filters, filtering antennas, dielectric resonator antennas, antennas for wireless/biomedical applications and metamaterials. Mr. Sahu is a student member of the Institute of Electrical and Electronics Engineers (IEEE) Microwave Theory and Techniques Society (MTT-S). Soni Singh was born in Deoria, UP, India in 1989. She received the B.Sc. degree in Electronics from DDU Gorakhpur University, India in 2009 and M. Sc. degree in Electronics from Banasthali University, Rajasthan, India in 2011. She completed her Ph.D. work in 2016 from the Department of Electronics Engineering, Indian Institute of Technology (Banaras Hindu University), Varanasi, India. She is author or co-author of twelve journal papers and twenty international/national conference papers. Her major research activity includes dielectric loaded metal horn antenna, medical applications of microwaves, and antenna technology. Manoj Kumar Meshram received the Ph.D. degree from the Department of Electronics Engineering, IT, BHU (now IIT (BHU)), Varanasi, India, in the year 2001. In August 2001, he joined as Assistant Professor in the Department of Electronics and Communication Engineering at DIT, Dehradun, India. In July 2002, he joined as a Lecturer in the Department of Electronics Engineering, IT, BHU, Varanasi, India, where presently, he is working as Associate Professor. His research interests include microwave and millimeter wave technology, meta-material, reconfigurable antennas, frequency scanning antennas for radar system and microwave imaging. He has published about hundred papers in International Journals. Dr. Meshram is a Senior Member of the IEEE, member of Indian Science Congress Association, India, Life Member of Indian Society of Technical Education (ISTE), India, and a Life Member of Institution of Electronics and Telecommunication Engineers (IETE), India. S. P. Singh (corresponding author) received the Ph.D. degree in Electronics Engineering (Microwave Engineering) from BHU, Varanasi, India, in 1989. He had been employed at Department of Electronics Engineering, IT, BHU (now IIT (BHU)), Varanasi since 1980 and retired as Professor on January 31, 2016. Presently, he is re-employed as Institute Professor in the same institute. He has been Coordinator of the UGC funded CAS program during 2005–2011, and Head of the Department during 2010–2013. He has supervised/co-supervised fourteen Ph.D. theses and sixty five M.Tech. Dissertation projects. He has authored/coauthored over two hundred research papers in international/national journals and conference proceedings. His areas of current research and publications include bio-electromagnetics, microwave antennas and microwave measurements. Prof. Singh is a Life Senior Member of IEEE, USA and a Life Fellow of IETE, India.