A dual-bit low-offset sigma delta analog-to-digital converter for integrated smart sensors

157

Sensors and Actuators A, 36 (1993) 157-164

A dual-bit low-offset sigma delta analog-to-digital integrated smart sensors F R RlediJk,

G Rademaker

converter for

and J H HuiJsing

Electronrc Instrumentatmn Laboratory, Department of Eiectrrcal Engmeermng, Derfr Unruersrty of Technology, PO Box 5031, 2600 GA Deift (Netherlands) (Recewxl

January

7, 1992, m rewed

form October 30, 1992, accepted November 27, 1992)

Abstract A signal-condltlomng clrcult IS presented which converts the small signal of a thermal-flow sensor mto a dlgltal slgnal A balanced chopper with bipolar transistors combined ulth a class B sigma delta converter keeps the offset beneath 10 pV Although the clrcmt has been designed especially for a thermal-flow sensor wth thermopdes, it LS well smted for other small-slgnal apphcatlons

Introduction SENSOR -

In industrial electronics there 1s an increasing demand for sensors capable of commumcatmg wth a microcontroller Conventional sensors require separate signal-condltlomng electronics and a separate analog-to-digital (A/D) converter, which makes the total system expensive In recent years, a large number of integrated &con sensors have become avallable Slhcon sensors offer the designer the opportumty of integrating slgnal-condltlomng electronics and an A/D converter together with the sensor on one chip These types of sensors are called integrated smart sensors They can be made suitable for direct commumcatlon with a microcontroller, which offers more advanced commumcatlon [l] Many (integrated) sensors generate only a small output signal, which can easily be spoiled by mterference or maladjustment Smart sensors can accurately buffer these small signals with their electronic circuits directly onto the chip [2] This paper presents an integrated electronic system, which converts the small signal of an mtegrated thermal-flow sensor mto a digital code The schematic is shown m Fig 1 The thermoplle elements of the flow sensor deliver a voltage m the microvolt range, without offset A novel balanced bipolar chopper removes the amplifier offset The A/D converter 1s Implemented as a class B version of the well-known sigma delta converter, also for offset reduction purposes 0924~4247/93/$6 00

CHOPPER

-AMPLIFIER

-

“’ CONVERTER

Fig 1 A block scheme of the proposed system

The principle of the thermal-flow sensor ~111be explained first, followed by a spotlight on some general demands concernmg the integration of electronic circuits on a sensor chip Fmally, the subsequent elements of the block scheme will be worked out

The thermal-flow sensor

A well-known method to measure flow with an integrated sensor 1s by using flow-induced temperature differences [3] The entire sensor chip, shown m Ftg 2, 1s heated by resistors to approximately 10 K above the ambient temperature Flow across the heated chip cools the front end more than the back end of the chip A temperature gradient occurs on-chip, whose magnitude depends on the flow velocity m a way described by AT = CTc,(iJ)“2

(1)

where U ts the magmtude of the flow, TCA 1s the temperature between chip and ambient flow and C 1s a constant depending on the physical parameters The small temperature differences are measured with integrated thermoplles The largest tempera@ 1993- Elsevler Sequoia All nghts reserved

158

integrated components and can even restrict the avallablhty of some types of components (5) The conversion of sensor signals mto digital form means the presence of dlgltal signals on the same chip Digital interference on the sensor analog electronics can cause large errors Chopper and amplljier

Fig 2 A mwophotograph resistors and thermopks

of the thermal flow sensor wth heatmg

ture difference can be found between both edges of the chip, which explams the composltlon of the sensor, with heatmg resistors m the middle and the thermoplles at the outside Thermoplles are basltally generating sources (hence wlthout offset) and have a relatively high sensitivity compared to other methods [4] Still, due to the small temperature signals, the output signal of, for example, a PSI-Al thermoplle will not exceed a few mllhvolts, which forces the electronic circuits to be able to dlstmgulsh between mlcrovolts

Design consideration of additional electronics Designing electromcs for use on a smart sensor chip ts often multi-faceted compared to other apphcatlons The design 1s mfluenced by unusual demands, followmg from the special smart-sensor environment The most important demands are (1) The output signals of the sensor are very small Electromc clrcults should often be able to handle signals m the rmcrovolt range to obtain substantial sensltlvlty (2) The output signals of the sensor are often slowly varymg Clrcultry where accuracy and resolution can be traded off against speed 1s very suitable (3) The sensor often consumes considerable chip space, which restricts the chip space avallable for the electromc clrcmts (4) Special process steps that are needed for the sensor can worsen the speclficatlons of normal

For the design of the amplifier, the first demand 1s by far the most significant Any Integrated amplifier can introduce an input offset much higher than the sensor signal itself The offset of the best possible input configuration, a quad bipolar stage, can only go down to about 0 1 mV This 1s still too high for handhng mmlmal sensor signals of a few mlcrovolts This problem has been solved by using a chopper Although the lmplementatlon of a voltage chop function 1s straightforward using MOS or JFET transistor switches, the lack of those components m the process used for the thermal-flow sensor forced us to design a new bipolar chopper In combmatlon with this bipolar chopper, a balanced voltage-to-current converter 1s used to buffer the sensor signal and convert It to a slgnal that 1s suitable as input for the analog-todigital converter Analog-to-dtgltal converter

Consldermg the foregoing demands, It 1s not surpnsmg that charge-balancmg converters became so popular for smart-sensor applications [ 5-71 Their prmclples of operation match the demands nearly perfectly They are simple and ease the chip-space consumption problems They are independent of component behavlour and convert the input signal wlthm a few mllhseconds, which 1s sufficient for most sensor apphcatlons and especially for the described thermal-flow sensor Their input mtegrator reduces noise m general and high-frequency dlgltal interference noise m partlcular A dlstmctlon has to be made between freerunning charge balancing, such as frequency or converters, duty-cycle, and clock-synchronized such as dual slope and sigma delta converters Free-runmng time signals are easily synthromzed with mterfermg digital signals, which can lead to large quantrzatlon errors In contrast, synchronized converters offer a total suppression of such synchromzmg Interference, which makes them more suitable for apphcatlons wth dlgltal signals on the sensor chip [8]

Basic synchronized charge-balancmg converters can only convert unipolar input signals, simply because the input signals are balanced with a umpolar reference current However, many sensors deliver bipolar signals A simple technique to handle bipolar signals IS to bias the converter with half of the reference current In that case the input conversion range IS divided equally between negative and positive signals A drawback of these class A converters IS then large offset due to the necessary bias current For this reason we shall mtroduce a new type of charge-balancing converter, which IS based on class B prmclples Therefore it has very low offset

only the sensor signal component In this way the offset IS reduced slgmficantly Because of the fact that MOSFETs are not available m the process m which the sensor ~11 be fabncated, a new bipolar voltage chopper, which IS shown m Fig 3, has been used Theoretically a saturation voltage VZatof some tens of m&volts can be obtained between the collector and emitter of normal-mode-driven blpolar transistors However, transistors dnven m reverse mode offer a much lower Vsat The saturation voltage of reversed-mode-driven transistors can be expressed as ff sat =IR b

Bipolar voltage chopper The small output slgnal of the thermoplle should of course be buffered and amplified first before the signal can be dlgtlzed However, m order not to spoil the resolution, the relatively hrgh offset of the amplifier should also be cancelled The normal way of offset cancellation IS chopping the sensor signal with respect to the offset First the total signal IS measured and next only the offset component of the signal IS measured by shortening the input of the amplifier Both results are to be subtracted, which leaves “CC

-r

Rg

3 A new balanced bipolar chopper, which uses mverted translstors

c

-cln 4

(2)

where & is the drive current, R, ISthe collector bulk resistance and pf IS the forward beta of the transistor As can be seen from eqn (2), larger transistors offer a smaller Vsat because of their reduced collector bulk resistance R, However, the maximum size IS limited because of mcreasmg swltchmg delays caused by the large Junction capacitances An optimally dimensioned configuration can offer a saturation voltage down to 0 3 mV Such a voltage drop over the switches cannot be tolerated The bipolar transistors have to be used m a balanced configuration m order to compensate for

160

the saturation voltages However, saturated transistor switches always need a current dram, hence they are basically non-floating A balanced chopper configuration would therefore not be amplementable with a grounded sensor slgnal source as mput However, a Seebeck element essentially generates a floating voltage, which allows us to design an alternative balanced chopper with bipolar transistors The remaining offset of this balanced configuratlon depends on the mlsmatchmg of the components and will normally be the order of 10 pV Thus this novel bipolar chopper can compete with JFET or MOS voltage choppers [9]

would lead to an offset of about 300 pV This offset will be cancelled by the chopper However, there are two other offset sources which cannot be cancelled out They are caused by loading the thermoplle with the voltage-to-current converter By using a balanced sensor, chopper and voltage-to-current converter, this load 1s mmlma1 and only dependent on small mismatches of the components Two mismatches are most sign&ant The first mismatch 1s caused by different Seebeck impedances The input current of the voltageto-current converter flows through these impedances and causes a differential voltage drop

Voltage-to-current

V,,, = AR,&,

converter

A voltage-to-current signal-condltlonmg block 1s required because the sensor produces an output voltage and the charge-balancing A/D converter needs an mput current The voltage-to-current converter 1s a balanced type and 1s shown m Fig 4 [lo, 111 The converter consists of two voltage followers, which transfer the differential input voltage to a conversion resistor R Although the converter 1s well balanced, offset due to mismatch of components and parameter variation, especially of the PNP mn-rors and input NPN transistors, will remam Normal mismatch

Fig 4 The balanced voltage-to-current

converter wth base-current

(3)

This error can be kept low by using symmetrical sensor thermoplles In that case the mismatch 1s only caused by the gradient of resistance over the chip The second mismatch 1s due to different currents flowing through the Seebeck impedances This will cause an offset voltage across these Impedances, given by Km = R, AL,

(4)

This error 1s kept low by usmg a base-current compensation clrcmt, where the base current 1s measured internally and fed back to the mput termmals of the amplifier, as 1s shown m Fig 4

compensation

161

In general the two offset sources lead to a remaining offset component of about 5 pV

Dual-bit sigma delta A/D converter In the foregoing we proposed the use of a synchromzed charge-balancing converter for use with smart sensors The sigma delta A/D converter 1s one of the most suitable types from this group because of Its snnple structure The drawback of conventional bipolar sigma delta A/D converters ts their relatively large offset This 1s because most sigma delta converters are basically class A types [ 6, IO] Offset 1s a severe drawback, especially when we want to use the converter for dlgltmng small sensor signals A novel way of dealing with these offset problems 1s using a class B approach Such an approach IS found m a dual-bit sigma delta converter Basically the forward path consists of a two-bit A/D converter, while the feedback path ~111contam

a two-bit D/A converter In audio apphcatlons such configuratlons are apphed for enhancement of the resolution [ 121 We shall use it for reduction of the offset by avoiding the class A blasmg problem The block diagram of the dual-bit sigma delta converter 1s shown m Fig 5 The inherent low offset of this configuration follows from Its performance at a zero mput In that case the capacitor will not be charged or discharged so no compensation action IS needed Both outputs Lo and Hi are Inactive and the two reference currents I,, and IrZ, which are opposite m sign, remam swltched off Because the converter 1s mactlve with low input signals, the offset will be determined entirely by the input currents of the comparators, which can be kept small The performance for non-zero mput signals 1s slmllar to that of a single umpolar sigma delta converter Connder, for example, a positive 1, chargmg the capacitor C, The voltage T/c, across C, will rise and cross the reference level V,,, At this moment the comparator C,,, turns over Then

VFFhi

Vclock rlIrlr-lIIr--l~ -

-

t

-

UP ,t

Rg

5 The total cmmt

mcludmg the low-offset dual-blt sigma delta converter

162

at the command of the clock the flip-flop FF,,, takes over the state of C,,, At the same time the output HI becomes acttve and Zr2will be switched on Zr2 discharges C, now and VcSdrops under the reference level V,, , and Ch, falls back mto its mltial state The next posltlve clock edge FFh, will switch off Zr2 and the output HI ~111become inactive agam From this operation prmclple It can be seen that the rate of the output pulses IS a measure for the mput signal So Z, can be expressed as a function of Zf by Z, = nZ,/N

(5)

where n IS the number of output pulses durmg a fixed time interval N The clock frequency ,f, IS 500 kHz For negative input signals the complementary second part of the clrcult 1s used for the conversion m a similar way By applying the dual-bit sigma delta converter, bipolar signals can be converted without using a biasing current for the converter This class B approach offers a small offset, because both the negative and the positive paths are mactlve at small input signals The polarity of the input signal can be determined from the fact that the HI output or the Lo output IS active By applying an up/down counter, the HI output ~111give count-up-pulses and the Lo output will give count-down-pulses to the counter Thus the output number will be available m two’s complement representation

Integrated circuit The electronics of the smart sensor chip have been realized m a bipolar process at a process facility of the Elcoma Group of Phlhps NiJmegen A microphotograph of the cn-cult IS shown m Fig 6 The capacitor, used m the charge-balancing A/D converter, can easily be seen The total chip area IS 1 5 mm x 2 4 mm, mcludmg bonding pads This size will not give any problems when sensor and electronics are actually integrated together on one chip Addition of the counter and dlgltal logic could give problems concermng the shortage of chip area However, for accuracy reasons there IS no real need to integrate this part on the sensor chip This function IS very suitable for lmplementation in a microprocessor

Rg

6

A mcrophotograph

of the signal-condmmng

crcult

The analog path, that IS the chopper and voltageto-current converter, has been laid out m a balanced way The comparator and Q-flop have been combined mto a master-slave flip-flop which performs the same function, but consumes less chip area Clock input and both output signals are TTL compatible, which digital standard 1s supported m many microprocessors

Performance The system that has been tested consists of a thermal-flow sensor with integrated thermoplles, the described integrated signal-condltlomng chip and a breadboard clrcmt containing the necessary digital parts The tests have been performed at room temperature, a supply voltage of 5 V and a clock frequency of 500 kHz Without flow, the remammg offset of the total clrcult was only 10 pV, correspondmg to the mentioned expectations The circuit has also been measured with the sensor disconnected Voltages up to 2 mV were converted m a linear way, with a useful resolution of 2 pV or 12 bits Figure 7 shows the capacitor signal and output pulse-rate signal with an input voltage of 0 1 V Typical cn-cult features are presented m Table 1 Flow measurements have not been performed with this system Measurements on the thermalflow sensor have been done earlier and corresponded to eqn (1) [2] Because the total system still consisted of three separate components, any measurements now would not lead to extra mformatlon They would also not give reahstlc mformatlon about the behavlour of the smart sensor,

163 Mode

Range Delay Graph 1

IReal Time I 0 00000 s [ 1 I

[Ghan 2

10 0

mV/dlv

Auto Scale Reference Sampling

@

I I

Perlod Left kHz

1 00 ms/dlv

178 nv

1 1

100

0 000 s

tlomng chip to undergo a test run and the Dutch Foundation for Technical Sciences for their financial support

I

References 1 S Mlddelhoek, P J French, J H HulJsmg and W J Llan, 2 3 Fig 7 Capacitor TABLE

and output waveforms at a typlcal mput slgnal

I TypIcal values at

Power supply voltage Power consumption Input voltdge range Resolution Total offset Output code Clock frequency ConversIon tune (a

1I bit

T, = 300 K

f5V cl

mw

+ZmV 2PV IOpV 11 bit 2’s complement 500 kHz 4ms

with thermal-flow sensor and signal condltlonmg combined on one chip, which will be the future design However, the high dlgltal Interference rejection of the total system does not stress a successful combmatlon of sensor and electronics In addltlon to this, clrcmtry symmetrically placed on the smart sensor chip 1s not likely to interfere with the thermal characterlstlcs of the sensor, as its power consumption is at least a hundred times smaller than that of the thermal sensor

4 5

Sensors with a dlgtal or frequency output, Sensors and Actuators, lS(1988) 119-133 J H Huysmg, SIgnal condltlonmg on the Sensor chip, Sensors and Actuators IO (1986) 219-237 B W van Oudheusden and J H Huysmg, Integrated slhcon flow dIrectIon sensor, Sensors and Actuators, 16 (1989) lO9- I19 A W van Herwaarden, The Seebeck effect m s~hcon ICs, Sensors and Actuators, 6 (1984) 245-254 G C M MetJer, R van Gelder. V Nooder, J van Drecht and H Kerkvhet, A three-termmal wlde-range temperature transducer with mlcrocomputer mterfacmg, ESSCIRC 1986. Del@ Nefher-

landi 6 J Robert, G C Temes, V Valenclc, R Dessoulavy and P Deval, A I6 bit low voltage CMOS A/D converter, IEEE J Soled-Smte Cwcutts SSC-22 (1987) 157m 163 7 J H HugsIng, G A van Rossum and M van der Lee, Two-wire bridge to frequency converter IEEE J So/&Smte Cmutrs SSC22 (1987) 343-349 8 F R RledlJk and J H HulJsmg, An Integrated absolute temperature sensor with pulse rate output, Tech Dlgesr 6th Int Conf S&d-State Sensors and Actuators (Transducers 91), San Franczxo, CA, USA, June 24-28, 1991, pp 479-482

9 J de Brum, Temperature difference to frequency converter for flow measurements Internal Report (request A87-I3), Electromc Instrumentation Laboratory TU Del0 10 R J van de Plassche dnd R E J van der Graft, A five-dlglt analog dlgttal converter, IEEE J Sohd Srare Czrcurts, SSC-12 (1977) 656-662

II S Pookalyaudom, Integrable electromcally vanable general-reslstance converter, IEEE Tram C~rcurrs Syst , CAS-2.5 (1978) 344353

I2 R W Adams, Design and Implementation of an audio IS-bit analog-to-dlgltal converter usmg oversamplmg techmques J Audio Eng Sot , 34(3) (Mar 1986)

Conclusions

Biographies

The reahzatlon of a multi-bit sigma delta A/D converter has been described The low offset ( 10 pV> has been obtamed by using a novel bipolar voltage chopper The circuit can be combmed with a thermal-flow sensor on one chip The output signal of such a smart sensor 1s directly compatible with a microprocessor or a dlgltal bus system

Frank R Rtedyk was born m Delft, The Netherlands, in 1965 He received his M SC degree m 1988 from the Delft Umverslty of Technology Smce then he has been a STW research asslstant at the Electronics Instrumentation Laboratory, and IS working towards his Ph D degree on the sub@ of smart sensors and a dlgltal sensor bus

Acknowledgements The authors wish to thank the Elcoma Group of Phlhps Nqmegen for allowmg the agnal-condl-

Geerlof Rademaker was born m Vlaardmgen, The Netherlands, m 1966 He received his M SC degree m 1989 from the Delft Umverslty of Technology After that he became a research assIstant at the Electromc Instrumentation Laboratory on

164

the topic of low-offset sigma delta converters Currently he 1s workmg for Peekel automation m Rotterdam Johan H Huyszng was born m Bandung, Indonesia, on May 21, 1938 He received an M SC degree m electrical engmeermg from Delft Umverslty of Technology, Delft, The Netherlands, m 1969, and a Ph D degree from the same university m 1981 for work on operational amplifiers Since 1969 he has been a member of the Research and Teaching Staff of the Electronic Instrumentation

Laboratory, where he 1s now professor of electronic instrumentation He teaches courses on electrical measurements techniques, electronic mstrumentatlon, operational amplifiers and analog-to-dlgltal converters Ha field of research 1s analog clrcmt design (operational amplifiers, analog multlphers, etc ) and mtegrated smart sensors (signal condltlomng on the sensor chip, frequency and digital converters which incorporate sensors, bus interfaces, etc ) He 1s the author or co-author of some 70 sclentlfic papers and 12 patents